

























































































DECLASSIFIE D 
By authority Secretary of 

SEP 2 3 1960 

Defense memo 2 August 1960 
LIBRARY OF CONGRESS 


WJ1 ‘nserarJITTON: BEFORE SERVICING 



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D °CVMENTAU G m ANY PAHT Wt ' 

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SUMMARY TECHNICAL REPORT 
OF THE 

NATIONAL DEFENSE RESEARCH COMMITTEE 


SEP 2 3 I960 

defense memo2 a W( lm 

******* congr ess 


This document contains information affecting the national defense of the 
United States within the meaning of the Espionage Act, 50 U. S. C., 31 and 
32, as amended. Its transmission or the revelation of its contents in any 
manner to an unauthorized person is prohibited by law. 


This volume is classiin accordance with security 
regulations of the War and Navy Departments because certain chapters 
contain material which was at the date of printing. 

Other chapters may have had a lower classification or none. The reader is 
advised to consult the War and Navy agencies listed on the reverse of this 
page for the current classification of any material. 



Manuscript and illustrations for this volume were prepared for 
publication by the Summary Reports Group of the Columbia 
University Division of War Research under contract OEMsr-1131 
with the Office of Scientific Research and Development. This 
volume was printed and bound by the Columbia University Press. 

Distribution of the Summary Technical Report of NDRC has been 
made by the War and Navy Departments. Inquiries concerning 
the availability and distribution of the Summary Technical Report 
volumes and microfilmed and other reference material should be 
addressed to the War Department Library, Room 1A-522, The 
Pentagon, Washington 25, D. C., or to the Office of Naval Research, 
Navy Department, Attention: Reports and Documents Section, 
Washington 25, D. C. 


Copy No. 


238 


This volume, like the seventy others of the Summary Technical 
Report of NDRC, has been written, edited, and printed under 
great pressure. Inevitably there are errors which have slipped past 
Division readers and proofreaders. There may be errors of fact 
not known at time of printing. The author has not been able to 
follow through his writing to the final page proof. 

Please report errors to: 


JOINT RESEARCH AND DEVELOPMENT BOARD 
PROGRAMS DIVISION (STR ERRATA) 
WASHINGTON 25, D. C. 


A master errata sheet will be compiled from these reports and sent 
to recipients of the volume. Your help will make this book more 
useful |o other readers and will be of great value in preparing any 
revisions. 





SUMMARY TECHNICAL REPORT OF DIVISION 13, NDRC 


VOLUME 2A 


COMMUNICATION 
RESEARCH 

DECLASSIFIED 
By authority Secretary of 

SEP 2 3 I960 

Defense memo 2 August 1960 
LIBRARY OF CONGRESS 

OFFICE OF SCIENTIFIC RESEARCH AND DEVELOPMENT 
VANNEVAR BUSII, DIRECTOR 

NATIONAL DEFENSE RESEARCH COMMITTEE 
JAMES B. CONANT, CHAIRMAN 

DIVISION 13 
HARADEN PRATT, CHIEF 


, jHE SERVICE 

, f part of T 

OR , » t V "classification 


WASHINGTON, D. C., 19 46 






NATIONAL DEFENSE RESEARCH COMMITTEE 



James B. Conant, Chairman 
Richard C. Tolman, Vice Chairman 
Roger Adams Army Representative 1 

Frank B. Jewett Navy Representative 2 

Karl T. Compton Commissioner of Patents 3 

Irvin Stewart, Executive Secretary 


x Army representatives in order of service: 

Maj. Gen. G. V. Strong Col. L. A. Denson 

Maj. Gen. R. C. Moore Col. P. R. Faymonville 

Maj. Gen. C. C. Williams Brig. Gen. E. A. Regnier 

Brig. Gen. W. A. Wood, Jr. Col. M. M. Irvine 

Col. E. A. Routheau 


2 Navy representatives in order of service: 

Rear Adm. H. G. Bowen Rear Adm. J. A. Furer 

Capt. Lybrand P. Smith Rear Adm. A. H. Van Keuren 

Commodore H. A. Schade 
3 Commissioners of Patents in order of service: 

Conway P. Coe Casper W. Ooms 


NOTES ON THE ORGANIZATION OF NDRC 


The duties of the National Defense Research Committee 
were (1) to recommend to the Director of OSRD suitable 
projects and research programs on the instrumentalities of 
warfare, together with contract facilities for carrying out 
these projects and programs, and (2) to administer the tech¬ 
nical and scientific work of the contracts. More specifically, 
NDRC functioned by initiating research projects on requests 
from the Army or the Navy, or on requests from an allied 
government transmitted through the Liaison Office of OSRD, 
or on its own considered initiative as a result of the experi¬ 
ence of its members. Proposals prepared by the Division, 
Panel, or Committee for research contracts for performance 
of the work involved in such projects were first reviewed by 
NDRC, and if approved, recommended to the Director of 
OSRD. Upon approval of a proposal by the Director, a con¬ 
tract permitting maximum flexibility of scientific effort was 
arranged. The business aspects of the contract, including 
such matters as materials, clearances, vouchers, patents, 
priorities, legal matters, and administration of patent matters 
were handled by the Executive Secretary of OSRD. 

Originally NDRC administered its work through five 
divisions, each headed by one of the NDRC members. 
These were: 

Division A—Armor and Ordnance 

Division B—Bombs, Fuels, Gases, & Chemical Problems 
Division C—Communication and Transportation 
Division D—Detection, Controls, and Instruments 
Division E—Patents and Inventions 


In a reorganization in the fall of 1942, twenty-three ad¬ 
ministrative divisions, panels, or committees were created, 
each with a chief selected on the basis of his outstanding 
work in the particular field. The NDRC members then be¬ 
came a reviewing and advisory group to the Director of 
OSRD. The final organization was as follows: 

Division 1—Ballistic Research 

Division 2—Effects of Impact and Explosion 

Division 3—Rocket Ordnance 

Division 4—Ordnance Accessories 

Division 5—New Missiles 

Division 6—Sub-Surface Warfare 

Division 7—-Fire Control 

Division 8—Explosives 

Division 9—Chemistry 

Division 10—-Absorbents and Aerosols 

Division 11—Chemical Engineering 

Division 12—Transportation 

Division 13—Electrical Communication 

Division 14—Radar 

Division 15—Radio Coordination 

Division 16—Optics and Camouflage 

Division 17—Physics 

Division IS—-War Metallurgy 

Division 19—Miscellaneous 

Applied Mathematics Panel 

Applied Psychology Panel 

Committee on Propagation 

Tropical Deterioration Administrative Committee 


iv 


CONFIDENTIA-I 


Library of Congress 



2015 


490941 











DOCUMENT Air ANY PAn r D C 

H5552JW JltH* i LASSl r?CAT,nl 

NDRC FOREWORD 


A s events of the years preceding 1940 revealed 
- more and more clearly the seriousness of the world 
situation, many scientists in this country came to 
realize the need of organizing scientific research for 
service in a national emergency. Recommendations 
which they made to the White House were given care¬ 
ful and sympathetic attention, and as a result the 
National Defense Research Committee [NDRC] was 
formed by Executive Order of the President in the 
summer of 1940. The members of NDRC, appointed 
by the President, were instructed to supplement the 
work of the Army and the Navy in the development 
of the instrumentalities of war. A year later, upon the 
establishment of the Office of Scientific Research and 
Development [OSRD], NDRC became one of its units. 

The Summary Technical Report of NDRC is a 
conscientious effort on the part of NDRC to summarize 
and evaluate its work and to present it in a useful and 
permanent form. It comprises some seventy volumes 
broken into groups corresponding to the NDRC Divi¬ 
sions, Panels, and Committees. 

The Summary Technical Report of each Division, 
Panel, or Committee is an integral survey of the work 
of that group. The first volume of each group’s report 
contains a summary of the report, stating the problems 
presented and the philosophy of attacking them, and 
summarizing the results of the research, development, 
and training activities undertaken. Some volumes may 
be “state of the art” treatises covering subjects to 
which various research groups have contributed in¬ 
formation. Others may contain descriptions of devices 
developed in the laboratories. A master index of all 
these divisional, panel, and committee reports which 
together constitute the Summary Technical Report 
of NDRC is contained in a separate volume, which 
also includes the index of a microfilm record of per¬ 
tinent technical laboratory reports and reference 
material. 

Some of the NDRC-sponsored researches which had 
been declassified by the end of 1945 were of sufficient 
popular interest that it was found desirable to report 
them in the form of monographs, such as the series 
on radar by Division 14 and the monograph on sam¬ 
pling inspection by the Applied 

Since the material treated inBbpimifehOfit^u^^^^ry 
in the Summary Technical Report of NDRC, the 

SEP 2 31960 


monographs are an important part of the story of 
these aspects of NDRC research. 

In contrast to the information on radar, which is 
of widespread interest and much of which is released 
to the public, the research on subsurface warfare is 
largely classified and is of general interest to a more 
restricted group. As a consequence, the report of Divi¬ 
sion 6 is found almost entirely in its Summary Tech¬ 
nical Report, which runs to over twenty volumes. The 
extent of the work of a Division cannot therefore be 
judged solely by the number of volumes devoted to it 
in the Summary Technical Report of NDRC; account 
must be taken of the monographs and available reports 
published elsewhere. 

Of all the NDRC Divisions, few were larger or 
charged with more diverse responsibilities than Divi¬ 
sion 13. Under the urgent pressure of wartime require¬ 
ments, the staff of the Division developed navigation 
and communications devices and systems which not 
only contributed to the successful Allied war effort, 
but which will continue to be of value in time of peace 
in the fields of transportation and communications. 
The work of the Division, under the direction first of 
C. B. Jolliffe and later of Haraden Pratt, furnishes a 
foundation for what promises to be even more radical 
developments than those of the war—for one example, 
direction finders which will operate at all elevations 
and azimuth angles, in other words, hemispherically. 

The Summary Technical Report of Division 13 was 
prepared under the direction of the Division Chief and 
authorized by him for publication. The report presents 
the methods and results of the widely varied research 
and development program, and, in the case of work 
with speech scrambling and decoding, it presents for 
the first time a comprehensive review of the state of 
the art. The report is also a notable record of the skill 
and integrity of the scientists and engineers, who, with 
the cooperation of the Army and Navy and Division 
contractors, contributed brilliantly to the defense of 
the nation. To all of these we express our sincere 
appreciation. 

Vannevar Bush, Director 
Office of Scientific Research and Development 

0 f J. B. Conant, Chairman 

National Defense Research Committee 


Defense memo 


V 


LIBRARY OF CONGRESS 









FOREWORD 


A r the outset of the national emergency, which 
- made it desirable to focus attention on military 
requirements in communications, developments in this 
field had reached a very advanced state in commercial 
applications. It was felt that because of this, and be¬ 
cause of the existence of many excellent research and 
development organizations in industry, it would not 
he necessary to establish any special research activity 
or central laboratory, but only to assess through Divi¬ 
sion 13 of the National Defense Research Committee 
the special problems for which military requirements 
needed solution, and present them to these already 
staffed groups. 

Naturally, many military needs differed radically 
from those encountered in civil life. There was also 
the urge to push ahead as fast as possible in the region 
of very high and ultra high frequencies where industry 
was feeling its way, being guided largely by the needs 
of commerce as such fields slowly unfolded. If the 
problems of generation, control, modulation, and re¬ 
ception in these frontier frequency areas could be 
harnessed, it was felt that results of inestimable value 
to the war effort would quickly follow. 

Another factor that influenced the types of research 
activity was the need to implement the new and 
greatly enlarged role of aircraft by giving airplanes 
the utmost in communication and in protection from 
adverse electrical weather conditions, and by utilizing 
the mobility of aircraft to create radio interference to 
enemy operations. 

Thus the Division adopted a very wide field of re¬ 
lated activities and, at the outset, initiated projects 
of fundamental importance such as development of 
ultra-high-frequency generators and measuring ap¬ 
paratus, precipitation static research to free aircraft 
radio reception from blackouts due to charged particles 
in the air, reception methods to make easy the identi¬ 
fication of strange radio signals, and propagation 


studies of radio waves on a world-wide coordinated 
basis. Many individual special studies in apparatus 
design and miscellaneous subjects were taken up as 
needs arose. 

Recognizing the need for radio interference gener¬ 
ators the Division initiated two projects on this subject 
which were carried to conclusion with successful field 
demonstrations. This general activity became so im¬ 
portant that a new Division 15 was organized which 
took over this particular activity. 

As the Division’s activities progressed and the array 
of communication, navigation, and identification needs 
became more complex, it was found that the study of 
systems as a whole required special attention. The 
problem of an adequate communication network and 
its proper integration into the early air warning sys¬ 
tems of the Army Air Forces loomed large and became 
a very important Division project. As time went on, 
the emphasis shifted more and more from equipments 
and instrumentalities to systems engineering problems. 
This led to the need of evaluating systems so that 
future planning could be done in the light of intel¬ 
ligent appraisal of the respective values of methods 
of both friend and foe. 

Towards the end of the war, this shift towards sys¬ 
tem problems led to a definite need for a central labo¬ 
ratory. The war progress in special military commu¬ 
nication developments had opened up new fields re¬ 
quiring the attention of specialists outside of existing 
laboratories. This brought about the establishment in 
1944 of Central Communications Research at Harvard 
University, where a very considerable staff of workers 
was active under the direction of Professor Chaffee, up 
to the time of demobilization. Much of this program 
continued under direct contracts between Harvard 
University and the military services. 

Haraden Peatt 

Chief, Division 13 



vii 






















































































PREFACE 


Tn summarizing the several hundred reports of con- 

tractors on the hundred-odd research projects spon¬ 
sored by Division 13 of the National Defense Research 
Committee [NDRC], the editor has had to settle in 
his own mind how much or how little of each project 
report should be included; in other words, how far 
the boiling-down process should go. 

The editor has an abhorrence for seeing good scien¬ 
tific or technical material go unpublished. Only by 
publication can the facts or methods developed by a 
few researchers become available for all researchers. 
On this basis, substantially all Division 13’s program 
should be included in the volumes, of which this is 
one, summarizing the work of the Division. On the 
other hand, time moves forward inexorably so that 
it is quite likely that, by the day of publication, much 
of the data would already be out of date. Furthermore, 
time and human energy are always scarce. On these 
bases, all that might be required would be a paragraph 
or two summarizing the aims of the project and its 
accompl ishments. 

A middle course was steered, a course between the 
easiest solution of publishing practically all of each 
report and the more difficult job of really digesting 
the project purpose and results. The editor, however, 
deliberately chose to publish too much rather than too 
little. In most cases it will be unnecessary for the 
reader to search out the original source material unless 
he wishes to dig deep into the subject. In those cases 
where fundamental information was assembled and 
printed in the project report, that is, information on 
which future research might be based, the summaries 
have been permitted to take as much space as required. 

This particular volume contains summaries of proj¬ 
ects dealing with a very wide variety of subjects and 
illustrates the editorial technique. 

Among many reports dealing with a whole series of 
projects on panoramic reception, the editor found a 
concise thesis called The Fundamentals of Panoramic 
Reception. This report is basic; all present and future 
schemes of panoramic reception draw upon the facts 
in it. The greater part of this report appears in this 
volume. On the other hand, only brief descriptions of 
actual apparatus developed by the Division using these 
principles are found here. 

Early in the life of the Division it was realized that 
the microwave region of the ether spectrum would be 
extensively used during the war. At that time very 


little was known of the propagation characteristics, 
or, in fact, of how to build microwave apparatus. 
Equipment was designed, built, and field-tested at fre¬ 
quencies of the order of 3,000 megacycles. Out of this 
came an omnidirectional system useful as an additional 
channel of communication; a highly portable trans¬ 
mitter-receiver with searchlight directivity; and the 
first field-strength measuring equipment for 3,000 
megacycles. Near the end of World War II, numerous 
systems were in operation or proposed for frequencies 
above 1,000 megacycles. An extensive analysis of all 
these systems was made by the Division. All this work 
is summarized in this volume. 

Another important subject covered here is the work 
done toward discovering the causes of and alleviating 
the effects of airplane precipitation static. Part of the 
Division’s work led to the “block and squirter” system 
for reducing radio interference; another part dis¬ 
closed the fact that uncoated airplane surfaces pick up 
less static than those which have been painted. An 
important end result was the acquisition of exact in¬ 
formation on structural changes to airplanes and their 
appurtenances, such as antennas, needed to avoid the 
generation of corona discharges except at controlled 
points. 

One of the biggest jobs of the Division was to aid 
the Air Forces in designing, procuring, installing, and 
operating airplane warning systems abroad. Under the 
general subject of “systems” research, each of the com¬ 
ponent parts was studied, and all these parts were 
properly integrated into a system. In fact, the Divi¬ 
sion was forced to aid in packaging some of the com¬ 
ponents so that they arrived in working condition at 
the European Theater of Operations! So vast was the 
work done under this general subject that only the 
merest summary could be included in this volume. 

A great amount of cooperative work on studying the 
vagaries of the ionosphere, with particular reference 
to the use of high frequencies for direction finding, 
was coordinated by the Division, all aimed at the 
production of a service prognosticating radio transmis¬ 
sion conditions on a world-wide basis. This work was 
finally taken over by the Interdepartmental Radio 
Propagation Laboratory [IRPL] and continues under 
sponsorship of the National Bureau of Standards. 

Other topics studied were the effects of trees and 
hills on radio communication in the region of 4 to 
116 megacycles; means of shielding diathermy ma- 


ix 


Confident! \ i a .. 





X 


PREFACE 


chines to reduce or prevent radio interference; the 
possibility of growing frequency-standard crystals to 
augment the natural sources of quartz; the develop¬ 
ment of a remarkably stable oscillator which needed 
no crystal at all; reconnaissance television; airplane 
facsimile equipment; the existing means of recording 
sounds on magnetic materials; the psychological bases 
on which effective jamming of enemy radio signals 
could occur—including the design and construction 
of two sets of jamming apparatus. Jamming and coun¬ 
termeasures became of such vital importance that a 
new Division (15) of NDRC was organized to carry 
forward at an accelerated pace the early work of 
Division 13. 


Among the noteworthy accomplishments of the Di¬ 
vision are the production of a means of “flash” teleg¬ 
raphy by which short radio messages can be trans¬ 
mitted and received at the rate of 3,000 to 9,000 words 
per minute; a new method of locating faults in field 
wire, and an elegant method of laying such wire by 
airplane; and the development of new batteries which 
were useful at temperatures as low as —40 F. 

Finally, the reader’s attention is directed to material 
supplementing this volume, which is published as 
Volume 2B of the Division 13 Summary Technical 
Report, Electronic Navigation Systems. 

Keith Henney 
Editor 





CONTENTS 


CHAPTER 

1 


2 


4 


o 


6 


8 

9 

10 

11 


12 

13 

14 

15 


16 

17 


18 


PART I 

GENERAL COMMUNICATIONS RESEARCH 

PAGE 

Systems Engineering for AAF Communications ... 3 

Propagation Studies . 5 

Frequency Modulation Versus Amplitude Modulation . 8 

Microwave Communication Systems.11 


PART II 

3,000-Mc COMM UNICA TION 

Field Tests and 3,000-Mc Equipment . 

P-F Generator for 2,000 to 3,000 Me .... 

PART III 

PRECIPITATION STATIC PROBLEMS 

Precipitation Static Reduction. 

Precipitation Static Research. 

Effect of Aircraft Surface Treatment .... 

Noise Eliminator Tests. 

The Block-and-Squirter System. 


19 

37 


45 

47 

49 

51 


b'7 


PART IV 

PANORAMIC RECEPTION 61 

The Fundamentals of Panoramic Reception.63 

Panoramic Receiver with Moving-Screen Indicator . . 88 

Improved Panoramic Receiver.98 

Receiver for Pulse Signals.108 

PART V 

INTERFERENCE GENERATION 117 

Study in Interference Generation.119 

Radio Interference Generators.122 

PART VI 

RADIO TRANSMISSION FORECASTING 

Ionosphere Studies.131 



xi 















CONTENTS 


PART VII 

APPARATUS DESIGN 

CHAPTER PAGE 

19 U-H-F Field-Intensity Measuring Equipment .... 137 

20 Aircraft Facsimile Systems.141 

21 Ultra-High-Speed Flash Telegraphy.144 

22 Frequency-Stabilized Master Oscillator.154 

23 Pickup Tube for Reconnaissance Television.162 

24 Sound Recording on Magnetic Materials.166 


PART VIII 

MISCELLANEOUS STUDIES 


25 Substitutes for Natural Quartz for Frequency Control . 173 

26 Shielding for Diathermy.175 

27 Locating Faults in Wire Lines.176 

28 Storage Batteries for Cold Climates.181 

29 An Electrical Cancellation and Indicating System . . 183 

30 Laying Field Wire by Airplane.191 

31 Floating Insulated Wire.192 

32 Noise Investigations.193 

33 Transient Response of Band-Pass Amplifiers .... 196 

34 Measurements of Magnetic Properties of Ferrite Core 

Materials.198 

35 Aircraft Antenna Power, Impedance, and Tuning-Net¬ 
work Survey . 200 

Bibliography.203 

OSRD Appointees.208 

Contract Numbers.209 

Project Numbers .211 

Index.213 





















PART I 

GENERAL COMMUNICATIONS RESEARCH 










Chapter 1 

SYSTEMS ENGINEERING FOR AAF COMMUNICATIONS 


A broad study of the communications requirements for a 
proposed air warning system for European and Pacific theaters, 
including improvements to existing equipment, reduction of 
radio interference problems, development of new antennas, 
field-strength surveys, study of engine-generator noise, laying 
field wire by airplane, and many other subjects important to 
combat communications. 

i i HISTORY OF THE PROJECT 

B efore the work under this project 3 - was begun, 
the Bell Telephone Laboratories [BTL] had had 
experience in providing the Army Air Forces [AAF] 
with communications systems such as the extensive air 
warning and information center networks established 
and placed in operation in the United States before 
Pearl Harbor. Alien the Army Air Forces School of 
Applied Tactics was established in Orlando, Florida, 
BTL provided a complete information center com¬ 
munications system for training purposes. 

When, therefore, plans were under way for opera¬ 
tions overseas, the Air Forces needed the same kind of 
consultation. Now, however, the equipment would have 
to be portable and much of the service would have to 
be performed by means of radio instead of by wire 
telephone plant. The Signal Corps at that time had 
many communication components primarily suited to 
ground force operation, for the most part in the high- 
frequency (h-f) band and usually for continuous-wave 
(c-w) operation. The Air Forces on the other hand 
were primarily interested in very lightweight equip¬ 
ment that would be transportable by air and would be 
suitable for voice operation so that in the Air Warning 
Service either wire or radio telephone could be used at 
will, depending upon which was available at the time. 

To implement a study of the systems aspects of com¬ 
munications, NDRC Project C-79 was set up, effective 
February 16, 1943. The objectives of this project were 
to “conduct research on engineering problems of the 
Army Air Forces communications systems and by oral 
or written communication cooperate in supplying in¬ 
formation on special engineering problems to officers 
of the Army Air Forces/’ 

The first assignment was improvement of the com- 

"Project C-79, Contract OEMsr-1018, Bell Telephone Lab¬ 
oratories, Inc., Western Electric Co., Inc. 


munications system (largely radio) for tactical Air 
Warning Service. This phase of the work was com¬ 
pleted, and equipment for systems of the type recom¬ 
mended was procured by the Signal Corps for the Air 
Forces. The second assignment was similar to the first 
but concerned the complete tactical Air Force com¬ 
munications system. Work on this problem led to the 
conclusion that each tactical Air Force would have its 
own special requirements which would have to be con¬ 
sidered individually. It was concluded that the best 
solution was to provide sufficiently complete and basic 
systems engineering information so that communica¬ 
tion officers in the field could adapt and integrate into 
a satisfactory working system equipment available in 
the theaters. 

A most valuable by-product of this effort was that 
it focused attention on the systems approach to com¬ 
munications problems and emphasized its importance 
to those responsible for Army communications. While 
the work on Project C-79 related solely to Air Forces’ 
needs, the demonstration of the systems approach re¬ 
sulted in a subsequent direct request to the Bell Tele¬ 
phone Laboratories for the preparation of systems en¬ 
gineering information on a broader scale which, in 
turn, was instrumental in initiating a project between 
the Signal Corps and Bell Laboratories to provide this 
information in the form of Army manuals. b 

Project C-79 covered a wide variety of subjects, in¬ 
cluding such problems as the avoidance of interference 
between radio sets in close proximity, the development 
of handy methods of estimating radio transmission 
over irregular paths, the use of h-f antennas adapted to 
transmitting short distances by means of sky waves, 
and the development of a new single-channel teletype¬ 
writer system which can be used on tactical radio sets 
working on a push-to-talk basis. The subjects of the 
individual reports making up the final report 1 indi¬ 
cate the wide range of problems considered under the 
project. 


b Electrical Communications Systems Engineering (TM11-486), 
preliminary issue dated February 25, 1944, revised edition, 
April 25, 1945. Much of the background for this compre¬ 
hensive manual, particularly the v-h-f radio section, was 
related to the work done under Project C-79. The list of 
Signal Corps equipment was made into a separate manual, 
TM11-487, October 2, 1944. 


3 






4 


SYSTEMS ENGINEERING FOR AAF COMMUNICATIONS 


12 PROJECT ACCOMPLISHMENTS 

For tactical Air Warning Service, the li-f band was 
obviously overcrowded. Preliminary studies indicated 
that while both the 30- to 40-mc or 70- to 100-mc 
very-high-frequency (v-h-f) bands were suitable, com¬ 
mercial 30- to 40-mc equipment already available was 
the most readily adaptable. The principal contribu¬ 
tion was in providing the Signal Corps and the manu¬ 
facturer with information on means for minimizing 
interference effects. 

The Air Forces were faced with the situation in 
which a large number of point-to-point radio circuits, 
each on a different frequency, converged at one loca¬ 
tion, resulting in a heavy concentration of transmit¬ 
ters and receivers. It became apparent early in this 
study that with current designs of both commercial 
and military v-h-f sets in the portable category, spuri¬ 
ous transmitter radiations and receiver responses con¬ 
stituted a major interference problem when operating 
a number of sets in close proximity. The work dealt 
with the sources of these spurious effects, their mag¬ 
nitudes, set design factors, and practical aids such 
as the separation between sets and arrangements of 
antennas. The final step was the tedious process of 
determining workable groups of frequency assign¬ 
ments. This work provided a procedure for treatment 
of similar problems by both the Army and Navy. 

In the military application of v-h-f radio sets, it 
is important to be able to predict whether or not a 
radio system can be expected to operate satisfactorily 
over specific paths. This problem was investigated, 
particularly for situations where the radio path in¬ 
cludes intervening hills and so is not line-of-sight. 
Information for making such predictions was prepared 
in a form that does not require extensive technical 
knowledge for its understanding and is therefore very 
useful in the field. In this work, simple principles for 
the selection of good antenna sites were also stated. 

In the latter phases of the study many of the same 
problems outlined above for the 30- to 40-mc band, 
were also studied for the 70- to 100-mc band for 
which equipment became available. Study of this 
equipment indicated that its characteristics with re¬ 


gard to mutual interference were comparable to those 
encountered in the lower frequency band. Further¬ 
more, it did not possess certain operating and packag¬ 
ing features needed for the Air Forces’ applications. 
A single-channel set more nearly meeting the require¬ 
ments was designed. 

Remote control arrangements were designed for 
both the 30- to 40-mc and 70- to 100-mc equipment 
discussed above. 

The use of v-li-f, although well suited to the Euro¬ 
pean theater, was not well adapted to situations fre¬ 
quently met in the Southwest Pacific where it was 
necessary to transmit through jungle territory or be¬ 
tween islands beyond v-h-f range. An investigation 
was made which led to a recommendation that, for 
such cases, reliance be placed on transmission by sky 
waves in the 2- to 8-mc band, for which antenna ar¬ 
rangements which radiate well at high vertical angles 
were devised. Estimates were prepared of the field 
intensities necessary to override atmospheric static in 
various parts of the world. 

A single-channel teletypewriter system was developed 
which permits changing a radio circuit back and forth 
at will between teletypewriter and voice operation, 
and which can be used on tactical radio sets working 
on a push-to-talk basis. Also included in the telegraph 
studies was the consideration of c-w Morse operation 
at v-h-f under conditions where speech is jammed. 

Antennas for use with the various sets were con¬ 
sidered. This included selection of suitable types, cali¬ 
brations, and packaging arrangements. A somewhat 
related activity was the development of a lightweight 
50-ft ply wood antenna mast which can be erected by two 
men. This mast was procured in substantial numbers. 

Another investigation was that of laying field wire 
from airplanes. This study, covering one method of 
paying out the wire, was carried on concurrently with 
work which the Bell Laboratories were doing under 
direct contract with the Army Air Forces covering 
other methods. Promising results were obtained from 
both these studies and the work was continued under 
a new contract between the Army Air Forces and the 
Bell Laboratories. The work under this contract, Proj¬ 
ect C-72, is summarized in Chapter 30. 





Chapter 2 

PROPAGATION STUDIES 


21 EFFECT OF HILLS AND TREES ON 
RADIO PROPAGATION 

211 Introduction 

U nder project C-79 a general study of the com¬ 
munications system of the air and ground forces 
was undertaken. Since the effect of obstructions as well 
as terrain characteristics is important from the stand¬ 
point of proper selection of radio-terminal sites, the 
investigations under Project 13.2-83 a were essential 
as part of an overall study. 

The final report 1 is divided into two major sections, 
Part I dealing with the transmitters, power supplies, 
methods of monitoring, field-intensity measuring 
equipment, antenna systems, calibration of the meas¬ 
uring equipment and radiation measurements. Part 
II is divided into sections dealing with the effect of 
trees, mountains, elevation of antenna system, and 
effect of receiver and transmitter locations. 


212 Results of the Investigation 

The woods used to study the effect of trees on radio 
propagation covered a sufficient area, and were such 
that the results should be valid for the great majority 
of cases in a region similar to that east of the 
Allegheny Mountains. The possible exception where 
these results would not apply would be a dense forest 
of tall trees with an abundant undergrowth, such as 
is found in some swampy areas and in tropical jungles. 

213 Horizontal versus Vertical Polarization 

Measurements at 28 and 116 me with antenna 
heights of 19 to 29 ft demonstrated the general supe¬ 
riority of horizontally polarized waves in relatively 
broken country, both as to magnitude of the field in¬ 
tensity and uniformity in the presence of nearby in¬ 
terfering objects such as trees and houses. In the case 


“Project 13.2-83, Contract OEMsr-1010, Jansky & Bailey. 


of 116 me, this is also true even in open country free 
from any outstanding obstruction. 

A comparison of the results with theoretical com¬ 
putations for propagation over a smooth spherical 
earth shows that the horizontally polarized waves are 
not only attenuated less but also are in much closer 
agreement with theory for the antenna elevations used 
in the study. This is particularly true at 116 me, where 
the actual field intensity for vertical polarization was 
11 db lower than calculated while for horizontal polar¬ 
ization it was only 4 db lower. In rolling country 
vertically polarized waves were approximately 7 db less 
than horizontally polarized waves. 

For horizontal polarization, trees of the type en¬ 
countered would cause a loss in transmission of only 
about 10 per cent. For vertical polarization at 4 me 
the loss is approximately proportional to the heights 
of the trees; in woods with 25-ft trees, approximately 
two and one-half times the power would be required, 
and with 50-ft trees ten times the power would be re¬ 
quired to maintain communication at the same maxi¬ 
mum range, as in open country. At 28 me, the trans¬ 
mission loss due to trees was not serious. A lateral 
movement of antenna, a 5-ft increase in antenna 
height or a 100 per cent increase in power would 
overcome the effect of the trees. The signals were ap¬ 
proximately normal if both antennas were above the 
foliage. 

At 116 me, vertical polarization is unsuitable in 
woods with 25-ft trees unless a position of minimum 
signal can be avoided. With 50-ft trees it would be 
necessary to operate from a position of maximum 
signal. A change in location of 2 ft would often cause 
the signal to change from maximum to minimum. 
On windy days the signal was often nullified even if 
a favorable location was employed. 


214 Effect of Mountains and Hills 

If a communication link is capable of maintaining 
communication over an unobstructed path for a dis¬ 
tance so great that the field intensity of a wave propa¬ 
gated over an obstructing ridge is not less than the 
normal value, then the service range of the link will 


5 





6 


PROPAGATION STUDIES 


not be affected or impaired by the presence of the 
mountain. If this condition is not satisfied, a reduc¬ 
tion in range will be experienced which at times may 
be very pronounced. 

Furthermore, when extreme range is desired and a 
position on a ridge is not accessible, if the minimum 
distance requirement has been exceeded a skillful 
operator may pick an advantageous position behind a 
ridge and exceed the maximum distance obtainable 
from a position that avoids the ridge. This apparently 
anomalous behavior results from the signal gain due 
to constructive interference resulting from the in¬ 
creased number of ray paths arriving at the receiv¬ 
ing point. 

215 Effect of Antenna Height 

Increasing the height of both transmitting and re¬ 
ceiving antennas is equivalent to increasing trans¬ 
mitter power, the governing factor being the propor¬ 
tional change in antenna heights , unless the absolute 
transmitting heights are specified. For example, an 
increase in antenna heights from 19 ft to 29 ft is 
equivalent to a transmitting power increase of five 
times. Naturally, the same equivalent increase in 
transmitter power would not be realized if the antenna 
heights were increased 10 feet from a 100-ft elevation. 

At 28 me this gain is quite dependable under all 
circumstances. At 116 me, under ideal conditions the 
results are consistent; in rolling terrain there is a 
positive gain but less than that experienced in level 
country; in woods in level country the results are 
erratic but on the average about what is found in level 
unobstructed country. Where the path of transmission 
crosses a mountain the gain is random and may be 
negative. If a fixed antenna height is to be used in 
mountains the average results will be as effective 
with antenna heights of 15 to 19 ft as they will be 
at greater heights. For maximum results a variable- 
height antenna is necessary, the gain being as great 
as 5 to 10 db above average and as much as 20 db 
above the minimum values with fixed-height antennas. 

216 Choice of Transmitter and Receiver 

Locations 

In wooded areas and over short distances (4 miles) 
there is little advantage in choosing a clearing rather 
than a location among the trees provided the foliage 


is at least 10 ft from the antenna. At greater distances 
a location in a clearing will give some advantage and 
a distance of 50 ft from trees 25 ft high is desirable. 

The best location under all conditions is the crest 
of a mountain. In rolling country where a single pre¬ 
dominant ridge is not available, the crest of a knoll 
is desirable. In some cases a site in front of the actual 
crest is better than on the crest, but the rear side 
should always be avoided. On a small level plateau 
there is little advantage in locating the site on the 
forward edge of the level area and no special effort is 
warranted in reaching it unless the situation is 
extreme. 

2.2 RADIOTELEPHONE COMMUNICATION 
BETWEEN MOBILE UNITS 

221 Introduction 

The original purpose of Project C-30 a was to de¬ 
termine the most desirable frequencies for radio¬ 
telephone communication between mobile vehicles over 
distances up to ten miles as well as to study the an¬ 
tennas in use on such vehicles and noise levels en¬ 
countered by receivers under such conditions. The 
events of December 1941, however, required immediate 
military decisions, together with a tremendously ac¬ 
celerated procurement program and tended to shift 
the emphasis in the project away from the original 
purpose and toward the more practical problem of 
how best to use the available equipment and how to 
measure its performance. 

A considerable portion of the final report 2 on this 
project is devoted to descriptions of equipment em¬ 
ployed in the studies and methods of calibration uti¬ 
lized. In 1942 the Armed Forces were already using 
frequencies of 4 and 28 me, approximately, for mobile 
ground communication, whereas the 116-mc frequency 
was mainly employed between plane and ground. 

Throughout the project and in the final report 2 at¬ 
tempts were made to correlate the measurements and 
propagation data with theoretical curves because, by 
the use of such curves, the performance of radio sys¬ 
tems over territory where studies have not been made 
can be predicted. Graphical methods of solving the 
complicated equations for ground-wave propagation 
over the surface of the earth were worked out and 
reported. 


“Project C-30, Contract OEMsr-174, Jansky & Bailey. 






STUDIES ON 116 MEGACYCLES 


7 


2 2 2 Work Accomplished 

Following construction, procurement, and calibra¬ 
tion of necessary transmitting and measuring equip¬ 
ment, field strength measurements on 4 and 28 me 
were made in three areas, one in the Beltsville, Mary¬ 
land, area characterized by rolling terrain, one in 
Bridgeville, Delaware, characterized as flat, and one 
near Twiggtown, Maryland, which was mountainous. 

After these field intensity studies had been made, a 
means for measuring the free-space field from mobile 
antennas was worked out so that a study of the radiat¬ 
ing properties of mobile Army antennas could be car¬ 
ried on with the object of comparing certain installa¬ 
tions. The free-space field was useful as a figure of 
merit since it eliminated the effects of the ground and 
focused attention on the physical properties of the 
antennas. 

The next step in the development of the project was 
a comprehensive study of the antenna characteristics 
of radio installations in Army vehicles including old- 
and new-style command cars, the M-3 light tank, and a 
lMj-ton panel truck with and without trailer. The effect 
of having the whip antennas vertical or in the semi¬ 
horizontal running position was determined, showing 
that the radiating properties fell off badly with the 
antenna in the running position. 

A study of ignition and atmospheric disturbances in 
mobile Army communications systems followed. 


223 Results on 4 and 28 Megacycles 

As a result of the studies made, it was recommended 
that transmitting installations be rated in terms of 
“twice the free-space field” produced at unit distance 
or the equivalent “unattenuated field at one mile.” A 
method for doing so was worked out and is described 
in the final report. 2 

At 28 me, for the receivers tested, it was found that 
input terminal voltages varied from 1.1 to 2.2 /av for 
each microvolt per meter of field intensity in which 
the antenna was immersed. The tests indicated that 
for vehicles in good condition, well isolated from 
sources of man-made noise, field intensities of the 


order of 1 /x\’ per meter were entirely satisfactory for 
intelligible f-m communication; that in convoy ser¬ 
vice in the presence of vehicles with less satisfactory 
ignition-noise suppression, 4 to 6 fiv per meter were 
necessary for an equivalent class of service. It was felt 
that situations in which more than 6 /av per meter 
would be required would be infrequent and could be 
eliminated by corrective measures on the vehicles or 
by a change in location. It was felt, furthermore, that 
atmospheric noise was not a factor in limiting com¬ 
munication over f-m circuits at 28 me and could, 
therefore, be ignored. 

When all factors were taken into account, the con¬ 
clusion was reached that f-m signals in the 20- to 40- 
mc range were more consistent and reliable for ranges 
of 3 to 12 miles in level and rolling terrain than a-m 
transmission of equivalent power at 4 me because they 
were not limited by atmospherics nor were they so sus¬ 
ceptible to variations of ground conductivity. In 
mountainous terrain, the higher-frequency signals suf¬ 
fered in comparison with lower-frequency signals be¬ 
cause of deep shadows and general reduction in the in¬ 
tensity of average signals. 

Atmospheric noise was definitely the limiting factor 
in 4-mc communication. 

2.3 STUDIES ON 116 MEGACYCLES 

In general the work on 116 me followed in outline 
that on 4 and 28 me. Field studies were made in the 
rolling country near Beltsville, Maryland. As a result 
of these field measurements it was reported that re¬ 
liable communication could be carried on over a dis¬ 
tance of about 9 miles with a transmitting antenna 
not over 6 ft above ground, and with antenna power 
in the neighborhood of 25 watts, assuming that a 1-ju.v 
per meter received signal would give good communi¬ 
cation. 

Use of 116 me was indicated only when the re¬ 
duced size of the antenna structure or the secrecy 
secured by lack of sky wave would have tactical ad¬ 
vantages and then only in gently rolling or reasonably 
flat terrain. Mountainous territory would affect the 
116-mc signals to an extent even greater than that 
on 28 me. 


CONFIDENTIAL 





Chapter 3 

FREQUENCY MODULATION VERSUS AMPLITUDE MODULATION 


A comparison of the properties of a-m and f-m communica¬ 
tion at very high frequencies, particularly for airborne use. 
Theoretical study of the maximum possible range of commu¬ 
nication and of the properties of the two systems as to sup¬ 
pression of noise such as static and electrical disturbance aris¬ 
ing from other equipment in the airplane, c-w jamming, etc. 

3 1 PROPERTIES OF AMPLITUDE 
MODULATION SYSTEMS 

I n an a-m system, the final report 1 on Project 13- 
110 a shows that, in the presence of random noise, 
the narrower the i-f pass band the better the perform¬ 
ance, but that the impairment in performance due to 
widening the pass band is not great. For strong sig¬ 
nals there is relatively little difference in the perform¬ 
ance of a wide-band and a narrow pass band system. 
If the carrier strength is such that the carrier-to-noise 
ratio is unity, then the wide-band system has a ratio 
of signal to audible noise which is poorer by a factor 
of 2. Articulation tests made by the Psycho-Acoustic 
Laboratory, Harvard University, bear out this ana¬ 
lytical criterion. 

Random noise cannot be balanced out, for example, 
by using random noise from one channel to oppose and 
supposedly cancel the random noise in another chan¬ 
nel. Against a man-made noise which is not random, 
balancing schemes may be of value. 

If interference takes the form of short sharp pulses, 
their effect on intelligibility can be kept to a mini¬ 
mum, especially when they are limited or clipped in 
the receiver so that their maximum amplitude does 
not exceed the amplitude of the voice wave. Analysis 
shows that to keep the i-f transient pulse as narrow as 
possible, it is necessary that the i-f response curve be 
as broad as possible. Since, however, a flat-topped rec¬ 
tangular response curve is not realizable, the effect of 
the slope of the response curve is important. In fact, 
the slope of the sides of the i-f response curve is more 
important than the width in minimizing the width of 
the i-f transient. 

If provision has been made for keeping the width 
of the i-f transient pulse as narrow as possible, the 
next step is to minimize its effect on the audio output. 
The final report 1 discusses clippers, counter-modu- 

•Project 13-110, Problem No. 2, Contract OEMsr-1441, Har¬ 
vard University. 


lators, and balancers as being useful in reducing the 
effect on the audio output. 

The analysis concludes that a good noise limiter 
and provision for minimizing the desensitization of 
the receiver due to a-v-c voltage developed by the 
pulses are essentials for a good communications re¬ 
ceiver. The performance of an a-m receiver is not 
greatly affected by mistiming, provided the carrier 
and side bands do not come too close to the edges of 
the pass band of the receiver. 

32 PROPERTIES OF FREQUENCY 
MODULATION SYSTEMS 

Frequency modulation has certain noise-suppressing 
properties which are inherent. In the presence of ran¬ 
dom noise only, when the ratio of carrier to noise is 
large, the f-m system yields' a signal-to-audible-noise 
ratio which is better than that of an a-m system by 
the expression 

V3 X i-f band width 
B X a-f maximum frequency * 

where B is a factor depending upon the deviation 
ratio. For a deviation ratio of 5, B is 1.6; for a de¬ 
viation ratio of 15, B is 1.3. 

Wide-band frequency modulation is capable of pro¬ 
ducing an extremely noise-free audio output at the 
price of a high r-f signal strength. Narrow-band fre¬ 
quency modulation cannot attain so much freedom 
from noise but yields a usable output at a lower r-f 
signal strength and hence is better adapted to those 
applications where distance of communication is more 
important than perfection in the audio output. 

The superiority of f-m over a-m systems is due to 
the fact that the noise voltage spectrum of the f-m 
system is triangular with zero value at zero frequency, 
.increasing linearly with frequency. The result is that 
most of the noise output is of such high frequency that 
it is inaudible. 

When the carrier amplitude is greater than the 
noise amplitude, the frequency deviation of the sum of 
the carrier and noise is the same as that of the carrier; 
but when the noise exceeds the carrier, the frequency 
deviation is the same as that of the noise alone. Thus 


CONFIDENTIAL 


8 






GENERAL CONCLUSIONS 


9 


noise effectively decreases the average frequency de¬ 
viation and in this manner decreases the audio output. 
Analysis shows that, for weak carriers, noise suppress¬ 
es the signal less when no limiter is employed. Thus 
there is an optimum level at which the limiter should 
operate. 

So far as pulse noise is concerned, an f-m receiver 
equipped with a balanced discriminator is, under cer¬ 
tain conditions, capable of complete suppression of 
pulses. Thus, there is no output when no carrier is 
present; if an unmodulated carrier exists at the center 
of the i-f band, no output is caused by a pulse at either 
peak of the carrier wave. These conclusions are true 
whether or not a limiter is used. 

A pulse occurring at a position other than the peaks 
of the carrier tuned to the center of the i-f band pro¬ 
duces noise from a balanced discriminator. Under this 
condition the noise suppression depends upon the 
limiter action and not upon the balance of the dis¬ 
criminator. 

When the carrier is detuned from the center of the 
i-f band and there is no limiter, a pulse produces some 
output regardless of its position with respect to the 
carrier cycle. When the carrier is detuned in the pres¬ 
ence of random noise, the a-f noise increases. Over- 
modulation, therefore, will increase noise, since the 
excessive frequency deviation producing the over¬ 
modulation is equivalent to excessive detuning. 

33 COMPARISON OF F-M AND A-M 
SYSTEMS 

In the presence of random noise only, it is certain 
that when the carrier is large compared with the noise, 
the f-m receiver is quieter in output. The minimum 
rms carrier strength at which this improvement is ob¬ 
tained is termed the threshold level and is about twice 
the rms value of the random noise. This threshold level 
is lower for a narrow-band system than for awide-band 
system, and at carrier levels below the threshold of the 
narrow-band system the audio output of the narrow- 
band receiver is quieter than that of a wide-band set. 
For maximum range, therefore, the narrow-band sys¬ 
tem is preferred. 

At carrier levels below the threshold, it seems that 
the f-m receiver is superior to an a-m receiver. Under 
this project, the receivers of SCR-508 f-m transmitter- 
receiver equipment were compared when the BC-603 
receivers were employed first as an f-m receiver and 
then as an a-m receiver. The f-m performance was 


consistently better. At an r-f input of 1 fiv, the ratio 
of signal-plus-noise to noise was 13 db for amplitude 
modulation and 37 db for frequency modulation. 

Against pulse interference, the a-m receiver has no 
protection except that which may be added in the form 
of clippers or limiters. An f-m receiver with an i-f 
limiter is vastly superior in this respect. 

With respect to jamming, there seems to be no 
marked superiority of amplitude or frequency modu¬ 
lation. Both can be jammed. So long as the desired 
signal is stronger than the jamming signal and docs 
not take over the limiter, the signal from an f-m re¬ 
ceiver is clean, but beyond this point interference rises 
rapidly. Both f-m and a-m sets are thoroughly jammed 
when the interfering signal is from 6 to 10 db greater 
than the desired signal. Amplitude-modulation receiv¬ 
ers are less vulnerable to c-w jamming when the beat 
frequency is inaudible. Such interference can take over 
the limiter in the f-m set, but will take over the a-m 
set only when it is strong enough to desensitize the 
receiver through a-v-c action. 

In the presence of pulse interference, it appears that 
the f-m receiver is quieter even if the a-m receiver has 
the best limiter possible. 

Communication by f-m apparatus is more suscep¬ 
tible to multipath transmission when the difference in 
the time of transmission is of the order of the modu¬ 
lation cycle. When this time difference is of the order 
of one r-f cycle the effect is to alter the intensity of 
the modulated carrier identically, in the two systems 
and not to distort the signal. 

In the presence of a carrier in the f-m system, ran¬ 
dom a-f noise is reduced considerably and for this 
reason it is possible to balance a carrier voltage against 
a noise voltage so that a very weak carrier can open the 
“squelch” but noise cannot. On the other hand, in the 
absence of carrier a weak noise voltage can open the 
squelch. The balancing action is not possible in an 
a-m receiver. 

With regard to size and weight there is definite ad¬ 
vantage in frequency modulation. 

3.4 GENERAL CONCLUSIONS 

Narrow-band frequency modulation is preferable to 
either a-m systems or to wide-band systems for general 
communication purposes, especially if crystal control 
of the receiver is possible so that the r-f band width 
can be narrow. Frequency modulation requires some¬ 
what closer tuning but since the improvement against 


CONFIDENTIAL 







10 


FREQUENCY MODULATION VERSUS AMPLITUDE MODULATION 


noise is so great, precise tuning is not necessary. The 
capture effect prevents simultaneous reception of two 
signals and renders the f-m receiver somewhat less 
susceptible to jamming. 

Noise limiters are necessary in a-m receivers. In an 
f-m receiver, on the other hand, the discriminator 
inherently cuts down noise and limiters may be em¬ 


ployed in both the r-f and a-f circuits. At the time of 
this report, it had not been necessary to use limiters 
in the a-f end of the receiver. 

An extensive bibliography is part of the final re¬ 
port , 1 which also contains numerous oscillographs, 
etc., giving visual comparison of the two systems of 
modulation. 


f 


ONFIDENTIAL 





Chapter 4 

MICROWAVE COMMUNICATION SYSTEMS 


A brief survey which evaluates microwave communication 
systems® in use, under development, or being considered in the 
United States prior to August 30, 1945. Pertinent facts con¬ 
cerning the established and contemplated systems for com¬ 
munication at carrier frequencies above 1,000 me including 
a cross-band microwave system. 

41 GENERAL CONSIDERATIONS 

T he usefulness of a given type of communication 
equipment is governed by a number of practical 
considerations, such as: 

1. The type of communication service it supplies. 

2. The economy with which it uses band space. 

3. The intelligibility of reproduction and the noise 
level at the reproducer. 

4. Its maximum reliable range. 

5. Its ease of operation. 

6. Its size, weight, cost. 

7. Its ruggedness and ease of maintenance and re¬ 
pair. 

These practical aspects depend upon a number of tech¬ 
nical considerations which will be reviewed in turn. 

411 Types of Operations 

A microwave system may embody one or more of the 
following types of operation: 

General call (every station in a network receives a 
call addressed to one of them). 

Selective call (only the called station receives a 
call). 

Unidirectional operation (flow of intelligence in 
only one direction in a given communication circuit). 

Duplex operation (flow of intelligence in both di¬ 
rections simultaneously in a given communication 
circuit). 

Relay operation (instantaneous retransmission of 
an incoming message). 

Simplex operation (only one wave of intelligence 
on a given r-f carrier). 

Multiplex operation (several waves of intelligence 
on a single r-f carrier). 


•Project 13-110, Problem 9, Contract No. OEMsr-1440, 
Harvard University. 


412 Types of Emission 

The type of emission is an important consideration 
in microwave transmission. The emission may have 
any one of a number of forms, such as: 

Continuous wave, amplitude modulated [a-m]. 

Continuous wave, frequency modulated [f-m]. 

Continuous wave, phase modulated [p-m]. 

Pulsed wave, pulse-amplitude modulated [p-a-m], 
in which equally spaced short pulses undergo varia¬ 
tions in amplitude corresponding to the instantaneous 
value of the modulating signal voltage. 

Pulsed ivave, pulse-length modulated [p-l-m], in 
which pulses of constant amplitude are varied in 
length (i.e., duration) in accordance with the instan¬ 
taneous value of the modulating signal voltage. 

Pulsed wave, pulse-number modulated [p-n-m], in 
which short pulses of equal amplitude and equal spac¬ 
ing are transmitted. Signaling is accomplished by 
emitting a varying number of these pulses in accord¬ 
ance with the instantaneous value of the modulating 
signal voltage. 

Pulsed wave, pulse-frequency modulated [p-f-m], in 
which short pulses of equal amplitude are transmitted 
at a variable rate corresponding to the instantaneous 
value of the modulating signal voltage. 

Pulsed wave, pulse-position (phase) modulated 
[p-p-m], sometimes referred to as pulse-time modu¬ 
lated [p-t-mj. Pulses of constant amplitude and 
length are advanced and retarded in time from a uni¬ 
form repetition rate by amounts of time which are 
proportional to the instantaneous value of the modu¬ 
lating signal voltage. The displacements of the pulses 
are measured either in time or in terms of the dis¬ 
placement of the r-f wave in radians. 

413 Spectra and Spectrum Widths 

The spectrum width W of an a-m signal is twice as 
great as the highest frequency of the modulating sig¬ 
nal f m if the modulation does not exceed 100 per cent. 
Thus, 

IF = 2/ (for a-m). (1) 

The width of the significant spectrum of an f-m 


■ 


CONFIDENTIAL 


11 








12 


MICROWAVE COMMUNICATION SYSTEMS 


signal is, to a first approximation, slightly more than 
twice the maximum frequency deviation A/ (the 
maximum displacement of the frequency when voice 
signal is loudest). Thus, 

TV == 2A/ (forf-m). (2) 

A reasonable approximation for the spectrum of a 
pulsed wave can be determined from the pulse dura¬ 
tion a, without regard to the modulation. Assuming 
the pulses to be rectangular, the width TV of the cen¬ 
tral portion of this spectrum is 

TV (for pulsed emission). (3) 

a 

In a-m transmission, for /„ = 3 kc, TV — 6 kc. 

In f-m transmission in the microwave region the 
most suitable value for A/, and therefore the value of 
TV, will be influenced by a number of factors. The most 
suitable value had not yet been determined at the time 
of this report. Values of A/ of the order of 100 kc are 
contemplated. 

The values of a employed in the existing pulsed- 
emission equipments result in calculated central spec¬ 
trum widths TV of the order of 3 me, with a tail on 
each side of the central part having appreciable height 
over a range of the order of 3 me. The actual spectra 
are apt to be broader and perhaps unsymmetrical as a 
result of undesired reflections due to imperfect impe¬ 
dance matches and from undesired alteration of the 
carrier frequency due to the pulsing of the transmit¬ 
ter. It appears that simplex pulse transmission could 
be characterized by a considerably narrower spectrum 
than that just described, although the spectrum would 
still be many times broader than that of a-m or f-m 
transmission. 

This may be reasoned as follows. In simplex pulse 
transmission it is not suitable for the pulse repetition 
rate f p to be less than two or three times f m , which sets 
a lower limit on f p . For example, let us assume a mini¬ 
mum value of f p of 8 kc. A duty factor must next be 
assumed. A large duty factor tends to narrow the spec¬ 
trum and at the same time to lessen any increase in 
transmission range resulting from the use of pulses, 
and vice versa. If a duty factor of 5 per cent is as¬ 
sumed, the pulse length a is 6.25 /i,sec, and, by equa¬ 
tion (3), TV = 0.32 me. 

41,4 Multiplexing of Signals 

The various schemes for multiplexing of signals on 
a single r-f carrier may be divided into two general 


groups: frequency-division multiplexing, and time- 
division multiplexing. 

In frequency-division multiplexing, several intelli¬ 
gence signals are combined into one signal which is 
applied to the transmitter as a single modulating 
wave. Usually the frequencies of the signals to be com¬ 
bined are elevated to different degrees and are then 
added together. Two examples follow: 

“Spiral-four” audio-frequency multiplexing. Four 
voice-frequency signals, each ranging from 200 to 
2,800 cycles per second, are combined into a modulat¬ 
ing signal having a frequency range from 200 to 11,- 
600 cycles per second. Channel 1 contains an ordinary 
voice spectrum. Channels 2, 3, and 4 contain single 
side-band (lower side-band) suppressed-carrier signals. 

Supersonic-frequency multiplexing. In this scheme 
the modulated subcarrier frequencies are supersonic 
frequencies; for example, 30, 40, and 50 kc. 

Time-division multiplexing is best described by 
means of an example. In the AN/TKC-5 equipment, 
the emission consists of a 2.0-fisec marker pulse fol¬ 
lowed by eight 0.4-^u.sec channel pulses which, in the 
absence of modulation, are equally spaced in time. The 
complete cycle of nine pulses requires 100 /xsec. Once 
each 100 fisec, each of eight voice-frequency circuits 
is “sampled,” and the channel pulse assigned to it is 
position-modulated. 

Frequency-division multiplexing has been used only 
in connection with continuous-wave emission, al¬ 
though it is applicable to all types of emission. Time- 
division multiplexing is especially suited to pulsed 
transmission. 

In general, multiplexing provides “trunk-line” ser¬ 
vice and some degree of privacy. It results in little if 
any economy of band space, other factors being equal. 
Calculations 1 indicate that to retain a fixed value of 
signal-to-noise ratio, the theoretical output spectrum 
must be widened directly as the number of signals 
combined by multiplexing. In the case of frequency- 
division multiplexed a-m transmission consisting of 
one carrier and an upper and lower side band for each 
voice channel, carriers are eliminated but guard space 
must be provided between the frequency ranges allo¬ 
cated to the various voice channels. In f-m transmis¬ 
sion, calculations show that the value of A/ and hence 
that of TV must be increased in the same proportion 
as f m to maintain a fixed signal-to-noise ratio. In time- 
division multiplexing of pulsed transmission, the min¬ 
imum number of pulses required for simplex trans¬ 
mission must be multiplied slightly more than N 


\ CONFIDENTIAL 





GENERAL CONSIDERATIONS 


13 


times if N voice channels are to be accommodated. 
Hence, multiplexing of pulsed transmission requires 
a decrease in a and an increase in W. 

Crosstalk Problem 

An inherent difficulty in connection with frequency- 
division multiplexed transmission is crosstalk in the 
form of modulation products resulting from nonline¬ 
arity in the amplitude characteristics of the equip¬ 
ment. In a-m transmission this is due to nonlinearity 
in the tube characteristics, and in f-m transmission to 
nonlinearity in the phase response of the circuits. In 
time-division multiplexed transmission under proper 
conditions only one voice signal is handled at a time, 
so that crosstalk ideally is impossible. However, dis¬ 
tortion due to tube or circuit characteristics is not 
impossible in pulsed transmission, for it may occur 
in the tubes and circuits of the modulating portion 
of the transmitter. 


41 - 6 Use of Band Space 

While in theory the number of channels possible in 
a given amount of band space depends solely upon the 
type of emission and the desired signal-to-noise ratio, 
in practice the number of possible channels depends 
also upon other practical factors. Perhaps the most 
important are the accuracy with which the transmitter 
carrier frequency can be reset and stabilized, and the 
stability of the local oscillators in the receiver. The 
following data are based upon the premise that all 
such practical difficulties are surmountable. 

Table 1 represents an attempt to determine approxi¬ 
mately the relative ultimate possibilities of the use of 
band space in simplex communication. Guard bands 
of estimated minimum practicable size have been 
allowed between the spectra. These values represent 
combinations of calculations and estimates, which is 
as far as one can go with safety at the present stages 
of both theory and practice. 


415 Establishment of Carrier Frequency 

In microwave communication, the carrier frequency 
must be readily and precisely selectable and extremely 
stable. A precision of the order of one part in 10 (i or 
better is required to realize fully the possibilities in 
the microwave region predicted by theory. 

Crystal control yields the required stabilization, but 
a huge number of crystals would be required for flex¬ 
ible operation of a system consisting of hundreds of 
stations and hundreds of frequency channels if each 
channel is to be available to every station. Crystal sav¬ 
ing has been attempted in connection with several 
v-h-f systems; for example, a few crystals may be used 
to provide a large number of frequencies through the 
processes of beating and multiplication. Another pos¬ 
sibility is to mix a crystal-stabilized signal with the 
output of a tunable oscillator and use the sum or dif¬ 
ference frequency as the carrier frequency. In this 
system all but a very small part of the carrier fre¬ 
quency is crystal-controlled. 

If crystals are used, there still remains the problem 
of selection of the correct frequency multiple. At the 
high level of multiplication required, this calls for 
considerable equipment. Also, if the r-f output power 
tube is not a power amplifier, crystal stabilization may 
be had only in the form of an automatic-frequency- 
control (a-f-c) circuit, which decreases the stability 
obtained. 


Table 1. Estimates of the minimum carrier-frequency spacing 
for simplex transmission in the microwave region, assuming 
the absence of all practical difficulties. 


Type of 
emission 

Estimated 
minimum space 

Description of transmission between carriers 

AM 

Carrier and two side bands; 

f m = 3 kc; 4-kc guard space allowed 



between spectra. 

10 kc 

FM 

Carrier and two side bands; 

/,„ = 3 kc; A/ = 3 kc; 4-kc guard 



space between spectra. 

10 kc 

Pulsed 

fm = 3 kc;/p = 8 kc; duty 
factor = 5%; a = 6.25 /xsec.; 

IF = 0.32 me; 0.18-mc guard space 



allowed between spectra. 

0.5 me 


The data in Table 2 show the estimated minimum 
amounts of band space required per voice channel in 
various forms of multiplex transmission, again assum¬ 
ing the absence of all practical difficulties. 

It is emphasized that the values shown in the two 
tables must be regarded only as estimated limits. Also, 
the values do not mean too much on a relative basis 
because the amounts of difficulty encountered in trying 
to achieve these limits will not necessarily be in the 
same ratios. 

It is possible theoretically to operate several pulsed 
transmitters on the same carrier frequency simultan¬ 
eously without interference by using different pulse 
repetition rates. However, this provides little if any 


CONFIDENTIAL 









14 


MICROWAVE COMMUNICATION SYSTEMS 


possibility of increase in the number of voice channels 
in a given amount of band space because the trans¬ 
mitter which operates at the highest repetition rate 
would call for a channel width comparable to the sum 
of the channel spaces that would be required if each 
transmitter were operated at the same minimum repe¬ 
tition rate with nonoverlapping spectra. 

The shift in carrier frequency caused by the dop- 
pler effect is investigated in an appendix of the final 
report. 7 The frequency shift is not likely to be great 
enough to affect the values in Tables 1 and 2, for it 
is not probable that the pass band of the receiver will 
be narrow enough for a noticeable effect on communi¬ 
cation to be produced even in the most extreme case, 
where a frequency shift of the order of 12 kc is noted. 


Table 2. Estimates of the use of band space in multi¬ 
plex transmission in the microwave region, assuming the 
absence of all practical difficulties. 




Required 
band space 

Type of 


per voice 

emission 

Description of transmission 

channel 

AM 

4 voice '.channels, frequency-division 
multiplexed by use of the spiral-four 
scheme; f m = 11.6 kc; 6.8-kc guard 
space allowed between r-f spectra; 
carriers 30 kc apart. 

7.5 kc 

FM 

4 voice channels, frequency-division 
multiplexed by use of the spiral-four 
scheme; fwi = 11.6 kc; A/ = 11.6 kc; 
6.8-kc guard space allowed between r-f 
spectra; carriers 30 kc apart. 

7.5 kc 

Pulsed 

8 voice channels, time-division multi¬ 
plexed; f m of each voice channel = 

3 kc; one marker pulse for each 8 chan¬ 
nel pulses gives f p = 9 X 8,000 or 
72 kc; average duty factor = 5%; 
a = 0.7 nsec, IF = 2.8 me; 1.2-mc 
guard space allowed between spectra; 
carriers 4.0 me apart. 

0.5 me 


41 ' Required R-F Power and Its 
Development 

A maximum range of 100 miles is generally consid¬ 
ered satisfactory for microwave communication. A dis¬ 
tance of 25 miles frequently is regarded as sufficient. 

The amount of r-f output power required is a sub¬ 
ject on which there is some theoretical disagreement, 
and is a matter that is governed largely by the kind 
of antenna permitted by the type of communication 
service desired. If a directional antenna may be em¬ 
ployed at the transmitter or the receiver or at both, 
an average r-f output power of the order of 1 or 2 
watts may be sufficient to achieve the desired range. 


Experience indicates that if omnidirectional antennas 
are employed at both stations, the average r-f output 
power should be of the order of 50 watts or more to 
achieve maximum range. 

At present the most practical types of power tube 
available for microwave communication are the kly¬ 
stron and the magnetron. The klystron is limited to 
an average output power of the order of 2 watts. High- 
power magnetrons intended for pulsed transmission 
have been well developed. Tunable magnetrons in¬ 
tended for c-w transmission, having an output power 
of the order of 1 kw or more, have recently been 
designed. Certain existing models of klystron and 
magnetron lend themselves readily to frequency 
modulation. 

418 Cross-Band Signaling 

The cross-band principle of signaling has been used 
in carrier systems and may be used in any of the radio 
communication systems to be described. 

Cross-band signaling will be described with the aid 
of Figure 1. Stations 1 arid 2 are alike in all respects, 



Figure 1 . Cross-band system of communication. 


except that the antenna system at either station may 
be directional or not, as required. Transmission and 
reception go on simultaneously at each station. If a 
beamed antenna is used for transmission, either the 
same antenna or an identical antenna aimed in the 
same direction is used for reception. The two stations 
are in contact when they are within range and when 
their carrier frequencies and F 2 differ by an 
amount equal to the intermediate frequency employed 
by both stations. At each station, a portion of the 
modulated output signal is applied to the mixer in¬ 
stead of a signal from a local oscillator for frequency 


CONFIDENTIAL 



























SUMMARY 


15 


conversion. The signaling system is operationally bi¬ 
lateral and offers the following features of operation: 

1. Establishment of contact becomes reasonably cer¬ 
tain to the calling party when he finds that he can hear 
himself talk. The presence of the other station’s car¬ 
rier makes his own voice audible, and this implies that 
his own signal is also being received by the other party. 
A request for verification of contact usually is unnec¬ 
essary. 

2. Duplex operation, i.e., two-way communication 
similar to an ordinary telephone conversation, is 
achieved. Push-to-talk operation is eliminated; either 
party can interrupt the other at will. 

3. The strength and quality of the voice signals 
heard bv the two parties are practically the same, so 
that each one usually knows how well he is being 
received without asking for a report. 

An outstanding virtue of cross-band signaling in the 
microwave region lies in the fact that it eliminates the 
necessity for having a separate first local oscillator for 
mixing in reception and the attendant necessity of 
stabilizing its frequency. x4n associated advantage is 
that if either party’s oscillator is slightly off frequency, 
this weakens the signal equally at both stations, and 
readjustment of the frequency of either oscillator im¬ 
proves or weakens the reception at both stations. 

419 Protection from Interference 
and Jamming 

Obvious measures that may be taken in the micro- 
wave region to forestall interference and jamming of 
communication service are (a) the use of high power, 
(b) the use of directive antennas, and (c) the use of 
widely separated frequencies for transmitting and re¬ 
ceiving. Other measures are (d) the use of very nar¬ 
row pulses and hence the widest practical spectrum, 
(e) adjustable signal-amplitude selection circuits 
(clippers and limiters) to aid in the elimination of 
pulsed interference, and (f) some form of manual or 
automatic volume control which can be used to pre¬ 
vent overloading of amplifiers by a c-w signal. 

4.2 SUMMARY 

421 Comparisons of Microwave 

Communication Equipment 

At the time of this report, there has been little 
production or use of microwave communication equip¬ 
ment. 


The few experimental mobile-ground, shipborne, or 
airborne microwave communication sets which have 
been built conform basically to the existing push-to- 
talk v-h-f equipment, and their characteristics are in 
such an indefinite state that critical comparisons are 
not timely. 

No broadly applicable model of microwave commu¬ 
nication equipment has been developed. The sets so 
far developed are relatively bulky equipment intended 
primarily for transportable point-to-point radio relay 
service and are not suitable for other applications. 

It appears that there would be use for a simple, 
small, lightweight, rugged, versatile, inexpensive 
microwave communication set that could be produced 
easily in large quantities. The delay in the develop¬ 
ment of such a broadly applicable set has not been 
occasioned by lack of need or desire, but because the 
development of radar equipment was given higher 
priority. The radar developments have led to the cre¬ 
ation of numerous microwave components which now 
could be reassembled into a system designed for com¬ 
munication. 

4 2 2 The Choice of the Type of Emission 

It appears that a set intended for broad application 
and for production and use in large quantities should 
employ c-w emission. Continuous-wave emission per¬ 
mits simplicity in equipment and achieves economy in 
band space. At this time, it is easier to produce an 
f-m signal than an a-m signal, but this condition may 
disappear. There are some indications that f-m may 
remain preferable to a-m. For example, f-m may offer 
more promise in regard to signal-to-noise ratio and 
antijamming properties. 

The use of pulsed emission for relay and trunk¬ 
line service appears to be satisfactory. The ultimate 
possibilities in multiplexed c-w transmission appar¬ 
ently have not been determined. It is possible that 
developments in equipment, for example, in the field 
of carrier-frequency stabilization, may make multi¬ 
plexed c-w transmission more practical than it appears 
to be at present. It would then be a competitor with 
multiplexed pulse transmission because of the con¬ 
comitant economy of band space and simplicity of 
equipment. 

Theory indicates that p-p-m is the best all-around 
choice among the five types of pulsed emission that 
were considered. Also, extensive laboratory tests made 
by the Federal Telephone & Radio Corp. [FT&R] and 
the Radio Corporation of America [RCA] indicate 





16 


MICROWAVE COMMUNICATION SYSTEMS 


that when maximum protective devices are used, the 
antijamming effectiveness of p-p-m and p-f-m ap¬ 
proach equality. According to an FT&R report, the 
p-p-m system has many advantages over the p-f-m 
system. They may be listed as follows. 

1. No change of average power during transmission. 

2. Greater degree of privacy. 

3. Possibilities of multiplexing, which is extremely 
difficult to obtain with p-f-m. 

4. Ease of center pulse-frequency stabilization. 

5. Ease of obtaining and applying blocking po¬ 
tentials. 

6. Further advanced state of development of the 
p-p-m system. 

4 - 2 - 3 Stabilization of Carrier Frequency 

In pulsed transmission, the stability and accuracy 
of the carrier frequency need not be so great as in c-w 
transmission because of the breadth of the spectrum. 
When the production of an extremely accurate and 
stable carrier frequency becomes a simple matter, this 
advantage of pulsed emission over c-w emission will 
disappear, and also the possibility of several thousand 
c-w channels in a given microwave frequency band 
will become a reality. 

While a crystal-control scheme may be applied to 
microwave communication equipment, there is much 
promise in cavity stabilization using the microwave 
frequency discriminator described by R. V. Pound. 6 
This a-f-c system is simple and straightforward, is 
not limited to steps of frequency, and will make avail¬ 
able at every station an unusually large number of 
stabilized carrier frequencies. The success of the micro- 
wave discriminator a-f-c system will depend largely 
upon the development of a tuned cavity whose reson¬ 


ant frequency may be very finely adjusted in a reset- 
able manner, and whose frequency is affected negli¬ 
gibly by changes in temperature. 

4 - 2 - 4 Use of Cross-Band Operation 

Cross-band operation requires simpler and less 
equipment. The first local oscillator of each receiver 
is eliminated, and the a-f-c system that stabilizes the 
transmitted carrier frequency also stabilizes the signal 
used for the first conversion in reception. 

Cross-band operation could be instituted in connec¬ 
tion with almost any one of the existing or contem¬ 
plated equipments. There would be no great advantage, 
however, in applying it to pulsed or to multiplexed 
point-to-point trunk-line systems. It would be particu¬ 
larly useful in connection with a communication sys¬ 
tem composed of hundreds or thousands of stations, 
any one of which is to be able to contact any other 
station. Cross-band operation is achieved readily now 
that r-f power-dividing waveguide sections such as the 
“magic T” have been developed. 

From the operator’s point of view, cross-band opera¬ 
tion simplifies and speeds up the establishment of 
contact. The called party does not have to touch or 
adjust anything when called, but can simply start 
talking. With any other existing or planned two-way 
radio system having the same individual-channel pri¬ 
vacy offered by the cross-band system, the called party 
has to make a frequency selection of some sort before 
he can start responding, and he has no indication that 
he is being heard by the calling party until the calling 
party acknowledges his response. The cross-band sys¬ 
tem would minimize interference between stations and 
would result in economical use of the available band 
space. 


CONFIDENTIAL 





PART II 


3,000-MC COMMUNICATION 














Chapter 5 

FIELD TESTS AND 3,000-MC EQUIPMENT 


A study of the suitability and possibilities of 3,000-mc com¬ 
munication over land and over sea water; development of an 
omnidirectional and a directional communications system. 


STATE OF THE ART 


C-42 2 had about the same communication performance 
as a 400-mw directional system.) 

5. Two-way voice communication was demonstrated 
by means of equipment used for the preliminary tests 
(Project C-24) over a distance of 14.5 miles. 


W hen this WORK was started in June 1941, there 
had been little if any mobile communication on 
microwaves. Point-to-point circuits had been operated 
on wavelengths in the centimeter region, in particular 
the 1,600-mc (18-cm) link across the English Chan¬ 
nel; microwaves had been extensively used in radar. 
The object of this research, a therefore, was to investi¬ 
gate the possibilities of the microwave region for 
mobile communication. 

5.2 INTRODUCTION 

As a preliminary to the development of actual 3,000- 
mc communication systems, a study was made of 
propagation characteristics and circuit requirements, 
with the following results P 

1. Vertically polarized waves were found best over 
salt water; over land there was little difference be¬ 
tween vertical and horizontal polarization. 

2. At these frequencies it was found that a substan¬ 
tially optical line-of-sight path was required, that 
signals were greatly absorbed by trees and houses, and 
that the signals disappeared very quickly beyond the 
optical horizon. 

3. Power gains of 100 to 200 are obtainable with 
antennas of relatively small dimensions, making it 
practicable to communicate over distances of 15 to 20 
miles with power of the order of 1 watt. This high 
gain results in high directivity, which makes unau¬ 
thorized interception difficult. 

4. Where high directivity cannot be used, the trans¬ 
mission power must be increased considerably. (A 30- 
watt omnidirectional system developed under Project 


“Projects C-24 and C-42, Contract Nos. OEMsr-32 and 
OEMsr-442, RCA Laboratories; Project C-2, Contract No. 
NDCrc-75, RCA Manufacturing Co., Inc.; Project C-10, 
Contract No. NDCrc-191, Westinghouse Electric & Manu¬ 
facturing Co. 


5-3 PROPAGATION OVER SALT WATER 

To study propagation at 3,000 me over salt water, a 
klystron transmitter with a power output of 2 watts 
and a 30-in. parabolic reflector with a power gain of 
23 db over a half-wave dipole were installed on land, 
and a receiver consisting of a crystal converter with a 
1221-Y local oscillator and a 30-mc i-f amplifier were 
installed on a 38-ft Coast Guard boat. The receiving 
antenna consisted of a small horn with a 1-ft square 
opening having a gain of 15 db over a half-wave 
dipole. 

In the first test the transmitter antenna was in¬ 
stalled on a hill south of Port Jefferson Harbor, 192 ft 
above sea level, and directed toward Bridgeport, Con¬ 
necticut. Difficulty was experienced in keeping the 
boat in the beam of the transmitter, as some of the 
runs were made after dark and because no markers 
were available for checking the position. In a second 
test, the antenna was moved to a bluff on Mt. Misery, 
Long Island, almost directly over the water and 125 
ft above it. The course ran east from there to Horton’s 
Point. Figure 1 shows the profile of this course; signal 



Figure 1. Profile, Mt. Misery Point and Long Island 
Sound based on 4/3 of earth’s radius. 


strengths using vertical polarization over the course 
are given in Figure 2. On this run voice communica¬ 
tion could have been carried out to grazing incidence 
at 24 miles. Keyed tone modulation could have been 
carried out somewhat further, but no signal was heard 
at a distance of 34 miles. 


1 


CONFIDENTIAL 




19 









20 


FIELD TESTS AND 3.000-MC EQUIPMENT 



Figure 2. Signal strengths with vertical polarization 
over sea water‘transmitter at Mt. Misery, Long Island, 
course east to Horton’s Point. 

A run using horizontal polarization was made over 
approximately the same course (to Mattituck, Long 
Island), the resulting measurements (Figure 3) indi¬ 
cating the maximum range to be about the same as 
for vertical polarization. 

These tests of the two types of polarization showed 
that with horizontal polarization the minima were 
very deep and short, the signal being inaudible at 
these points indicating almost complete cancellation 
between the direct and the reflected rays. This means 
that the coefficient of reflection for water for this 
polarization is almost 100 per cent for the rather small 
angles which the rays made with the water. With ver¬ 
tical polarization the minima were not nearly so deep. 

The conclusion to be reached from these tests is that, 
for a communication circuit over salt water, vertical 
polarization is definitely indicated. 

5 4 PROPAGATION OVER LAND 

Three tests were made to investigate propagation 
over land. These consisted of a variable antenna height 


test, a test of signal strength as received in an auto¬ 
mobile at various locations, and recordings of signal 
strength between fixed locations versus time. 

Variable Antenna Height Test 

The klystron transmitter and reflector were mounted 
in a small elevator suspended from one of the towers 
located at Rocky Point. The receiving antenna used 
was a horn with an opening of 2.14x2.65 ft and having 
a gain of 23 db over a half-wave (A/2) dipole. This 
was mounted 70 ft above ground and connected to the 
receiver by means of a waveguide. The transmitter 
and receiver were 14.5 miles apart. The gain of the 
parabolic receiving antenna was also 23 db over a 



Figure 3. Received signal strengths for horizontal 
polarization; transmitter at Mt. Misery, course east 
to Mattituck, Long Island. 


half-wave dipole. As the 2-watt transmitter and an¬ 
tenna were raised and lowered, signal strengths were 
measured at the receiver. The maximum received 
signal occurred within a few feet of the height cal¬ 
culated from the profile. Vertical polarization was 
employed. (See Figure 4.) 


ONFIDENTIAL 






































































CRYSTAL-RECEIVER TESTS 


21 


Automobile Tests 

The transmitter antenna was mounted 80 ft above 
ground (Building No. 10 at Rocky Point) and was 
fed through a waveguide. The receiver with a small 
horn antenna was transported by automobile to vari¬ 
ous locations. The signal strengths were quite variable, 
in fact “shotgun” in pattern. The signal disappeared 
quite often when the car was moved a foot or two. 
Local reflections and absorption were quite noticeable 
from objects in the vicinity. 



HEIGHT OF TRANSMITTING ANTENNA ABOVE GROUND IN FEET 

Figure 4. Signal strength versus height for 10-cm 

signals, vertical polarization, Rocky Point to Riverhead, 

Long Island. 

Recouping Tests 

Two sets of recordings of the signal strengths 
versus time were made over a distance of 14.5 miles 
(between Building 10 at Rocky Point and Building 
9 at Riverhead). The receiver employed a 23-db horn 
antenna. The recordings were of chief interest because 
of the severe refraction occurring on the evening of 
November 5. (See Figure 5.) Similar conditions were 
observed over nearly the same path on 500 me which 
was also being recorded. 

Severe refractions were noted again on the night 
of February 15 and 16. No effect was noticed due to 
heavy rain, fog, or light snow. No selective fading or 
static was noticed at any time. Ignition noise entered 
through the 30-mc i-f amplifier but this could be 
removed with adequate shielding. 

The conclusion reached is that 10-cm communica¬ 
tion over land woidd be satisfactory with either hori¬ 
zontal or vertical polarization, provided the terminals 
of the system were within line-of-sight of each other. 



AM PM AM PM 

Figure 5. Recording of 10-cm propagation, vertical 
polarization, Rocky Point to Riverhead, showing abnor¬ 
mal conditions on evenings of November 5 and 9. 


55 CRYSTAL-RECEIVER TESTS 

Since it has been suggested quite often that a simple 
crystal detector followed by an audio-frequency am¬ 
plifier could be used for portable reception, some tests 
were made to determine the feasibility of such a simple 
receiver. The tests showed that such a receiver had a 
very poor equivalent noise side-band input [ENSI] b in 
the neighborhood of 100 fix across 75-ohm input ter¬ 
minals, which is over 40 db poorer than the crystal 

b ENSI (equivalent noise side-band input) is the equivalent 
input magnitude of all random noise which is transferred to 
the output circuit, and therefore, of all such noise within the 
frequency side-bands passed by the receiver. 




























22 


FIELD TESTS AND 3,000-MC EQUIPMENT 


heterodyne receiver employed in the tests or 65 db 
poorer than an ideal receiver in which the noise would 
be proportional to the absolute temperature T and 
the frequency band A/. 

This receiver with a 15-db antenna was carried 
about in the vicinity of an omnidirectional trans¬ 
mitter having an antenna with a gain of about 6 db 
with an input of 2 watts. Over a distance of about one 
mile, reception was very unsatisfactory due to the 
presence of trees and other obstructions. Behind large 
trees no signals could be heard, and even in fairly clear 
places the standing-wave pattern was such that the 
receiver had to be moved to keep away from zero- 
signal points. 

The results indicated that reception with such a 
simple receiver could be expected up to a distance of 
about a mile but that continuous reception by a walk¬ 
ing or moving person was not practical. 

56 EQUIPMENT EMPLOYED 

Two transmitters and three receivers plus accessory 
equipment were employed in these tests and a consid¬ 
erable number of measurements were made on stabil¬ 
ity, modulation capabilities, receiver sensitivity, and 
frequency modulation versus amplitude modulation. 

5.7 TRANSMITTERS 

One of the transmitters furnished for the prelimi¬ 
nary investigation of 3,000-mc propagation character¬ 
istics was a Westinghouse klystron unit supplied 
under Project C-10. 3 The other was a Western Elec¬ 
tric unit which is described in Chapter 6. 

Table 1 gives salient facts about the 2 transmitters. 


Table 1. Comparison of 3,000-mc transmitters. 



Western Electric 
transmitter 

Westinghouse 

klystron 

transmitter 

Size 

66^x22x17 in. 

37x21x15 in. 

60-cycle power input 

775 watts 

450 watts 

Power output at 3,059 me 4 watts 

4 watts 

Plate supply ripple 

—60 db 

—68 db 

Plate supply bounce 

About 3.5 volts 

About 6 volts of 

Accuracy of 

of 1,500 volts 

2,500 volts 

frequency setting 

±1.0 me 

No provision 


The frequency drift of the two transmitters during 
the warm-up period is shown in Figure 6. Both trans¬ 
mitters were operated from a voltage-regulated power 



Figure 6. Frequency drift after starting 3,000-mc 
transmitter. 


supply. The instantaneous frequency stability of the 
two transmitters was comparable, the klystron trans¬ 
mitter being frequency-modulated ±0.2 me when it 
was amplitude-modulated 35 per cent, the Western 
Electric transmitter having a band of 1 me. These 
figures were obtained in listening tests measuring the 
width of the frequency spectrum with a receiver. The 
wider band of the Western Electric transmitter can 
easily be explained by the harmonics in its 16-kc 
pulses (CFVD).° 

5 - 71 Frequency versus Temperature 

Frequency and ambient temperature measurements 
on the klystron transmitter showed that there was no 
definite correlation between these two factors, indicat¬ 
ing that other causes existed for the frequency varia¬ 
tions. At the time of the measurements, varying 
weather conditions disturbed the impedance match 
between the transmitting antenna and its waveguide 
feed which, in turn, presented a variable load to the 
transmitter, at times causing the transmitter to stop 
oscillating. 

When the horn antenna was used, greater stability 
was experienced, probably because it was more water¬ 
proof than the parabolic reflector antenna. With the 
horn, recordings indicated that the transmitter stayed 
within the 2-mc pass band of the recording receiver 
over a temperature variation of approximately 63 to 


c See Chapter 6. CFVD indicates a continuous-frequency vari¬ 
able-duration pulse system. 


CONFIDENTIAL* 















TRANSMITTERS 


23 


75 F. During this period the transmitter did not 
stop oscillating of its own accord. Both transmitter 
and receiver were operated from a voltage-regulated 
power source. 

An increase of 1 per cent in the 115-volt power 
supply to the klystron transmitter caused —180 parts 
per million change in output frequency. This corre¬ 
sponds to a 540-kc frequency shift at 3,000 me. 

5 7 2 Automatic Frequency Control 

The commutator of the automatic frequency control 
of the Western Electric transmitter was replaced with 
the balanced detector circuit shown in Figure 7. In this 
circuit a synchronous motor drives a four-pole vari¬ 
able condenser at 1,800 rpm. Thus, when the frequency 
is off the resonant frequency of the monitor cavity, a 
120-cycle voltage is generated and is impressed in push- 
pull upon the grids of the balanced detector. Plate 
voltage for the detector is obtained from the input to 
the magnetic field rectifier supply and while it should 
be a 120-cycle sine wave, the 60-cycle full-wave recti¬ 
fied voltage was satisfactory. The performance of the 
automatic frequency control system was the same as 
with the commutator and had no sliding contacts. It 
used the same number of tubes as the original circuit. 

5 7 3 Transmitter Modulation 

A modulator similar to the Western Electric trans¬ 
mitter modulator was built for the klystron transmit¬ 
ter. In this system the transmitter is keyed with 50 
per cent square dots under the no-modulation condi¬ 
tion. At the peak of the audio cycle, the keying mark 
goes to 100 per cent (up) and to zero (down). An 
ordinary a-m receiver with a diode detector followed 
by a 5-kc low-pass filter will receive such transmission. 

The advantages of such a CFYD system are: 

1. Frequency modulation of the transmitter is min¬ 
imized which greatly eliminates distortion in the re¬ 
ceiver due to its not having a perfectly flat band-pass 
characteristic. 

2. Full modulation capabilities are easily realized 
without undue distortion. 

3. Loading and tuning of the transmitter are less 
critical. 

In this system the pulse frequency was 21 kc; the 
build-up time of the pulses was 1.0 /xsec . The klystron 
grid was driven with a pulse amplitude of 100 volts 


peak to peak. Since there was no direct current on the 
grid, the effective modulating voltage was half this 
value because the klystron is cut off on the negative 
swing. By applying positive bias voltage to the kly¬ 
stron, the maximum positive voltage reached by the 



Figure 7. Automatic frequency control for Bell Labora¬ 
tories transmitter. 

grid is increased with a corresponding increase in 
average power. Adding 15 volts positive to the grid, 
in this case, increased the modulating peak voltage 
to 65. 

57 4 FM versus AM 

To compare frequency modulation with pulse modu¬ 
lation, the frequency of the klystron was modulated 
by varying the plate voltage through the output trans¬ 
former of the modulator placed in series with the plate 


c 


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i 
















































































































24 


FIELD TESTS AND 3,000-MC EQUIPMENT 


supply to the klystron cathode. The resultant ampli¬ 
tude modulation in the receiver was limited out. A 
modulating voltage of 50 volts peak to peak gave a 
peak frequency deviation of 300 kc. This corresponded 
to 100 per cent modulation with a receiver pass band 
of 600 kc. 

A signal-to-noise ratio improvement of 60V 3 should 
be obtained using frequency modulation; the narrow 
frequency band-pass of the receiver, however, made 
it apparent that automatic frequency control would be 
necessary to keep the transmitter from drifting out 
of the pass band of the receiver. 


5 - 7 - 5 Push-to-Talk Circuit 

To turn the transmitter off, the length of the co¬ 
axial line to the antenna change-over switch was ad¬ 
justed so that when the switch was opened the reactive 
load placed upon the transmitter caused it to stop 
oscillating. This maintained constant input power to 
the transmitter cavity. The transmitter did not come 
on at the same frequency at which it went off, how¬ 
ever, and special means had to be taken to increase 
the power to the transmitter during the stand-by 
period. This was accomplished by increasing the plate 
voltage from 1,600 to 1,775 volts by means of a relay 
operated by the change-over switch. 



Figure 8. Ultra-high-frequency converter using 1221-Y 
oscillator and crystal detector. 


Receiver Performance 


58 RECEIVERS 

Three receivers were tested during this study. One 
of them employed a crystal converter with a 1221-Y 
oscillator; another used a special tube containing a 
resonant-cavity oscillator into which was coupled a 
hairpin circuit tuned to the signal frequency, the out¬ 
put being amplified in an 8-mc amplifier; the third 
receiver used a beam deflection tube. 

5 - 81 Crystal Converter 

This receiver (Figure 8) was made from parts sup¬ 
plied by the Western Electric Company. Small changes 
in oscillator frequency were made by means of a screw 
driver on one of the oscillator plate inductors. The 30- 
mc i-f amplifier (IR-202), developed and manufac¬ 
tured by RCA under Project C-2, 5 had a mid-band 
frequency of 30 me, an equivalent band width of 2.75 
me at full gain and 3.25 me at somewhat lower gain. 
The automatic gain control characteristic was quite 
flat above 15 to 20 ju,v input. 


Frequency Stability. The frequency change was 
2,070 kc or 28 parts per million per degree (ppm/°C) 
over an ambient temperature variation from 36 to 
81 F. A -J-l per cent change in plate voltage caused 
a change of —|—55 parts per million. A -j-l per cent 
change in filament current caused a frequency change 
of —|—73 parts per million; changing the line voltage 
to the oscillator power supply by -j-l per cent pro¬ 
duced a frequency change of —J—69 parts per million. 

Input Noise Characteristic. The ENSI of this re¬ 
ceiver with its i-f amplifier was 0.8 fix across 75 ohms 
for an r-f band width of 10 kc. This is 23.2 db poorer 
than the best ideal receiver (KTAf base). This meas¬ 
urement was made with a 3,060-mc r-f carrier well 
above the peak noise level in the i-f amplifier, so that 
the intermediate-frequency threshold was not a factor 
in determining the noise. The ENSI of the i-f ampli¬ 
fier itself was 0.21 fix under the same conditions, 
showing that most of the noise came from the crystal 
converter. 

Sensitivity. To determine the weakest signal that 
could be received, taking the threshold into account, 


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WAVEGUIDES AND ANTENNAS 


25 


an a-f amplifier having an equivalent band width of 
5 kc was used. Under these conditions a 100 per cent 
modulated signal of 3.7 /xv across the 75-ohm input 
gave a signal-to-noise ratio of 10 db. 

RCD-23 Receiver 

The 3,000-me converter of this receiver made by 
the RCA Manufacturing Company was a resonant 
cavity oscillator plus an i-f amplifier with a mid-band 
frequency of 8 me. Due to the fixed oscillator cavity, 
tuning over a limited range was accomplished by 
varying the cavity voltage and the input tuning. 

Stability. Oscillator stability with temperature was 
less than 7 ppm/°C. Changes in cavity and cathode 
voltages of —J—1 per cent produced frequency changes 
of -j-400 parts and —100 parts respectively. 

ENSI. At 3,060 me, ENSI was 3.8 yx across 75 
ohms for an r-f band width of 10 kc using a signal 
above the peak noise of the i-f amplifier. This is 36.8 
db poorer than the ideal receiver. With a signal of 
25 yx across the input (below the i-f threshold) the 
ENSI was 5 /xv for a 10-kc band. 

Beam Deelection Converter Receiver CR-301 

In this superheterodyne receiver the frequency con¬ 
verter was a beam deflection tube (H-2214-2A or H- 
2243-1) providing separate deflection circuits for the 
signal and for the local oscillator injection. The signal 
circuit cavity was toroidal in shape and was so con¬ 
structed that the signal voltage between opposite faces 
of the cavity deflected the electron stream as it passed 
through a slot in the cavity. The local oscillator volt¬ 
age was applied to a pair of rods which deflected the 
electron stream before it passed through the signal 
cavity. 

Between the signal cavity and the collector plate 
was a slot bisected lengthwise by a fine target wire. 
When the electron beam was at rest (without oscillator 
or signal deflection voltages) and properly focused and 
centered, most of the electrons were removed by the 
wire. Deflection of the electron beam to either side of 
this central partially masked position produced an 
increase in collector current. Thus when the beam was 
deflected through this central position the slope of the 
curve of collector current versus deflection voltage 
reversed. This reversal of slope produced a large value 
of conversion conductance. The collector plate was 
treated to accentuate secondary emission, thereby pro¬ 
viding one stage of electron multiplication. 


The local oscillator of this receiver was a ZP-446 
equipped with a regulated power supply. 

The ENSI of this receiver was 0.9 /xv across 75 
ohms for a 10-kc band width. This is 24 db poorer 
than an ideal receiver. 

5.9 WAVEGUIDES AND ANTENNAS 

Several types of waveguides were studied with a 
view to finding the most practical method of feeding 
antennas at a distance from the transmitter. In the 
region studied (3,000 me), feed lines of about 100 ft 
of ordinary coaxial cable have too high a loss. Horn 
and reflector types of antennas were examined as to 
their practicability for communication systems at 
these frequencies. 

5 - 91 Waveguides 

Because of the relative ease of bending rectangular 
guides and the better control of polarization compared 
to circular guides, two types of rectangular guides 
were used: 

1. Copper rain spouting 3%gx2% in. with 0.020- 
in. walls, corrugated lengthwise, was used in 10-ft 
lengths. The measured loss was 0.39 db per 100 ft at 
A = 9.8 cm. This figure compares with the calculated 
loss of 0.36 db. 

2. Commercial bronze rectangular tubing 3x1 x /z in. 
with 0.064-in. walls, hard-drawn, 90 per cent copper, 10 
per cent zinc. Calculated loss was 0.85 db per 100 ft 
at A = 9.8 cm; measured loss was 0.79 db. 

Various bends were tried. If the radius of curvature 
of the inner surface was 6 in. or more, negligible re¬ 
flection occurred. Tapered sections of various lengths 
showed reflections varying around 1 per cent. 

Small amounts of water in the waveguides increased 
the losses greatly. 

5 - 9 - 2 Antennas 

Two horn antennas with an aperture of 2.65x2.14 
ft were made of plywood and lined with copper foil. 
They had a power gain of 23 db over a A/2 dipole at 
3,000 me. Another horn with a 1-ft square aperture 
having a calculated gain of about 15 db was used on 
both the land and the salt water surveys. 

The paraboloid reflectors 30 in. in diameter were 
used for transmission. One was fed with a A/2 vertical 
dipole; the other reflector had the same feed system 
but with the addition of a parasitic tuned dipole in 


CONFIDENTLY 





26 


FIELD TESTS AND 3,000- MC EQUIPMENT 


front of the radiator substantially to eliminate direct 
forward radiation. The directive pattern of this an¬ 
tenna is shown in Figure 9 where it will be seen that 
the pattern is quite narrow. It measures ±7° wide 6 
db down. The horn antennas were down 6 db at ± 8 ° 
in the horizontal plane and at ± 6 V 2 ° in the vertical 
plane. 



ANGLE FROM BEAM CENTER IN DEGREES 

Figure 9. Radiation pattern of 30-in. reflector with 
dipole and parasitic unit. 


5 - 9 - 3 Loss in Plywood 

In testing antennas it was desirable to locate the 
units inside some sort of building and for this reason 
it was important to know the effects of plywood on 
radiation from antennas. Tests were made, therefore, 
to find out something about the losses in plywood. 

The effects of the material upon a free space wave 
would be very difficult to measure experimentally. 
Such properties, however, may be indirectly deter¬ 
mined by placing slabs of the material in a rectangu¬ 
lar waveguide. 

Various thicknesses of plywood were obtained by 
first making a thick slab from Vs- and Vi-in. pieces of 
board held together by small bakelite dowels. By using 
various combinations of thicknesses, steps in thickness 
of approximately V& in. were obtained. 

5 - 9 - 4 Measurement Procedure 

A rectangular waveguide was terminated in a horn 
at it's far end. Beyond the horn was a device for meas¬ 
uring radiated power. A standing wave detector was 


near the transmitter. The guide was perfectly matched 
by means of a variable capacitance. Then the slabs 
were inserted between the standing wave detector and 
the horn. The power output and both the magnitude 
and position of the standing waves back of the plywood 
were measured. 

From these data the ratio of power input to power 
output and the magnitude of the coefficient of reflec¬ 
tion could be determined. From the magnitude and 
position of the standing wave, the impedance looking 
toward the load from the back side of the plywood slab 
was calculated. These data are plotted in Figures 10 
and 11 . Note that the impedance is plotted in the com¬ 
plex plane and forms a spiral. The efficiency of trans¬ 
mission for a V 2 -L 1 . ( 12 -mm) slab of dry plywood is 
about 60 per cent. Soaking the slab over night in water 
reduced its efficiency to about 43 per cent. 



Figure 10. Transmission efficiency and reflection coeffi¬ 
cient, dry plywood at 9.7-cm wavelength. 


Minimum reflection takes place when the thickness 
of the wood is approximately A /2 for a wave propa¬ 
gated through plywood in the 2x3-in. waveguide. If 
the material were lossless the reflection would be zero 
for the A/2 thickness. 

From these data the dielectric constant of the ply¬ 
wood is estimated at about 2.33. There is some uncer¬ 
tainty about this figure since the data were not consis¬ 
tent, probably because this wood is nonhomogeneous 
and the properties of pieces probably vary substan¬ 
tially. Another method , 6 used at the Massachusetts In¬ 
stitute of Technology [MIT], in which standing waves 
in a waveguide terminated in a conducting sheet are 
measured, indicates a value of 1.9 for the dielectric 
constant of plywood at A = 6 cm. It is quite possible, 
however, that the plywood tested at MIT was drier 
than the samples examined under this project. 


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OMNIDIRECTIONAL MICROWAVE TELEPHONE 


27 


Power Measurement 


Power measurements were made with a load resistor 
consisting of 41 in. of No. 38 Nichrome wire (41.6 
ohms per ft) wound in a 20-turns-per-in. square 
thread on a 1 % 6 -in. diameter brass cylinder. The square 
thread was 0.027 in. deep by 0.020 in. wide. The wire 
was kept in the center of the groove by a 0.01-in. silk 
thread. A brass sleeve (% 6 -in. wall) was pushed over 
the cylinder. This sleeve had a coaxial fitting at one 
end with the inner conductor connected to one end of 
the Nichrome wire. The other end of the wire was 
connected to a terminal on the cylinder. The cylinder 
had a well to receive a thermometer. 

The section of the waveguide to which the resistor 
was attached was connected to the waveguide from a 
3,000-mc transmitter. The waveguide-to-coaxial-line 
transformer was adjusted so that the load resistor was 
matched to the waveguide (approximately 50 ohms). 



RESISTANCE 


Figure II. Impedance versus thickness, plywood in 
waveguide. 

The temperature rise with time was compared with the 
curves of Figure 12 to determine the power into the 
resistor. If the power flowing through the waveguide 
was known, the crystal probe could be calibrated. 


5.io OMNIDIRECTIONAL MICROWAVE 
TELEPHONE 

Following the preliminary investigations carried out 
under Project C-24 and described above, Project C-42 
was set up to develop an omnidirectional telephone 


system and a directional microwave telephone, using 
the data secured under C-24. 



Figure 12. Calibration of ultra-high-frequency load 
resistor used in power measurements. 


The omnidirectional system was desired by the 
Navy as an additional telephone communication chan¬ 
nel having a certain amount of secrecy, operating 
somewhere in the region of 1,200 to 2,700 me and hav¬ 
ing a range in any direction of 10 miles, all stations 
transmitting on the same frequency. The equipment, 
designed to operate at 1,400 me, consists of two trans¬ 
mitter-receiver combinations mounted with their as¬ 
sociated control equipment. Overall system tests were 
made between Rocky Point and Riverhead, a distance 
of 14.5 miles. The transmitter antenna, a vertical 
doublet, was mounted 77 ft above ground. The receiver 
antenna was mounted 49 ft above the ground and con¬ 
sisted of a vertical doublet with a parabolic reflector. 
With 30-watt average power output from the trans¬ 
mitter at Rocky Point, the power delivered to the re¬ 
ceiver at Riverhead was 3.3 g/iw. A block diagram of 
the system is shown in Figure 13. 

5101 Transmitter 

The transmitter tube was a magnetron having an 
average output of 30 watts on a carrier frequency of 


CONFIDENTIAL 


I 














28 


FIELD TESTS AND 3,000-MC EQUIPMENT 



Figure 13. Block diagram of nondirectional system 
working on 1,400 me. 

about 1,400 me. The output signal consisted of 0.8- 
H see pulses having an unmodulated pulse rate of 20 
kc. The pulse rate was frequency modulated, the maxi¬ 
mum deviation being ±3 kc. The frequency-modu¬ 
lated pulse system provided an improved signal-to- 
noise ratio at the receiver over that obtained by other 
pulse systems and simplified the transmitter design. 


Furthermore this system lowered the noise threshold 
by 12 db compared to a c-w system of the same power, 
representing an increased effective range for the sys¬ 
tem. (A circuit diagram of the pulse transmitter is 
given in Figure 14.) 

The output from a single-button carbon microphone 
(Western Electric Type F3) was sufficient to modulate 
the transmitter 100 per cent. An audio limiter pre¬ 
vented overmodulation due to high input from the 
microphone hut permitted a reasonable per cent modu¬ 
lation for weak microphone input voltages. 

The transmitter operated from 110-volt 60-cycle 
mains and required 750 watts at a lagging power fac¬ 
tor of 0.9. 

5102 Magnetron Details 

The r-f oscillator and output tube was an air-cooled 
multianode magnetron, the output carrier frequency 



Figure 14. 


Circuit diagram of 1,400-mc pulse transmitter for nondirectional 


system. 


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OMNIDIRECTIONAL MICROWAVE TELEPHONE 


29 



Figure 15. Interior view of pulse transmitter. 

of which was determined by the dimensions of the 
tube. The magnetic field was supplied by permanent 
magnets of Alnico V and was adjusted by a variable 
shunt across each of the two magnets. 



CARRIER FREQUENCY CHANGE FROM OPTIMUM IN MC 


Figure 16. Effect on output of changing frequency by 
adjusting magnetron magnetic field and matching- 
circuit. 

The anode of the magnetron was at ground poten¬ 
tial, excitation being provided by negative pulses ap¬ 
plied to the cathode. These pulses were supplied by a 
pair of 813 beam-power tubes which were partially 
driven by regeneration from their own output and 
partly from a pulse oscillator using an 807 tube. 


Table 2. Magnetron data. 


Average power output 

35 watts 

Magnetic field 

about 900 gausses 

Average d-c current 

26 ma 

Peak pulse voltage 

9,700 volts 

Filament voltage 

6.3 volts 

Filament current 

1 amp 


An artificial line with a delay from one end to the 
other of about 0.5 /xsec, located in the cathode circuit 
of the 807, determined the pulse length. The frequency 
of the pulses was controlled by a 6SK7 frequency- 



0|2 3 4 5 

MINUTES FROM TIME PLATE VOLTAGE IS ON 

Figure 17. Frequency drift of transmitter. A, trans¬ 
mitter cold, filaments heated 1 minute; B, after 5 
minutes stand-by period. Room temperature, 20 C. 



90 100 110 120 130 

VOLTAGE OF 60 CYCLE POWER SUPPLY 


Figure 18. Relation between carrier frequency change 
and supply voltage changes. 



90 100 110 120 130 

VOLTAGE OF 60 CYCLE POWER SUPPLY 

Figure 19. Effect on power output of changes in power 
supply voltage. At each voltage, magnetic field was 
adjusted for maximum output. 

modulator tube. Another 6SK7 and a GH6 limited the 
maximum audio voltage applied to the control grid 
of the frequency-modulator tube. 

The magnetron gave its greatest output at a par¬ 
ticular frequency, and the output decreased if the load 
impedance and the magnetic field were changed to 
alter the frequency (Figure 16). The output fre¬ 
quency varied with changes in temperature (—13 
cycles per million per degree C) and with line voltage 
(see Figures 17 and 18). The output power varied, of 
course, with changes in magnetron voltage as shown in 
Figure 19. 


CONFIDENT! 


















































































30 


FIELD TESTS AND 3,000-MC EQUIPMENT 










* 

































90 100 110 120 130 

VOLTAGE OF 60 CYCLE POWER SUPPLY 


Figure 20. Effect on pulse frequency of changes in line 
voltage. 


510 3 Transmitter Operation 

Operation of the pulse transmitter (Figure 14) is 
as follows: 

An increase of plate current of the 807 (Fj) 
through winding 1-2 of transformer T x induces a 
positive voltage on the control grid. This further in¬ 
creases plate current and results in a rapid rise of 
plate current through the tube to a high value. 

The current flows from the cathode into the arti¬ 
ficial line which during the current pulse is equivalent 
to a 152-ohm resistance (the characteristic impedance 
of the line) connected from cathode to ground. The 
voltage pulse produced by the current flowing into the 
line travels along the line to the open end, that is, to 
the end to which the anode of the 6SK7 is connected. 
This end is in effect open-circuited because of the high 
impedance of the 6SK7 compared to the line imped¬ 
ance. The voltage pulse is reflected without phase re¬ 
versal and travels back along the line to the cathode. 
When the return pulse front, now at twice the original 
voltage, reaches the cathode, the cathode voltage rises 
and decreases the plate current. This causes a decrease 
in grid potential, which rapidly decreases the plate 
current to cutoff. The line will then be charged to a 
potential equal to about twice the cathode-to-ground 
potential at the beginning of the pulse. This charge 
will be reduced at a uniform rate by the plate current 
of the frequency-modulator tube V 2 . When the cath¬ 
ode potential of the 807 reaches about 50 volts, the 
tube will again conduct current. 

The rate at which the charge on the line is reduced 
and the value of the initial charge on the line deter¬ 
mines the pulse frequency. The plate current of V 2 
determines the rate of discharge of the line. The plate 
current is controlled by the screen potential and the 
cathode-to-grid potential. At the end of the pulse, the 
voltage charge on the line is proportional to the 500- 
volt supply. Thus if the supply rises in voltage it will 
be necessary to increase the plate current of V 2 to 


maintain constant pulse frequency. The plate current 
of F 2 is increased by increasing the screen-grid poten¬ 
tial obtained from the 5 00-volt supply through R 4 
and R 5 . 

Since it was necessary to by-pass the screen grid 
of V 2 for pulse frequency, C 2 was shunted across R 4 
and R 5 so that the phase and amplitude of ripple fre¬ 
quency voltages would be maintained at their proper 
value at the screen grid. In other words, the time con¬ 
stant on R 4 , R 5 , C 2 was made equal to the time con¬ 
stant of by-pass capacitor C 3 and the screen grid 
resistance to ground. 

Pulse frequency is modulated by voice frequencies 
by varying the grid voltage of V 2 . A peak voltage of 
2.2 volts on the control grid modulates the pulse fre¬ 
quency ±3 kc. The average pulse frequency is ad¬ 
justed by R 4 to 20 kc. 

Audio Limiter 

In order that modulation voltages applied to the 
control grid of V 2 should not exceed 2.2 volts, an 
audio voltage limiter is placed in series with the audio 
input. This limiter consists of V 3 whose amplification 
is controlled by the bias applied to its control grid by 
the left half of the 6H6 diode. The rate of decrease of 
this bias is governed by R 1 C 1 . This bias will drop to 
half value in about 1 second. The potential appearing 
across terminals 8 and 6 of the secondary of the modu¬ 
lation transformer is reduced by the divider R 2 R 6 . 
The volume control is set so that with maximum input 
from the microphone the peak voltage on the control 
grid of V 2 is 2.2 volts. 

When the microphone switch is closed, the surge of 
current through the primary of the microphone trans¬ 
former induces a high voltage in the secondary. The 
control bias resulting would keep the limiter cut off 
for several seconds and to prevent this from happen¬ 
ing, the biased diode (right half of the 6H6) limits 
the maximum voltage which can be applied to the grid 
of F 3 . 

Pulse Control 

The pulse output of V x is amplified in V 5 (two 813 
tubes in parallel) through T 2 . Load current of the 
amplifier tubes flows through R 3 which is in series 
with the pulse autotransformer T 3 and across which 
is one winding of T 2 . The voltage drop across R 3 is 
stepped up and reversed in polarity by T 2 . The plus 
pulse voltage from oscillator V x starts the amplifier 
tubes conducting. The increase in plate current lowers 


CONFIDENTIAL^ 



















RECEIVER 


31 


the plate voltage which is again lowered by the in¬ 
crease of plate current due to the voltage across ter¬ 
minals 1 and 2 of T 2 . The plate voltage thus drops to 
a few hundred volts. At the end of the pulse period 
the output of the oscillator drops to zero, reduces the 
plus voltage on the control grids of the amplifier tubes 
and their plate current increases. The regenerative 
action in T 2 causes the amplifier current to decrease 
rapidly to zero. 

Sufficient fixed bias is supplied to the amplifier 
tubes to keep emission at zero or nearly zero in ab¬ 
sence of pulse voltage excitation. During the opera¬ 
tion the bias is increased by emission current so that 
the tubes are definitely cut off between pulses. This 
shortens up the output pulse and insures against pass¬ 
ing any transients which may follow the pulse. Di¬ 
mensions and windings of the pulse transformers are 
given in the final report 2 on Project C-42. 

Output Circuit 


With no r-f voltage on the fuse the bridge is bal¬ 
anced and no current flows through the indicating 
meter. Radio-frequency voltages applied to the fuse 
change its resistance, upsetting the bridge balance. 
The resultant meter reading is proportional to the 
square of the r-f voltage. Polarizing voltage for the 
bridge is obtained from a 100-volt rectifier. The bolo¬ 
meter may he used to determine standing waves on the 
output transmission line or on the wavemeter to deter¬ 
mine the output frequency. 

Wavemeter 

The wavemeter was a %-wavelength section of coax¬ 
ial line. The length of the inner conductor could be 
varied 1 in. by a micrometer head at one end of the 
wavemeter, giving a frequency range of from 1,300 to 
1,500 me. At 1,400 me a micrometer change of 1 divi¬ 
sion (0.001 in.) corresponds to a change of 0.220 me 
of the resonant frequency. 


Output power is indicated by a bolometer capaci- 
tively coupled to the output transmission lines. The 
bridge circuit, shown in Figure 21, is used to indicate 


-DC 



Figure 21. Circuit of bolometer bridge. 


a change of resistance of the bolometer resulting from 
heating by the r-f currents. The bolometer itself is a 
5-ma vacuum fuse, one end of which is connected to 
ground through a section of coaxial line 1 in. long. 
This end is exposed to the r-f voltages. By acting as 
an inductance, the short length of line partially tunes 
out the capacitance to ground of the fuse cap. The 
other end of the fuse is by-passed to ground and is 
connected to the bridge through a flexible coaxial 
cable. 


5ii RECEIVER 

The receiver employed (Figures 22, 23, 24) covered 
a frequency range of 1,350 to 1,545 me and consisted 
of an r-f amplifier, converter, and oscillator using 
GL-446 tubes followed by a 30-mc mid-band i-f am- 



Figure 22. Block diagram of receiver. 

plifier having a band width of approximately 3 me. 
The i-f amplifier was followed by a biased detector 
having a hand-pass transformer in its plate circuit 
which converted received pulses into sine waves. The 
sine waves then passed through the limiter, discrimi¬ 
nator, low-pass filter, and audio amplifier diagrammed 
in Figure 25. 

The receiver had an excess noise ratio [ENR] d of 
from 16 to 17 dh (kTAf base) while the converter 
alone had an ENR of 21 db. 


d The excess noise ratio is a measure of the excess of the 
measured noise power output over the ideal noise power from 
the thermal agitation of the signal source. 


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32 


FIELD TESTS AND 3,000-MC EQUIPMENT 



Figure 23. Radio frequency unit of directional receiver. 
OSCILLATOR 

The oscillator frequency was lowered 340 ke from 
1,480 me (0.023 per cent) by a 10 per cent increase in 
line voltage and was lowered 1,280 kc (0.086 per cent) 
by a 10 per cent decrease in line voltage. Oscillator 
frequency decreased 780 kc from 1,470 me for a 20 
C increase in temperature, which is a temperature co¬ 
efficient of —26 ppm/°C. 

Intermediate-Frequency Amplifier 

The converter was followed by a 5-stage 30-mc i-f 
amplifier, the first stages of which had automatic gain 
control [a-g-c] and the last two had fixed gain. If all 
5 stages had a-g-c, there would not be enough voltage 
to operate the detector at high signal levels. The 
equivalent band width varied with received signal 
strengths due to the change in gain with a-g-c. The 
ENR of the i-f amplifier was 13.5 db. 

Detector-Discriminator 

The i-f amplifier was followed by a biased detector 
having in its plate circuit a tuned transformer with a 
pass band of 17 to 23 kc. This converted the received 
pulses into sine waves. The transformer fed a limiter 
tube which in turn fed a discriminator followed by a 
4-kc low-pass filter to keep the 20-kc subcarrier out 
of the audio amplifier as well as to limit the audio 
band width to the speech range. 

Automatic Gain Control 

Automatic-gain-control voltage was obtained from 
a 6H6 connected across the output of the last i-f am¬ 


plifier stage and was amplified by a 6SJ7 (Figures 26 
and 27). The a-g-c voltage was also used to control the 
grid bias of a 6J5 which operated a carrier-alarm re¬ 
lay. The relay served two functions. When no signal 
was being received, it short-circuited one-half of the 
audio output transformer and prevented the noise 
level in the receiver output from rising too high. Dur¬ 
ing a signal period, the short circuit was removed and 
a d-c voltage was supplied to the 600-ohm line to oper¬ 
ate the alarm relay in a distant callbox. 

Antenna 

The antenna consisted of a vertical dipole and two 
high-impedance chokes to keep energy from traveling 
down the outside of the transmission line. The an¬ 
tenna was designed to match the 75-ohm line. 

Callbox 

The receiver had a relay operated by a-g-c voltage. 
In the absence of a signal this relay shorted half of 
the audio-output transformer and acted as a partial 
squelch. When a signal was received, the squelch was 
removed and a voltage to ground was applied to both 
sides of the handset-earphone line. This voltage was 
utilized to operate a carrier-on relay in the callbox 
which in turn could operate any sort of alarm circuit. 

5.12 PERFORMANCE OF OVERALL SYSTEM 

In addition to testing two of the units as a system, 
distortion measurements were made using both single 
tone and double tones. The effect of c-w or 1,000-cycle 
pulse interference on the receiver was investigated. 
Data on these tests are in the final report. 2 

Calculations were made on the probable range if 
one end of the circuit were in an airplane. As the dis¬ 
tance is increased there is an area of continuous recep¬ 
tion followed by an area where variable reception is 
obtained because of cancellation effects between the 
direct and reflected waves. As the distance is increased 
further a point is reached which is the limit of recep¬ 
tion unless the antenna gain is increased. These sig¬ 
nals could be received by an airplane at a distance of 
124 miles if it were flying at an altitude of 1,000 ft. 

sis DIRECTIONAL MICROWAVE 
TELEPHONE 

A second part of Project C-42 was to develop a sim¬ 
ple portable directional microwave communications 




DIRECTIONAL MICROWAVE TELEPHONE 


33 



I^A/Wf-A/W 


itAA/V^a/VV 

3 9 


V 


-C3- 


osc 


L , - 30 t NO. 28 E ON | DIAM FORM 5 /lh 
l_ 2 = 9 ♦ N 0.26 SC C ON -I DIAM FORM 5/ih 
L 4 = II t N 0.26SCC ON -g- DIAM F0RM5yu.h 
Lg= 19 t N0.32E ON jj- DI AM FOR M 5/ih 


L 


1000 


5 


GL-446 


Figure 24. Input circuits to receiver. 

i CONFIDENTIAL 


TO FOUR SIMILAR 
STAGES 















































































































































34 


FIELD TESTS AND 3,000-MC EQUIPMENT 


DISCRIMINATOR 



system to replace visual signaling. The requirements 
were as follows: It should be light enough for two 
men to carry. It should have a minimum battery life 
of 4 hours, a maximum support height of 5 ft, a 
20-degree beam, and provision for six channels. It 



ON]" diameter form 
2 

Figure 26. Circuit of a-g-c (6H6) and detector. 

should provide reliable communication with a clear 
line of sight over a 10-mile distance, simplicity of 
operation, security from interception, and suitability 
for operation in a net. 

So far as these requirements are concerned these 
results were attained: Equipment was built into two 
packs, one weighing 24% lb and the other 31 lb. At 
80 F about 6 to 8 hours of battery life could be ex¬ 
pected which decreased to 4 hours at 20 F. The unit 
was mounted on standard Signal Corps tripods. (Fig¬ 
ure 28 is a photograph of the system with antenna 
and reflector.) The beam was 6 db down at 12° total 
width and 20 db down at 26° total width. Consistent 
reliable communication was attained over distances of 


30 miles from suitable elevations, and from favorable 
hills communication was carried on at 38 miles. The 
system had only one tuning knob and employed push- 
to-talk operation. Interception was difficult due to the 
narrow beam and to rapid attenuation beyond line of 
sight. The units could be adjusted by skilled personnel 
to any of 6 channels 4 me apart. In the field, however, 
the transmitter tuning was fixed and the receiver 
could be tuned about j±2 me from the transmitter 
frequency. Owing to the high directivity, it is doubt¬ 
ful that net operation would be possible. Units could 
operate on common frequency if separated about 50 
ft and if the path in front of each unit were clear of 
reflecting objects. 

5131 Antenna System 


The antenna was a A/2 dipole and parasitic radiator 
housed in a plastic weatherproof box with a paraboloid 
reflector about 30 in. in diameter spun from sheet 



Figure 27. Circuit of a-g-c amplifier and callbox 
control tube. 



































































































































DIRECTIONAL MICROWAVE TELEPHONE 


35 


aluminum. The dipole was fed by a concentric line 
with balanced feed obtained by a A./-t sleeve at the end 
next to the dipole. A short piece of flexible concentric 
line connected the antenna to the oscillator. 

513 2 Power Supply 

The primary power source was three 2-volt, 25- 
amp-hr, lead-cell storage batteries (Signal Corps type 
BA-54-d, Willard-type radio 27-2). The cells weighed 
4(3 lb each. A standard Mallory VP-540 vibrator 
converted this energy to 250 volts at a maximum 
transceiver load of about 25 ma. The battery load was 
approximately 3 amp, receiving or transmitting. 



Figure 28. View of directional telephone system with 
antenna and reflector. 


513 3 Radio-Frequency Unit 

A cavity-type oscillator using a GL-446 lighthouse 
tube furnished output to the antenna. For transmit¬ 
ting the oscillator was cathode-modulated by a two- 
stage audio amplifier and for receiving the GL-446 
was used as a superregenerative detector with separate 
quench oscillator and two-stage audio amplifier. The 


receiver had an ENR only 8 db worse than existing 
superheterodyne receivers using tube converters and 
no more than 16 db worse than the best receiver 
known at the time which used crystal converters. The 
ENR varied with signal strength because the a-g-c 
held the audio-frequency level down. Thus at 10-/iv 
input the ENR was 32 db and with 500-p.v input the 
ENR was 41 db. 



Figure 29. Cavity oscillator for directional system. 

1, hollow cavity resonator operating at X by X/2 mode; 

2, adjustable end plugs, set and clamped or soldered in 
in place; 3, inner conductor of short coaxial resonator 
carrying at one end spring fingers which contact tube 
shell, 5, of the GL-446 which is internally by-passed 
to cathode; 6, grid ring of GL-446 which contacts pair 
of 90-degree sector split contacts connected to left wall 
of cavity; 7, tube-plate ring which, through contact 
springs, 8, and mica by-pass ring, 12, connects to opposite 
wall. Thus coaxial, 3, in combination with slider, 10, 
tunes cathode-to-grid circuit, whereas cavity between 
grid and plate tunes that circuit. 9, spring-loaded 
plunger which holds tube against its spring contacts; 
14, removable cover for changing tube; 11, small wire 
loop coupling plate grid to cathode grid circuit for feed¬ 
back; 13, screw for setting channel frequency; 15 
through 20, solenoid system for tuning receiver and 
compensating transmitter frequency to make it same as 
receiver frequency. 


CONFIDENTIAL 







































































36 


FIELD TESTS AND 3,000-MC EQUIPMENT 


513 - 4 Transmitter 

A minimum transmitter power of 100 mw was 
estimated to be needed with 20-db antennas at each 
end. Measurements on this unit as a transmitter 
showed 400 mw output at 5 watts input. Amplitude 
plate modulation up to 50 per cent was obtained but 
with concommittant frequency modulation of about 
8 kc per volt. It was thought that CFVD modulation 
could be used to reduce the frequency modulation. A 
modulator made up of an 884 sawtooth generator op¬ 
erating at 20 kc, a 6SN7 pulse shaper, and a 6V6 
modulator reduced the frequency modulation of the 
carrier, but actual listening tests showed no better 
results than with straight amplitude modulation. 
Since the CFVD system added several tubes it was 
abandoned. Further amplitude-modulation tests showed 
that at the same modulation capability could be se¬ 
cured with smaller modulator requirements by using 
cathode modulation. A 9002 triode was finally em¬ 


ployed as a modulator. A quench output of 100 volts 
at 100 kc was decided on. 

Measurements indicated that the transmitted and 
received frequencies of the oscillator would differ by 
0.5 to 1.5 me. To overcome this common transceiver 
fault a compensating device was added to the cavity. 
This is illustrated in Figure 29. 

514 FIELD TESTS 

The longest path over which communication was 
held was along Long Island Sound over salt water. 
One terminal was on a 225-ft hill and the other on a 
120-ft hill. Total distance was 38 miles and the visi¬ 
bility was calculated to be 36 miles based on % earth’s 
radius. This path showed some fading, as would be 
expected for a path exceeding the optical range. Other 
tests gave good communication at distances of 29 
miles. Communication was also carried on over water 
with heights of about 4 ft and 15 ft, with the trans¬ 
ceivers on a beach 5.75 miles apart. 


CONFIDENTIAL 






Chapter 6 

R-F GENERATOR FOR 2,000 TO 3,000 MC 


A pulse-modulated telephone generator tuning from 1,980 
to 3,120 me, having a frequency stability better than ±0.05 
per cent over an ambient temperature range of ±10 C, 
producing approximately 10 watts, using a velocity variation 
type of tube. 

61 INTRODUCTION 

A t the time this project 11 was started there was no 
L available equipment for producing voice-modu¬ 
lated power of several watts in the 2,000- to 3,000-mc 
range, with provision for frequency change over so 
wide a range and meeting the requirement of 0.1 per 
cent in frequency constancy at any selected operating 
frequency. 

Under the project a model was developed the oscil¬ 
lator of which is tunable from 1,940 to 3,150 me, al¬ 
though the frequency range of the entire unit is lim¬ 
ited by the tunable range of the monitor cavity which 
is 1,980 to 3,120 me. Modulation up to 100 per cent 
is effected by varying the duration of superaudible 
16-kc pulses. The tube used as oscillator is of the 
velocity variation type, having a focused beam and 
using an external magnetic field. This tube was 
selected because it had reached the point where com¬ 
mercial application was feasible. Its construction 
permits the use of an external circuit so that wide 
tuning range is possible. Other tubes available at that 
time did not permit such a wide range or would not 
deliver so much power. 

6.2 THE oscillator tube 

The heart of the transmitter is the oscillator tube 
(1290-CT) with its associated cavity, two walls of 
which are adjustable by dial controls to permit tuning 
to the desired frequency. By splitting the cavity along 
its center plane the two halves can be pulled apart so 
that the tube can be removed and replaced easily. 

The output coupling system consists essentially of 
a short piece of coaxial cable projecting into the cavity 
and provided with a small coupling loop on its inner 
end. The orientation and location of the loop within 

a Project C-7; Contract NDCrc-177, Western Electric Co., Inc. 
The generator described herein was supplied to RCA Commu¬ 
nications, Inc., for field tests covered in Chapter 5 under 
Project C-24. 


the cavity may be adjusted from outside to vary the 
coupling between cavity and external load. 



Figure 1 . Schematic drawing of “three-gap” velocity 
variation oscillator tube with cavity. 


The cavity is made of rectangular bronze tubing 
4x1% in. outside dimensions and 7 in. long. Two 
plungers are employed having the shape shown on the 
left of Figure 4. The tuning range of the cavity with 
a “three-gap” tube is shown in Figure 5. Some trouble 
was had with parasitic oscillations because the con¬ 
nections to the two center disks were brought out 
through holes in the cavity. The arrangement shown 
in Figure 6 cured this difficulty. The leads from the 
two disks form a transmission line terminated at one 
end by the disks and at the other by a small capaci¬ 
tance equal to about 1 /i/if between the two leads. Re¬ 
sistors connected across this capacitance are isolated 
from the 10- to 15-cm oscillations by small choke 
coils. They stop any tendency to oscillate at about 40 
cm but have no effect on the desired oscillations. The 
length of the transmission line is adjusted by building 
into the cavity with a small copper block so that the 
line length lies between A/2 and A/4 throughout the 
operating range. 

63 MAGNETIC FIELD 

The field required for focusing the three-gap tube 
must be nearly uniform throughout the space occupied 
by the beam and must be variable between about 400 
and 600 gausses. This field is supplied by two field 
coils with a soft iron shell. The requirements on ripple 
suppression were not so severe as those imposed upon 



37 













































38 


R-F GENERATOR FOR 2,000 TO 3,000 MC 



Figure 2A. Photograph of oscillator tube showing demountable cavity. 


the supply to the tube disks, so that no electronic volt¬ 
age regulator was required on the power supply for 
the field. 

64 AUTOMATIC TUNING SYSTEM 


in. At 10 me, however, this motion would have placed 
the plunger beyond the glass of the tube, so the car¬ 
riage carrying the motor was arranged to be retracted 
as the cavity piston approached the tube. The actual 
means by which the frequency is controlled by this 
plunger will be evident from a description below. 


Early tests showed that considerable frequency vari¬ 
ation would be experienced during the warm-up period 
of the oscillator cavity and tube. Variations of about 
5 me at 10 cm and 2 me at 15 cm were encountered, 
indicating that some means of automatic tuning would 
be necessary to hold the transmitter within the speci¬ 
fied frequency. 

The final arrangement consists of a VVin. rod made 
of thin-walled brass tubing arranged to be driven into 
and out of the cavity by means of a reversible motor 
mounted on a carriage at one end of the cavity struc¬ 
ture. The motion of the tuning plunger was set at 1 


65 MONITOR CAVITY 

The standard of frequency about which the trans¬ 
mitter is designed to operate lies in the small so-called 
cavity monitor, consisting essentially of a small wave- 
meter tuned by means of a micrometer head (Figure 
7). This wavemeter consists of a coaxial transmission 
line shorted at one end and open at the other the 
length of which is adjustable to A/4 at the desired 
frequency by means of the micrometer. A small silicon 
crystal rectifies the r-f output of this cavity, the rec- 


CONFIDENTIAL 

















MONITOR CAVITY 


39 



Figure 2B. Photograph of assembled oscillator tube. 


titled crystal current being used for a side-tone circuit 
and for the automatic tuning system. 

A rotating cam enters a slot in this cavity and varies 
the resonant frequency at a rate corresponding to 
about 80 cycles per second. Thus if the mean fre¬ 
quency of the transmitter is on one side of the mon¬ 
itor-resonance curve, the result will be an 80-cycle 
modulation of the output of the cavity crystal. If the 
transmitter is on the other side of resonance with the 
cavity, the 80-cycle modulation will be reversed in 
phase. If the transmitter is at the top of the resonance 
curve, modulation current will vanish. Thus it is only 
necessary to amplify the 80-cycle crystal output and 
to commutate it synchronously, whereupon the direct 
current will be zero at resonance and will have a posi¬ 
tive or negative direction depending upon which side 
of the resonance curve the transmitter is operating. 
An electric filter and a polarized relay controlling the 


direction of motion of the automatic tuning motor 
mounted on the oscillator cavity structure, and there¬ 
fore controlling the position of the tuning rod in the 
transmitter cavity, make up the remainder of the 
tuning system. 

A two-stage amplifier made up of a 6SJ7 and a 
6Y6G tube provides sufficient gain for the automatic 
tuning system even when only one-quarter of the 
normal current is present in the monitor cavity. This 
system holds the average frequency of the transmitter 
to within ±300 kc of the frequency of the monitor. 
The range over which the transmitting frequency can 
be varied by the tuning motor is about 7 me at 15 cm 
and 25 me at 10 cm. This is sufficient to cover the 
variations during the warm-up period, to allow for 
changes in tubes, and for variations in the output 
coupling loop and in the magnitude of the disk 
voltage. 


COYTTDENTIAL 




















40 


R-F GENERATOR FOR 2,000 TO 3,000 MC 



Figure 3. Knocked-down velocity variation tube employed in voice-modulated pulse transmitter. 


CONFIDENTIAL 












CFVD MODULATION SYSTEM 


41 



DISK VOLTAGE IN VOLTS 



Figure 5. Approximate frequency calibration of oscillator. 



Figure 4. Details of two-plunger cavity showing (on 
left) construction of plunger finally used. 

66 CFVD MODULATION SYSTEM 

Various methods of modulating the velocity varia¬ 
tion tube were considered (described in the final re¬ 
port 1 ) with the result that the following scheme was 
selected and put into the model transmitter. 

This method consists, briefly, in modulating the 
length of square pulses of constant amplitude and fre¬ 
quency. One advantage is the fact that special forms 
of limiting can be used at the receiver that cannot be 
used with amplitude modulation. V ith a combination 
of modulation voltages on the accelerating anode and 
on the disks the best linearity and frequency modula¬ 
tion compensation can be obtained. In this way it is 
possible to reduce the frequency modulation from 
about ±1 me to 100 kc, but the addition of more 
filter stages to the disk voltage supply further reduces 
ripple voltage to about 85 db below the d-c output, 


with the result that frequency modulation due to 
ripple voltages is of the order of ±10 to 50 kc. Use 
of a-c heater voltages brought this undesired frequency 
modulation up considerably and was abandoned in 
favor of battery supply. A circuit diagram of the con¬ 
tinuous-frequency variable-duration [CFVD] pulse 
transmitter is given in Figure 9. 



CROSS-SECTION A-A 

C = CAPACITANCE BETWEEN 
1 DISKS*l/u*jf 


C = I p JU f 

R» IOO OHMS COATEO CERAMIC 
RESISTOR 

L' 5 TURNS ON IN. MANDREL 
(NOT CRITICAL) 

Figure 6. Details of cavity and equivalent circuit. 


CONFIDENTIAL 













































































































SIDE-TONE CIRCUIT 
TO HANDSET JACK 


42 


R-F GENERATOR FOR 2,000 TO 3,000 MC 



Figure 7. Mechanical details of 10- to 15-cm wavemeter used as monitor for obtaining automatic frequency control. 


MONITOR 



TO AUTOMATIC 
TUNING MOTOR 


Figure 8. Circuit of frequency monitor and amplifier. 


6-7 PULSE GENERATOR 

The pulse generator consists of a multivibrator 
generating an approximation to a square wave at about 
16 kc followed by a two-stage pulse shaper and a series- 
integrating circuit made up of capacitance and re¬ 
sistance which produces a voltage wave approximately 
triangular in shape. The distance along a horizontal 
line between two successive points on the triangular 
curve is almost exactly proportional to the height of 
the horizontal line above the axis. Thus it is only 
necessary to apply this wave to the input of a tube 
operating around cutoff and vary the bias on the 
tube to correspond with the modulation desired. 

The triangular wave and the audio voltage may, 
ordinarily, be applied to the modulator tube without 
excessive interaction but in this case the small ratio 
between the frequencies of the triangular wave (16 
kc) and the highest audio frequency make the problem 
somewhat more difficult. By means of a cathode fol¬ 
lower connected directly to the cathode of the modu¬ 
lator the triangular wave voltages appear across a low 
impedance, with the result that the triangular wave 
is practically independent of the cathode current. 
Audio modulating voltages are applied directly to the 
grid as a bias variation. 


ONFIDEXTIAJqJ 































































































































































POWER SUPPLY 


43 


EACH TAP BYPASSED TO GROUND 


220V 170V II8V 88V 35V 22V 10V TO MAIN 



Figure 9. Circuit diagram of continuous-frequency variable-duration [CFVD] pulse transmitter. 


Following the modulator is a two-stage pulse shaper 
which provides an output of 60 volts peak-to-peak, 
the shape of which is nearly rectangular. This voltage 
drives an 807 output tube. 

During a cycle of modulation the duration of the 
pulse varies from very nearly zero to a complete cycle, 
thus varying the average power output of the tube. 
Measurements indicated that both second and third 
harmonic distortion in this system are down about 20 
db at 100 per cent modulation. 

This system of modulation was chosen after prelim¬ 
inary experiments indicated that the modulation char¬ 
acteristic of a velocity-variation type of oscillator was 
exceedingly nonlinear and that there was considerable 
spurious frequency modulation. Even in the case of 
the modulation employed (CFVD) a certain amount 
of frequency modulation exists because of the fact 
that the beam in the tube does not reach its final con¬ 


dition until about 5 /xsec after it has been turned on. 
Therefore during the first 5 ^sec of each pulse the 
frequency of the transmitter differs somewhat from 
the frequency during the remainder of the pulse. The 
difference at 10 cm may be as much as 2 me but de¬ 
creases at longer wavelengths becoming negligible at 
15 cm. This frequency modulation does not occur at 
voice frequency rates as in some modulation systems. 

68 POWER SUPPLY 

The main power-supply system is capable of deliv¬ 
ering any desired voltage between 1,500 and 3,700 
volts and is electronically regulated to maintain low 
ripple voltages. The electronic regulator requires less 
space and weight than the chokes and capacitors which 
would be required to reduce the ripple an equivalent 
amount. 


CONE I DENT! AT 






































































































PART III 


PRECIPITATION STATIC PROBLEMS 


W ith the inauguration of long-range military 
flights incident to the problems of national de¬ 
fense and later of actual warfare, which had to be 
carried out in areas of prevailing bad or adverse 
weather conditions such as Alaska, Northwestern 
United States, etc., the radio interference phenomenon 
known as precipitation static became of primary con¬ 
cern and Division 13 undertook the prosecution of a 
general study. Precipitation static is caused by high- 
voltage electric discharges from planes due to the 
accumulation of charges picked up by flying through 
snow, rain, or dust. These accumulations raise the 
airplane to a potential which will break down the 
insulating ability of the surrounding atmosphere. 
When encountered, the interference caused usually 
entirely disrupts radio communications. 

Several contracts were entered into by Division 13 
on various phases of the project. It was soon found 


that military operation of aircraft at much higher 
altitudes and at much greater speeds resulted in 
greatly increased interference which could not be com¬ 
bated effectively by the means used by the commercial 
air transport operators. Further investigation with the 
‘hope of better solutions was required. Furthermore 
the hazard of lightning strikes to aircraft in flight 
constitutes a problem closely related to precipitation 
static and was made a subject for concurrent inves¬ 
tigation. 

Several of the studies were conducted entirely in 
the laboratory, while others employed both laboratory 
techniques and flight tests to examine the usefulness 
of the instruments developed and flight-tested. The 
progress of the project as a whole was delayed for 
some time due to the unavailability of aircraft having 
the required size and speed characteristics for carry¬ 
ing forward the flight-test program. 


CONFIDENTIAL 


45 
















Chapter 7 

PRECIPITATION STATIC REDUCTION 


Investigation of the fields around point-discharge sources; 
effects of air movement and radiation on point discharge; 
shock excitation of radio receivers; noise reduction by non¬ 
linear devices; charge dissipators, such as fine wires, the 
Bendix discharger, radioactive devices, and flame dischargers. 

INTRODUCTION 

U nder this project" several aspects of the precipi¬ 
tation static problem were studied. 1 Among these 
studies are the mechanism of high-voltage electric 
discharges, the approximate equivalent circuit for 
studying voltage-induction effects arising from corona 
discharges, the statistical current-frequency spectrum 
of the positive streamer discharge and the determina¬ 
tion of the corresponding current-burst shape, deter¬ 
mination of pulse rates for negative-point discharges, 
radio interferences to be expected in complex networks, 
effect of receiver detectors on measured interference, 
shock-excitation effects and the relationship between 
effective band width and circuit decrement, and noise 
reduction secured by means of limiters, dampers, and 
other nonlinear circuit elements. 


711 Apparatus Involved 

To supply the high voltages needed to produce elec¬ 
tric discharges at atmospheric pressure, rectifier-filter 
arrangements were provided producing 200,000 volts 
and discharge currents of 0.005 to 5,000 ju.a. Means 
were also provided for superimposing on the d-c po¬ 
tentials an oscillating voltage of several thousand 
volts and of any frequency between 100 and 4,000 
cycles per second. Vacuum-tube voltmeters were em¬ 
ployed for measuring interference. 

712 Preliminary Investigations 

Before studying actual means for reducing precipi¬ 
tation static noises, an extended study was made of 
various types of noise sources, that is, negative and 
positive points, the discharges from blunt points, and 
streamers and coronas. 

“Project C-21, Contract OEMsr-92, Oregon State College. 


Among other interesting facts discovered was that 
carrier-like bands of intensified noise existed at cer¬ 
tain radio frequencies. Furthermore, it was observed 
that for a particular operating condition the frequency 
intervals between all these bands were more or less 
the same. Thus it appeared that radio interference 
was being produced by a regularly recurring phenom¬ 
enon having a frequency of occurrence equal to the 
frequency difference between successive noise bands. 
For example, with a discharge of 52 /j.a from a nega¬ 
tive needle point, strong interference was experienced 
at 2, 4, 6, 8, 10, and 12 me, the noise voltage decreas¬ 
ing but the band width increasing with frequency. A 
linear curve could be drawn between discharge current 
and frequency in megabursts per second indicating the 
direct relationship. 

Interposing a liigh-Q inductor at the end of a trail¬ 
ing wire antenna was suggested as a means of reduc¬ 
ing static over a narrow band of frequencies. A resistor 
at the end of the trailed wire resulted in general noise 
reduction but not so much as was secured over the 
narrow band by the inductor. 

713 Nonlinear Elements 

The conclusion of the investigators was that the only 
hope of greatly improving radio reception through 
modification of the receiver circuits alone during pe¬ 
riods of precipitation static lay in the use of non¬ 
linear circuits or devices. 

A Hallicrafter receiver with an audio limiter was 
found to be less susceptible to noise of the type ex¬ 
amined than a receiver not so equipped. Analysis in¬ 
dicated that antenna limiters would probably not pro¬ 
duce much benefit. 

The fact that shock excitation produces long drawn- 
out, oscillations in the various stages of a radio receiver 
led the investigators to believe that “dampers” of 
various sorts, which restrict the oscillations in the first 
i-f circuit before they have time to set the following 
circuits into shock oscillation, might prove to be very 
useful. Even though an improvement in a signal-to- 
noise ratio of only several times is obtainable by means 
of a damper alone, it is entirely possible that dampers 


CONFIDENTIAL 


47 






48 


PRECIPITATION STATIC REDUCTION 


used in conjunction with antenna, i-f and audio limit¬ 
ers, and other devices could effect a very material 
improvement in radio reception during adverse cir¬ 
cumstances. 

7-2 STUDY OF EXISTING CHARGE 
DISSIPATORS 

A study was made of several types of devices, each 
of which had as its function the discharge of accu¬ 
mulated charge on an airplane without at the same 
time producing radio interference. These devices were 
of self-ionized and pre-ionized types. The former 
were simply discharge wires or other structures in 
which the physical or mechanical construction was 
varied in the hope of attaining better discharge with 
less accompanying noise. Examples of pre-ionized 


types are radioactive cups and a flame discharger. 

Point and fine-wire dischargers of several sorts were 
studied and compared to the conventional Bendix 
unit. 

The conclusion reached was that the only kind 
capable of discharging large currents in a rapidly 
moving airstream with absolutely no measurable 
radio interference is the pre-ionized flame discharger. 
It consists of a high-temperature oxyacetylene flame 
at the end of a long, slender conducting electrode. 

Another conclusion of the investigation was that 
the noise-current frequency spectrum obtaining quite 
generally for most gaseous discharges irrespective of 
pressure and polarity seems to be fairly constant up 
to roughly 5 me and to vary inversely as the square 
of the frequency at higher frequencies. 




Chapter 8 

PRECIPITATION STATIC RESEARCH 


Use of inverse vacuum-tube voltmeter for measuring high 
potentials and application to study of precipitation static. 

s i THE PROBLEM 

T he purpose of this project a was to attack the pre¬ 
cipitation static problem 1 by developing test equip¬ 
ment, making flights in storms, correlating the data 
secured and, if possible, develop means of eliminating 
the interference. Because a suitable airplane with 
which to collect systematic records under storm con¬ 
ditions was not available, the main part of the pro¬ 
gram could not be completed. The test equipment, 
however, was designed, built, and tested. 

8.2 EXPERIMENTAL PROCEDURE 

It was believed that an experimental investigation 
of the electric charges on and the electric potential 
gradients around airplanes flying through snow, rain, 
or sand storms would provide a basis for designing 
equipment to reduce communication failures in such 
storm conditions. 

The first step in the investigation was the develop¬ 
ment of a generating voltmeter, together with suitable 
indicating and recording equipment, particularly 
adapted to the measurement of potential gradients at 
the surface of an airplane. 

8.3 APPARATUS DEVELOPED 

The apparatus developed consisted of wind-driven 
generating voltmeters which generated voltages pro¬ 
portional to the surface potential gradient at particu¬ 
lar positions on the plane, vacuum-tube voltmeters for 
measuring the generated voltages, an instrument panel 
for mounting the microammeters for the several 
vacuum-tube voltmeters and standard aircraft instru¬ 
ments, a panel-mounted instrument for measuring 
vertical accelerations, and a motion-picture camera 
and drive for photographing the instrument panel. 

The generating voltmeter was of conventional de¬ 
sign and needs no description. After considerable ex¬ 
perience with conventional vacuum-tube voltmeters 

‘Project C-41, Contract OEMsr-678, University of New 
Mexico. 


with their positive-ion troubles, an inverse vacuum- 
tube voltmeter circuit was developed. 

831 Inverse Vacuum-Tube Voltmeter 

As is well known, the inverse vacuum-tube voltmeter 
employs a high-vacuum tube in a reverse connection, 
that is, the input signals are applied to the plate cir¬ 
cuit, the grid current being a measure of the applied 
potentials. The plate is biased negatively so that 
positive voltages applied to the grid result in greater 
grid current flow. The circuit for the voltmeters finally 
employed is shown in Figure 1. The input impedance 
of this circuit (neglecting insulation resistance) is of 
the order of 5,000 megohms. The 9002 tube was chosen 
after some experimenting, although the tubes must 
be individually selected for the job. Another suitable 
tube of small size is the 6C4. 


9002 



Figure 1. Circuit diagram of inverse vacuum-tube voltmeter. 

In operation, SW 3 and R± are adjusted to give mid¬ 
scale reading on the microammeter with no voltages 
applied to the input. R 3 is used to adjust all tubes 
and instruments to the same sensitivity. The range of 
the instrument is changed by closing SW 3 and adjust¬ 
ing R 4 again to give the mid-scale reading. Calibra¬ 
tion curves are shown in Figures 2 and 3. 


49 


CONFIDENTIAL 




















50 


PRECIPITATION STATIC RESEARCH 



VOLTS PER CM 

Figure 2. Calibration of voltmeter, low range. 

832 Camera Drive 

It was desired to photograph the entire instrument 
panel, containing five microammeters, air temperature, 
airspeed, and other essential data, at rates of 1 frame 
per second and 1 frame per 5 seconds. A Paillard 
Bolex 16-mm camera was employed, with a motor 
drive and gear box with shifting gears mounted di¬ 
rectly to the small motor. 

83 3 Other Equipment 

The g-meter for measuring vertical acceleration 
consisted simply of a magnetically damped arm cen¬ 
tered on two springs with a scale calibrated directly 
in g. 

An all-metal plane capable of swift ascent and de¬ 
scent, equipped with two-way radio, de-icing equip¬ 


ment, blind-flying instruments, and oxygen for high- 
altitude work was desirable. A Consolidated B-24 
stationed at the Alamogordo Air Base was made avail¬ 
able and the instruments were installed. A short 
flight was made when weather conditions were appro¬ 
priate and the plane returned to base. The total time 
devoted to gathering data by means of the plane 
amounted to 3 hours. The plane was then wrecked 
on a routine military mission not connected with the 
project. 

The brief experience with the instruments and the 
plane indicated that the techniques developed were 
satisfactory and that if another plane could have been 
secured data of worthwhile nature could have been 
developed. 



Figure 3. Calibration of voltmeter, high range. 






















Chapter 9 

EFFECT OF AIRCRAFT SURFACE TREATMENT 


Studies and tests yielding quantitative data upon the relation¬ 
ship between surface treatment of aircraft skin materials, 
specifically with military paint finishes, and the accumulation 
of electric charges known as precipitation static. 

9-1 EQUIPMENT AND METHODS OF TEST 

E quipment foe blowing snow or frost crystals over 
sample surfaces and measuring the charging rate 
was developed during 1938 under a cooperative proj¬ 
ect between the Bendix Aviation Corporation and the 
Purdue Research Foundation. This equipment was 
used in the work done under Project C-64. a 

The test equipment 1 consisted of a vacuum-cleaner 
motor and fan which supplied the air blast, a hopper 
with motor-driven screw-feed mechanism for feeding 
the snow into the air blast at a constant rate, an in¬ 
sulating support for the test sample, a flexible-tube 
connection for returning the snow to the upper part 
of the hopper where it is separated from the air stream 
by centrifugal action, and a portable galvanometer 
for measuring the charging current from the test 
sample to ground. The support for the test sample was 
semicircular in shape and made of paraffined maple 
IV 2 in- thick. A recess % 6 in. deep and D/i in. wide was 
turned in the edge of the block to take the test sam¬ 
ples, which were VA in. wide and 12 in. long. The 
radius of the bottom of this recess was 3% in. In the 
center of this recess a rectangular groove 1 in. wide 
and V 2 in. deep was turned to form a passageway for 
the snow-laden air along the inner surface of the 
sample. The test sample was held in place by an in¬ 
sulated wire spring which ran lengthwise along its 
outer surface. 

Since the test sample was bent to form the arc of 
a circle having a radius of 3% in. and the snow-laden 
air was blown along its inner surface, the snow was 
held in contact with the entire length of the sample 
by centrifugal force. 

The snow was fed into the air stream by the motor- 
driven screw feed at the rate of 30 cc of loose unpacked 
snow per second. 

a Project C-64, Contract OEMsr-679, Purdue University. This 
work led to a much more extensive investigation for the Special 
Devices Branch, Aircraft Radio Laboratory, W right field, 
Contract W-33-038-ac-19 under the title Precipitation Static 
Tests of Surface Coverings for Aircraft. 


The velocity of the air stream along the sample was 
estimated by measuring the air speed in a straight 
extension of the rectangular curved passage by means 
of a standard Pitot tube and was found to be approxi¬ 
mately 65 mph. 

At an air speed of 65 mph and a feed rate of 30 cc 
per second, the average depth of snow on the sample 
was less than 0.05 mm, which permitted ample oppor¬ 
tunity for the snow particles to come in contact with 
the test sample. 

The galvanometer used to measure the charging 
rate was a Leeds & Northrup portable type having a 
deflection sensitivity of 57 divisions per p. a. One ter¬ 
minal of the galvanometer was connected to the 
ground and to the metal part of the hopper containing 
the screw feed mechanism and to the fan and motor 
frames; the other was connected to the back side of 
the sample by means of a connection to the spring 
which holds the sample in place. 

The Aircraft Radio Laboratory supplied samples 
of the following camouflage paints from which test 
specimens were prepared: 

Dark O.D. No. 41 Red No. 45 

Light O.D. No. 42 (green) White No. 46 

Gray No. 43 Blue No. 47 

Black No. 44 Yellow No. 48 

The test specimens were prepared by spray-paint¬ 
ing a single coat of each paint on a carefully cleaned 
strip of Alclad skin-metal. To determine the effect 
of the metal under the paint, the Dark O.D. No. 41 
paint was sprayed on aluminum, copper, Dowmetal F 
cleaned, Dowmetal F with the oxide left on, Duralu¬ 
min and cold-rolled steel. The spray gun was carefully 
cleaned each time before changing to a different paint 
sample. 

Later the Naval Research Laboratory supplied 
samples already coated on 1 *4x12-in. strips of Dura¬ 
lumin. One sample of each of the following was 
provided. 


Acetobutyrate, clear 

2 coats 

L-12 Gray 

2 coats 

M-485 Gray 

2 coats 

71 Line Gray 

1 coat 

Dope Red 

2 coats 


To provide information regarding the effect of tem¬ 
perature on the charging rate when using snow or 


CONFIDENTIAL 


51 





52 


EFFECT OF AIRCRAFT SURFACE TREATMENT 


frost crystals, it was proposed to make tests at ap¬ 
proximately 20, 0, and —20 F. Since previous tests 
had shown that the same results were to be expected 
from the use of frost crystals from the refrigerator 
coils as from snow, frost crystals were used until snow 
became available. 


92 SUMMARY OF RESULTS 

The results of a large number of tests on camou¬ 
flage paints and metals, with and without special sur¬ 
face treatment, have led to the following tentative 
conclusions: 

1. A large percentage of the charging takes place 
at or near the point of initial impact of uncharged 
snow or ice crystals with the surface, indicating that 
the charging is due to the contact and separation of 
substances having dissimilar surface characteristics. 

2. All of the camouflage paints became negatively 
charged under most conditions of test and usually 
charged at rates of 1.25 to 5 times that of aluminum. 
This indicates that they fall below aluminum in the 
triboelectric series. 2 

3. Metallic lead was the only material found which 
always became positively charged with snow and frost 
crystals, indicating that snow and frost crystals fall 
between lead and aluminum in the triboelectric series. 

4. The charging rate obtained with snow varies 
widely, depending on the conditions under which the 
snow was formed and its subsequent history. 

5. The charging rate of a paint appears to be af¬ 
fected very little by the metal under it. 

6 . In general the charging rates of the paints de¬ 
creased with increase in temperature over the range 
from —10 to 20 F and fell to zero at some point be¬ 
tween 20 and 32 F. 

7. Dowmetal F that has just been cleaned with steel 
wool may charge either positively or negatively. How¬ 


ever, if the surface is covered with a heavy coat of the 
gray oxide, it usually charges negatively at a low 
rate. 

8 . Oxidized lead initially charges negatively with 
frost crystals, but the oxide wears away rapidly and 
the charge reverses in sign. 

9. The test results indicate that a large percentage 
of the charge acquired by an airplane flying through 
snow or ice crystals is generated on the frontal sur¬ 
faces and on the propellers near the points of initial 
impact of the particles with the plane. 

10. Under the predominant conditions where a 
plane flying through snow or ice crystals acquires a 
negative charge, it should be possible to neutralize 
largely the charging from the camouflage paints by 
metal-spraying a thin film of metallic lead on about 
half or two-thirds of the frontal surfaces, or by coat¬ 
ing the propellers with metallic lead. 

93 FURTHER INVESTIGATIONS DESIRABLE 

The results of the work on this project indicate that 
the following further investigations are desirable. 

1. Investigation should be made to find or develop 
a suitable binder for camouflage paints which would 
fall above aluminum in the triboelectric series when 
subjected to the same conditions of temperature and 
humidity that are encountered by a plane in flight. 

2. A more extensive investigation should be made 
of the charging rate at various points along a surface 
corresponding to a wing section, in an effort to find 
means of causing the snow or ice particles to discharge 
back into the plane before being carried away by the 
air stream. 

3. Since it is apparent from tests with segmented 
test samples that most of the charging occurs at or 
near the point of initial impact of the particles with 
the surface, the charging rate of the semiconducting 
rubber of the de-icers should be investigated. 





Chapter 10 

NOISE ELIMINATOR TESTS 


Investigation of the manner in which an airplane collects 
a charge from precipitation, testing and improvement of the 
effectiveness as noise eliminators of several discharging sys¬ 
tems, design and development of a simple form of voltage 
gradient indicator to be used as a lightning strike warning 
device. 

ioi INTRODUCTION 

A lthough a number of the factors causing precip- 
. itation static and means for reducing the effect on 
radio communication systems were known before this 
project 3 was begun, one of the major accomplishments 
of the project was to collect in one report data collected 
during actual flight in precipitation static areas. 

a Project C5-68, Contract OEMsr-893, United Air Lines. 


In brief, the project 1 proved that the use of high- 
voltage or high-frequency devices as a means for aid¬ 
ing in discharging the plane while in flight would be 
no more effective than an unenergized point dis¬ 
charger. The flight tests proved that high gradient 
fields can be detected and their relative position and 
strength indicated during flight with simple measur¬ 
ing instruments and exploratory prods. One of the 
conclusions reached during the investigation was that 
a painted surface of an airplane collects charge at a 
higher rate than a clean surface. This fact was sub¬ 
stantiated by other investigators and was instrumental 
in influencing the Military to remove paint from all 
noncombatant airplanes. (See Chapter 9.) 



NOISE METER ANTENNA 


END OF MAST WRAPPED WITH TAPE a C0VERE0 
/ WITH GRAPHITE PAINT 


RIGHT ACCUMULATOR-v 


^rLEFT ACCUMULATOR 


NOSE TEST 
AREAS ' 


WING TEST 
AREA 




Figure 1. Boeing 247-D all-metal plane used in precipitation static research. 


CONF1DENTI Alf;, 

.* . • v.H v \ 


53 








54 


NOISE ELIMINATOR TESTS 


102 APPARATUS EMPLOYED 

A major portion of the experimental work was con¬ 
ducted in a Boeing 247-D all-metal plane (Figure 1). 
Some auxiliary testing was carried out with a Douglas 
C-47 military cargo airplane and other tests were 
made in a Consolidated C-87 cargo version of the 
Liberator bomber. 

Test areas insulated from the airplane were faired 
into the surface so they would not interfere with the 
normal flow of air. These test areas were intended to 
show how charge is accumulated and what portion of 
the plan structure is responsible for the greatest ac¬ 
cumulation. The propellers were equipped with full- 
length de-icer shoes and fine bare copper wires were 
cemented in the leading edge of these rubber shoes. A 
slip ring was mounted on the propeller hub so that 
connections could be made to these fine wires. A sec¬ 
ond slip ring was connected to the propeller itself. 

Spurious corona discharges from uncontrolled 
points about the airplane were believed to be respon¬ 
sible for much of the radio noise. To prevent this un¬ 
controlled factor from entering the results of the in¬ 
vestigations on the several discharge devices, all un¬ 
shielded and sharp projections about the airplane were 
covered in some manner, usually by a generous appli¬ 


cation of rubber tape covered with a coating of con¬ 
ducting graphite paint. Antenna insulators, masts, 
and tension springs were all treated in this fashion. 
Filters were placed in the antennas so that the static 
current discharged by them could be measured with¬ 
out interrupting radio communication. 

Trailing wire dischargers were fitted to the wing 
tips, stabilizer tips, and tail position. These wires 
were retractable. 

Gradient indicators consisting of small point-ter¬ 
minated dipole explorers were located parallel to the 
major axis of the plane so that components of the 
electric field could be measured in all directions simul¬ 
taneously. The generating voltmeter discussed in 
Chapter 8, consisting of a rotating vane and ampli¬ 
fier, was mounted on an inspection door on the under 
surface of the right wing. The location of the search 
equipment is shown in Figure 2. 

io.3 INSTRUMENTATION 

Several methods of measuring the small currents 
encountered in precipitation static were employed. 
In one case, an RCA fiOl neon tube shunted by a 
capacitor flashed at a rate proportional to the cur¬ 
rent flowing (Figure 3C). This method was simple 


mm in i|H * 'WWHWjBIMMK 



ACCUMULATOR 






TRANSMITTING ANTENNA’ 



. GRADIENT DETECTOR 


DISCHARGER 


FRONT"V"ANTENNA 


TRAILING WIRES 
(RETRACTED) 




REAR V ANTENNA 








w 




- 






Jr 

" 




,v 


Figure 2. Location of research equipment external to cabin. 






TEST RESULTS 


55 


A 



TEST PRODS 


BALANCED VACUUM TUBE 
AMPLIFIER 


CROSS POINTER 
INSTRUMENT 




Figure 3. Electric and electronic measuring instruments employed in United Air Lines research. 


and economical but proved unreliable at currents of 
V 2 p,a or less. In other cases a single-stage amplifier 
with a plate milliammeter (vacuum-tube voltmeter) 
was found to be quite successful. Currents of 0.05 p.a 
per scale division were easily measured. Twenty-four 
of these amplifiers were built on a single chassis. 

10 - 31 Charge Indicator 

A basic requirement for the analysis of data obtained 
was a knowledge of the static potentialities of the 
various weather areas encountered. For this purpose 
a charge indicator was constructed and used on all 
flights. This instrument consisted of an accumulator 
element streamlined in form and insulated from the 
plane. It collected its charge from precipitation in the 
same manner as did the plane. That is, certain por¬ 
tions were subject to impact by snow, ice, or rain; 
other portions were subject to frictional effects, and 
other portions of the surface were subject to charge 
by separation. Charge rates were measured by noting 
the current flowing from collector to the airplane. 

m 


Once the airplane had acquired a high potential with 
respect to space, discharge would occur, reducing the 
plane potential with respect to the accumulator ele¬ 
ment so that the charge on this element would flow 
to the airplane. 

Provisions were made so that the information on the 
several current meters could be photographed simul¬ 
taneously by means of a single-frame 16-mm camera. 

io-4 TEST RESULTS 

Studies were made of several types of dischargers, 
all compared to a trailing wire consisting of a 5-ft 
length of very fine wire connected to a suppressor re¬ 
sistor made of approximately 5 ft of rubber-covered 
Aquadag-impregnated rope, the resistance of which 
was from 250,000 to 500,000 ohms. This trailing wire 
was definitely helpful in reducing static noise. Under 
severe conditions, currents as high as 200 to 400 
were observed, but in each case the trailing wire low¬ 
ered the noise level appreciably. 


CONFIDENTIAL 























56 


NOISE ELIMINATOR TESTS 


Other trailing wires of various types, such as those 
with terminals made up of a brass sphere or a cluster 
of three very sharp points, were used, but none seemed 
to have any advantages over the simple wire described 
above. 

A device of the Slayter Electronics Corporation 
consisting of a high-voltage supply and a discharge 
element made up of sharp points or long conducting 
rods was tested in flight. Both units were intended to 
produce ionization about the discharge points and thus 
to aid in discharging the plane. 

In test flights, it was demonstrated that the Slayter 
apparatus “produced no improvement in radio recep¬ 
tion, while the trailing wire discharger rendered the 
receiving equipment usable although all the static was 
not removed.” 1 

After many flights with several models of discharg¬ 
ers, including wick devices and those which are ener¬ 
gized by high-voltage 60-cycle or r-f power, the con¬ 
clusion was reached that random noises must be con¬ 
trolled by shielding to reduce the potential gradients 
at the points producing the noises; in this manner 
more current would be discharged by the controlled 


discharging system which must be capable of dis¬ 
charging current in the order of 200 to 300 /*a. 

Flight tests showed that the majority of the charge 
accumulated by an airplane comes from the frontal 
surfaces. The amount of accumulation is a function 
of speed and surface coating. A painted surface will 
collect greater charges than will an unpainted surface. 

Test flights indicated that the gradient indicator, 
consisting of prods attached to a vacuum-tube volt¬ 
meter, would indicate the presence of thunderclouds 
when they were several miles distant. In general, an 
instrument capable of registering 5 to 10 /xa would 
mark the existence of a thundercloud five miles away, 
this current increasing to 30 to 50 f*& when the plane 
approached within a half mile of the cloud. 

Using four prods, one each placed in wing tips, 
tail, and nose parallel to the longitudinal and lateral 
axis of the plane, and with the indicator taking the 
form of a cross-pointer instrument (Figure 3A) 
marked to show the direction toward the danger area 
and the safe limit of field intensity to avoid danger of 
lightning strike, warning of a possible lightning strike 
was found to be distinctly feasible. 




Chapter 11 

THE BLOCK-AND-SQUIRTER SYSTEM 


Study of the order of magnitude of corona discharge cur¬ 
rents from large military aircraft; flight tests to investigate 
the efficacy of various discharge systems; development of the 
“block-and-squirter” system in which the plane is discharged 
in pulses between which the radio receiver is operative, the 
system relying on the phenomenon of persistence of hearing 
by which an operator takes no notice of short intervals of 
silence in reception. 

hi INTRODUCTION 

recipitation static radio interference was con¬ 
sidered to be one of the most serious hazards en¬ 
countered in the training of Army pilots on large 
military aircraft. For this reason, the Commanding 
Officer of the Second Air Force assigned one of his 
staff planes, a B-24 Liberator four-engined bomber, 
and later a B-17 Flying Fortress, for use by Washing¬ 
ton State College [WSC] in conducting research 
studies of precipitation static. 8. 

Prior to undertaking this research with the Second 
Air Force and with OSRD, WSC had done consider¬ 
able research over a period of some five years in the 
field of radio static interference on commercial air¬ 
craft. With the cooperation of United Air Lines, 
studies had been made of receiver performance and 
of balancing networks for reducing static shocks. 
Therefore the work on this project 1 was, in effect, a 
continuation and enlargement of the work already 
underway. 

112 CAUSES OF PRECIPITATION STATIC 

The most serious type of aircraft radio interference 
occurs when the plane flies through precipitation, such 
as rain, snow, or frost crystals, or through dust-laden 
air. This kind of interference is especially serious be¬ 
cause it occurs when radio aid is most needed, that 
is, during times of limited or zero visibility. 

Various theories have been advanced as to the 
mechanics by which an electric charge is generated on 
a plane in flight, and how it is collected and dis¬ 
charged. The cause of radio interference from precipi¬ 
tation static is generally understood to be due to coro¬ 
na discharges taking place between various parts of 


‘Project C-70, Contract No. OEMsr-848, Washington State 
College. 


the plane and air. These commonly take place from 
sharp metal corners and edges of wings and ailerons, 
from antenna wires, or from any sharp or pointed 
metal part of the plane exposed to the outside air, 
and probably also from the sharp edges of propellers. 

There appear to be two general conditions under 
which a plane accumulates an electrostatic charge 
which causes radio interference. In the first case, the 
plane flies through a highly charged area such as a 
thundercloud in which there exists a high space gra¬ 
dient in potential, and the metal plane seems to act 
more or less as a conductor between differently charged 
areas. In such areas, measured charging currents on 
the moving plane usually change rapidly in value and 
polarity. In the second case, the plane appears to col¬ 
lect a potential charge at a more or less uniform rate 
and to discharge it at about the same rate. Under 
these conditions, the plane is usually negative with 
respect to any discharge which takes place. This sec¬ 
ond condition prevails more frequently than the first 
and therefore is considered the more serious hazard 
to flying, although both conditions create radio inter¬ 
ference in the receiver on the plane. 

A secondary source of radio interference arises be¬ 
cause the propeller surfaces engage the air at much 
higher speed than do the other surfaces of the airplane 
so that whether the accumulation of an electrostatic 
charge is due to impact, friction, or sweeping action, 
it is probable that a potential difference exists be¬ 
tween the propeller and the plane. Interference from 
this current flow through the propeller bearings would 
closely resemble “wheel static” in automobiles and 
could be corrected by providing an electrically con¬ 
ducting path around the oil film. Hydraulically op¬ 
erated propellers have no ground brush but electrically 
operated propellers do have a good ground brush and 
would not cause this type of interference. 

Radio interference occurs on all frequencies com¬ 
monly used on the plane, although not always to the 
same degree. However, under severe static conditions 
all radio contact is interrupted. It is agreed that the 
radio-beacon frequencies, 200 to 400 kc, are the most 
important to the pilot needing radio aid, and therefore 
all observations during these tests were made on this 
band. 






58 


THE BLOCK-AND-SQUIRTER SYSTEM 


113 PURPOSE OF THE PROJECT 

The expressed purpose of Project C-70 was to make 
flight tests of precipitation-static radio-interference 
phenomena, and of devices for their control. First, it 
was desired to measure the magnitude and charac¬ 
teristics of corona discharge currents which cause 
radio interference on large military aircraft. Before 
intelligent steps can be taken to suppress, neutralize, 
or otherwise control radio interference on these planes, 
it is necessary to have more information than was 
available on the subject. Measurements on four- 
engined bombers had not been undertaken prior to 
this contract. 

Second, it was desired to flight-test such devices as 
might be developed by WSC or by other contractors 
and having for their purpose the suppression or con¬ 
trol of precipitation static interference on large air¬ 
craft. Facilities were especially favorable for such 
tests because of prior arrangements with the Second 
Air Force where planes and experienced instrument 
pilots were made available for flying in inclement 
weather. 

The work naturally fell into three divisions: devel¬ 
opment of equipment for making the desired measure¬ 
ments in flight, organizing and conducting test flights 
in weather favorable to static interference, and carry¬ 
ing on supplemental laboratory tests of equipment 
and devices which had promise of reducing the in¬ 
terference. 

n.4 FLIGHT TESTS 

Flights were made to determine the magnitude of 
discharge currents which take place on large military 
aircraft, to ascertain the rate at which a charge ac¬ 
cumulates on a plane, and to test devices developed for 
producing a noiseless discharge, or for suppression 
or control of the discharge. Most of the flights were 
made in the Pacific Northwest, although some thirty- 
three states were crossed during the course of the 
work. Test data were secured at altitudes ranging 
from 3,500 to 32,000 ft, over mountains as well as 
plains, in rain, dust, snow storms, and through frost 
crystals, at temperatures from —J—20 C to below —40 C. 

11.5 TEST equipment 

The essential measuring equipment consisted of 
multiband radio receivers, a radio noise meter, d-c 
microampere amplifiers, and Esterline-Angus record¬ 
ing milliammeters. A standard 6K7 amplifier tube 


operated as a d-c amplifier increased the small cur¬ 
rents obtained from the test devices to a value where 
milliamperes could be recorded on the recording mil- 
liammeter. 

To measure the rate at which the plane accumulated 
an electrostatic charge, a standard radio-compass loop 
housing was copper-plated and mounted on a spar in 
free air in front of the nose of the plane. This charge 
collector, having a projected area in the line of flight 
of exactly 0.45 sq ft, was grounded to the plane 
through a 15-megohm resistor. The current through 
this resistor was measured by a d-c amplifier and rep¬ 
resented 45 per cent of the rate per sq ft at which 
the wings, elevators, etc., collect electrostatic charges. 
(See Figure 1.) 


Figure 1 . Copper-plated teardrop automatic direction 
finder housing mounted on wooden spar and adjusted in 
position in front of nose of plane. Wire through center of 
spar connects teardrop to d-c amplifier to measure rate 
of accumulation of charge on plane. 

Continuous records of all of the test instruments 
were made during flight. One recording meter regis¬ 
tered the charging rate, another the radio signal and 
interference noise, and the other two of the four 
meters were employed to test the several types of dis¬ 
charge devices tested. The final report 1 includes 220 
sheets of flight data collected by means of this equip¬ 
ment. The data include discharge rates obtained from 
Bendix wires, trailing antennas, rod dischargers, 
bristle devices, etc. In all, 23 different types of dis¬ 
charge systems were examined. 

H6 STUDY AND ANALYSIS OF 
FLIGHT TEST DATA 

1. Many flight tests were made on B-17’s and B-24’s 
in rain, snow, and frost crystals, and discharge cur- 




jCONFI DEMT1AL 








THE BLOCK-AND-SQUIRTER SYSTEM 


59 


rents were measured. These range from the usual 25 
to 100 /x a up to 500 or 600 /x a in more severe cases. 

2. When flying through a local thundershower, dis¬ 
charge currents were measured in excess of 2,500 /xa. 
Polarity of the charge on the plane reversed when fly¬ 
ing through a thundercloud. 

3. The polarity of the plane was usually negative 
with respect to the discharge wires extended from the 
tail of the plane. 

4. Padio noise usually started when the discharge 
current on the Bendix wire reached 35 to 100 /xa. 

5. The most severe radio noise was always encoun¬ 
tered when flying through frost crystals. 

6. The use of multiple discharge wires did not as¬ 
sure noiseless radio reception at high discharge cur¬ 
rents. 

7. The application of high-voltage alternating cur¬ 
rent to accelerate corona discharge did not reduce 
static interference. 

8. The application of high voltage or high fre¬ 
quency to the receiving antenna did not aid radio re¬ 
ception. 

9. An increase in discharge current accompanies an 
increase in airspeed of the plane. 

10. The cadence in static noise (“wow” static) 
noted on some of these flights appears to be associated 
with the difference in speed of adjacent propellers. 

11. The trailing antenna wire will discharge cur¬ 
rent approximately in proportion to its length and is 
no less noisy than a Bendix wire. 

12. The discharge characteristics of the trailing an¬ 
tenna can be improved by adding a tuft of small wire 
bristles to the antenna weight. 

13. A 100-ft Bendix wire will discharge approxi¬ 
mately in proportion to its length and was not found 
to be any less noisy than a 10-ft Bendix wire. 

The upshot of all this work indicated that: 

1. The magnitude of corona discharge current on 
a four-engined bomber appears to be too large to be 
dissipated noiselessly and at the same time keep un¬ 
controlled discharges from taking place elsewhere on 
the plane. 

2. Higher-speed planes with larger frontal areas 
will undoubtedly experience even larger corona dis¬ 
charges than take place on the B-17’s and B-24’s. 

3. Properly applied a-c potentials to a suitable dis¬ 
charger should keep the charge potential on the plane 
low enough to prevent uncontrolled discharges from 
taking place. 


THE BLOCK-AND-SQUIRTER SYSTEM 

Attempts to produce noiseless discharge as a means 
of solving the static problem having failed, attention 
was turned to another approach to the problem. Since 
persistence of hearing, like persistence of vision, takes 
no notice of small intervals of silence in reception, it 
was thought that if the discharge on the plane could 
be alternately blocked and accelerated and if the re¬ 
ceiver could be alternately opened and blocked in syn¬ 
chronism, then radio reception might be possible dur¬ 
ing the worst kinds of static troubles. 

The method of putting such a method into opera¬ 
tion is as follows: 

A suitable a-c potential is applied to a corona dis¬ 
charger system on the plane, depleting the electrostatic 
charge sufficiently during one half cycle so that during 
at least part of the opposite half, when the applied 
potential blocks the discharge current, the effective 
corona voltage on the entire plane will be low enough 
so that no corona discharge will take place from any 
other part of the plane. 

During the quiet, or blocking, part of the cycle the 
radio receiver is turned on and accepts a clear signal. 
During the squirter, or noisy, part of the cycle, the 
radio receiver is turned off and hence it is not cogni¬ 
zant of the severe noise accompanying the discharge 
of the plane. 


BLOCKING 



Figure 2. Pulsing and blocking, or “static squirter” system. 


The receiver valve is interconnected with the a-c 
generator supplying the squirter system and properly 
phased therewith so that the receiver is always turned 
off during the noise discharge. (See Figure 2.) The 
frequency of the a-c generator should be above the 
audio range of the receiver, perhaps between 10 and 
15 kc. 
























60 


THE BLOCK-AND-SQUIRTER SYSTEM 


The metal plane constitutes an effective electrostatic 
capacitance which various authorities have estimated 
at from 500 to 2,000 /x/xf. The electrostatic charge is 
accumulated at a measurable rate while flying through 
precipitation, the rate depending upon conditions. 

For a given set of conditions, the corona discharge 
from some point on the surface of a plane in flight 
will follow the equation: 

I = KE(E — E s ) 

in which I is current, E is the total effective voltage 
applied to cause I , and E s is the potential at which 
corona starts. K depends upon the geometry of the dis¬ 
charge point, and upon various other conditions, such 



Figure 3. Typical performance curve of sharp cutoff 
tube over-biased to be used as blocking valve. 

as air velocity and pressure, humidity, etc. Experi¬ 
mental curves derived from tests on a Bendix wire 
closely follow this equation. It is evident then that 
with a properly designed corona discharger system and 


with all sharp points and corners removed from the 
plane, the E g of the discharger system can be kept con¬ 
siderably below the E g of any other part of the plane. 

If the effective d-c potential E could be varied 
slowly up or down from some certain value, the dis¬ 
charge current would increase or decrease according 
to the above equation. Practically, this effective corona 
potential E may be varied by superimposing an a-c 
potential and the discharge current thus alternately 
increased and decreased. 

By means of laboratory tests it was proved that, if 
a high enough a-c potential is superimposed on the 
d-c corona, the discharge can be reduced during part 
of the cycle to a noiseless value, or even completely 
stopped. On the other half of the cycle, the discharge 
will be greatly increased. This ejects the electrons 
from the discharger in a succession of squirts. 

Theoretical analysis and some laboratory tests in¬ 
dicated that the effective corona potential for a dis¬ 
charge of 100 /xa from a Bendix wire would not be 
less than 45 to 50 kv; that the a-c power required for 
the block-and-squirter system would be of the order of 
10 watts or less; that the a-c potential should have a 
frequency of the order of 10 to 15 kc. 

ns FINAL CONCLUSIONS 

It was concluded at the close of the contract that 
radio interference on large military aircraft is caused 
by such large corona currents that a successful noise¬ 
less discharge system would not be practical and that 
a system based on the principle of discharging the 
plane in a series of pulses between which the receiver 
is turned on offers considerable promise of success. 

The block-and-squirter system was further devel¬ 
oped and flight-tested with the Second Air Force and 
a prototype unit built by General Electric Company 
for use on B-29’s. The original test equipment weighed 
about 750 lb. The prototype weighs less than 40 lb and 
occupies a very small space in the dorsal fin of the 
plane. Y-J Day came before the prototype was avail¬ 
able for installation on the B-29’s, but the Army ap¬ 
proved continuation of the tests. 


CONFIDENTIAL 



























PART IV 


PANORAMIC RECEPTION 


S everal projects in Division 13 were concerned 
with panoramic receivers, either for monitoring 
purposes or for more active participation in communi¬ 
cation taking place in war zones, i.e., for jamming. 
The following projects dealt with receivers per se: 

C-27, a moving-screen receiver for the region 500 
to 600 me. 

C-36, an improved receiver for the band 3 to 10 me, 
also a similar improved receiver for the band 0.1 to 
30 me. As part of the project, a report, Fundamentals 
of Panoramic Reception, was issued. 

C-39, a scanning and stopping receiver for the 350- 
to 750-mc range. 

Uses of panoramic principles for interference gen¬ 


eration (jamming) are covered in Part V. 

In the summaries of this work to follow, the chrono¬ 
logical order in which the work was done has not been 
followed. Instead, the portion of the C-36 final report 
dealing with fundamentals is reviewed first, then fol¬ 
low the reports on the several receiver projects. The 
summaries of reports on the applications of panoramic 
principles have been confined, largely, to abbreviated 
descriptions of the apparatus developed. Where sig¬ 
nificant statements or quantitative data contained in 
the final reports of the receiver projects relate to 
principles of panoramic reception, they have been 
lifted from the report in which they appear and in¬ 
cluded in Chapter 12. 


CONFIDENTIAL 


61 











Chapter 12 


THE FUNDAMENTALS OF PANORAMIC RECEPTION 


A summary of the parts played in panoramic reception by 
such factors as proper frequency allocations, design of scan¬ 
ning filters on resolution and signal-to-noise ratio, types of 
indicators, determining frequency of received signals, record¬ 
ing signals, reception of pulse signals, receivers without fre¬ 
quency sweep, uses of panoramic receivers, and other factors 
of importance to the general subject. 

12.1 INTRODUCTION 

T his report, based to a large extent upon work car¬ 
ried out in certain NDRC projects, 1,2,3 summarizes 
some of the fundamentals of the art of panoramic- 
radio reception. a By panoramic reception is meant the 
reception of signals present within a band of frequen¬ 
cies and the display on a visual indicator of informa¬ 
tion concerning these signals. The received signals are 
usually radio signals but may be other types. Devices 
in which a variable oscillator is employed to obtain a 
visual diagram or record of the frequency characteris¬ 
tic of an equipment unit or circuit component will not 
be included under the term panoramic receivers. 

There are two general methods of panoramic recep¬ 
tion: (1) that in which the signal frequencies are 
swept in succession past a relatively narrow selecting 
circuit, the output of which is used to actuate the vis¬ 
ual indicator, and (2) that in which the signal fre¬ 
quencies are applied simultaneously to a number of 
selecting circuits of adjacent frequencies, and the out¬ 
puts of these circuits are applied in rapid succession 
to the visual indicator. In this report principal con¬ 
sideration will be given to the former method, only a 
brief section being devoted to the latter. A theoretical 
alternative to (1) is to move the selecting circuit past 
the signals, but the practical utility of this appears to 
be very slight except possibly for special problems. 

The process of moving the frequencies past a select¬ 
ing circuit is commonly termed scanning. It is accom¬ 
plished by automatically varying the frequency of a 
beating oscillator, known as a sweep oscillator. The 
essential elements of a panoramic receiver of the fre¬ 
quency-scanning type are shown in Figure 1. Other 
functions frequently provided in a panoramic receiver, 
in addition to those of scanning and visual indication, 
are aural reception and precise frequency determina¬ 
tion. 

“Project C-36, Contract No. OEMsr-357, Bell Telephone 
Laboratories, Inc., Western Electric Co., Inc. 


The process of determining the different frequencies 
present in a given band by heterodyning the signals 
in succession past a sharp selecting circuit is quite old 
in the art. 4,5 Almost as old is the provision of auto¬ 
matic sweep for the beating oscillator. 6 Panoramic re¬ 
ception adds to this frequency-analyzer technique the 
simultaneous display on a visual indicator of the dif¬ 
ferent signals in the band which is being scanned. 7,8 



Figure 1. Essential elements of panoramic receiver. 


Panoramic reception has much in common with 
ordinary radio reception. Thus in each case (a) pro¬ 
vision must usually be made for handling a wide range 
of signal strengths; (b) high sensitivity is desirable, 
the limiting factor being random electric disturbances 
in the medium or the equipment (conveniently re¬ 
ferred to, even in a visual device, as noise) ; (c) heter¬ 
odyne technique is advantageous to obtain the desired 
gain and selectivity. 

Much more important than the similarities, however, 
are the differences between the two types of reception. 
Some of these are: 

1. Whereas the ear is effectively able to appreciate 
only one signal at a time, the eye can receive at almost 
the same time information concerning a number of 
signals. 

2. In panoramic reception it is necessary to swing 
one or more beating oscillators continuously, and fairly 
rapidly, over a band of frequencies, whereas in ordi¬ 
nary radio reception the beating oscillator frequency 
is adjusted manually and remains fixed during a de¬ 
sired response. 

3. Unless the circuits which precede the scanning 
of the radio band are tuned in synchronism with this 


CONFIDENTIAL 



63 




















64 


THE FUNDAMENTALS OF PANORAMIC RECEPTION 


scanning, these circuits must have a reasonably flat 
response over a frequency band as wide as the scan¬ 
ning band. This is important in the design of anten¬ 
nas, transmission lines, and r-f amplifiers. 

4. A wide range of signal intensities presents 
greater problems in panoramic reception. 

5. Because of the finite build-up and decay times of 
a selecting circuit, the width of the selecting circuit 
desirable for use in the scanning process depends upon 
the rate at which the frequency range is scanned. 
Hence the width of the receiver band is usually con¬ 
siderably greater for a panoramic receiver than for an 
ordinary receiver. 

6. As a result of the various factors noted above, the 
selectivity or resolution obtainable in a panoramic re¬ 
ceiver is poorer. 

7. The signal-to-noise ratio and useful sensitivity 
are poorer than can be attained in an ordinary re¬ 
ceiver. 

8. In panoramic reception there are many different 
ways of presenting the information, and different types 
of indicators may serve different purposes. 

9. The characteristics of vision must be taken into 
account in panoramic reception. 

122 FREQUENCY ALLOCATIONS 

Choice of a frequency allocation for a panoramic re¬ 
ceiver, that is, choosing of the frequencies for beating 
oscillators, i-f circuits, etc., is apt to prove difficult. 
Factors entering into this choice are (1) the location 
and width of the frequency band or bands to be 
scanned, (2) the capabilities of sweep oscillators, (3) 
the avoidance of multiple responses, and (4) the selec¬ 
tivity realizable at different frequencies. 

12-21 Scanning Band 

The frequency band to be scanned may be located 
almost anywhere in the frequency spectrum. The width 
of the scanning band is ordinarily determined either 
by the fineness of resolution desired or by the number 
of signals that can be simultaneously monitored. 

Frequently when a panoramic receiver is to cover a 
wide frequency band, it may also be desired to scan 
alternatively narrower bands capable of being posi¬ 
tioned anywhere within the wider band. This is be¬ 
cause a narrowing down of the band provides better 
resolution and fewer signals to claim the operator’s 
attention. However, either breadth or variety in the 


scanning band is apt to be reflected in a multiplication 
of i-f circuits and complexity of equipment. 

12-2-2 Sweep Oscillators 

Sweep oscillators used in panoramic reception are 
mostly of two types: 

1. Electronically controlled oscillators, that is, f-m 
oscillators whose frequency is varied by means of a 
reactance tube which applies a quadrature voltage to 
the oscillating circuit. The amplitude of the quadra¬ 
ture voltage is ordinarily determined by means of a 
sawtooth wave applied to the reactance tube so that 
each sweep of the frequency band is followed by a 
rapid return. 9 With such an oscillator, the location 
and width of the scanning band may be readily 
changed by simple potentiometer controls acting upon 
the sawtooth wave. Using conventional oscillator cir¬ 
cuits, electronic sweep oscillators can be built for fre¬ 
quencies up to approximately 100 me and reasonable 
linearity can be obtained for a sweep of the order of 
20 per cent of the top frequency. 

2. Mechanically controlled oscillators, that is, oscil¬ 
lators whose frequency is controlled by using mechani¬ 
cal motion to change capacitance, inductance, or both, 
in the oscillator circuit. The element which is varied 
may be a capacitor, a Lecher wire, or a cavity. Nor¬ 
mally with this form of oscillator the sweep through 
the frequency band in one direction and the return in 
the opposite direction are at about the same rate. Since 
it is difficult to obtain coincidence of the traces for the 
two directions of scanning, blanking of one scan may 
be necessary. Mechanically controlled oscillators are 
particularly useful in the frequency range above that 
of electronically controlled oscillators. The percentage 
of frequency swing may be quite large. 

Other ways of controlling the frequency of an oscil¬ 
lator are possible, but have not been used to any sub¬ 
stantial extent. One of these, for example, is to control 
the oscillator inductance by varying the d-c flux 
through the core of an h-f inductance coil. 

By means of one or more frequency multipliers the 
frequency band of a sweep oscillator may be moved 
upward, the percentage frequency band remaining un¬ 
changed. Suitable suppression of unwanted frequen¬ 
cies is necessary. In case the percentage frequency 
band covered by a sweep oscillator is less than that de¬ 
sired, it is usually possible to obtain a greater percent¬ 
age by building the sweep oscillator for a higher fre¬ 
quency. The sweep frequencies may then be modulated 


c 


CONFIDENTIAL 


i 





FREQUENCY ALLOCATIONS 


65 


down to a lower location if desired. Inversely, the per¬ 
centage sweep necessary to scan a given frequency 
band may be reduced by shifting the signal band up¬ 
ward in frequency. 

If two or more different widths of scanning band 
are desired, and if the ratio of the maximum scanning 
band to the minimum scanning band is quite large, it 
may be difficult to provide both wide and narrow 
sweeps in the same oscillator. This is because the fre¬ 
quency control for the narrow sweep, whether electri¬ 
cal or mechanical, becomes too tine. A practical solu¬ 
tion of this problem may be to provide more than one 
scanning oscillator and a corresponding number of i-f 
amplifiers. Thus, a first oscillator might scan the en¬ 
tire band and a second oscillator a part of the band. 
All scanning oscillators must, of course, be controlled 
by the same sweep circuit. The different scanning os¬ 
cillators may be used alternatively or, if desired, com¬ 
binations may be used for wide-hand scanning. Mul¬ 
tiple scanning oscillators furnish an effective way of 
reducing the ratio of top frequency to scanning band 
for any individual oscillator but they tend to compli¬ 
cate filtering and shielding problems. All circuits 
which precede any scanning oscillator must be as wide 
as the scanning band of that particular oscillator. 

12.2.3 Heterodyne Methods—Multiple 

Responses 

Heterodyne technique is desirable to obtain the 
selectivity and gain for panoramic reception. It is well 
known that one of the most important problems in 
using the heterodyne method is the avoidance of mul¬ 
tiple responses. Let us consider first a single hetero¬ 
dyne receiver, one in which a single intermediate fre¬ 
quency is employed. The most important type of un¬ 
wanted response is the so-called image or second chan¬ 
nel response which occurs when the intermediate fre¬ 
quency is less than half the width of the band to be 
covered. 10 If the frequency band to be covered has a 
ratio of top to bottom frequency of more than 3 to 1, 
there is no intermediate frequency below the band for 
which image response will not occur. In addition to 
image response, there are a number of other types of 
response which may be serious, including higher-order 
combinations of signal and oscillator frequencies. If 
the intermediate frequency of a single heterodyne re¬ 
ceiver is above the band to be scanned, the selecting 
circuit employed in the scanning process must be at a 
relatively high frequency and it is usually impractical 
to obtain the desired selectivity characteristics. 


Image response, as well as other undesired re¬ 
sponses, can he reduced by using r-f tuning, controlled 
either electronically or mechanically so as to track 
with the beating oscillator. This, however, is difficult 
in a wide-hand panoramic receiver. 

A way of avoiding the multiple response difficulties 
of single heterodyne reception is to employ instead a 
double superheterodyne method, with the first inter¬ 
mediate frequency placed above the signal band. This 
method, however, leads to other possibilities of mul¬ 
tiple response which must be taken into account. 11 The 
principal trouble in this case is due to spurious re¬ 
sponses resulting from higher-order combinations of 
the two beating oscillators. In practice it has been 
found difficult, even with extremely careful shielding, 
to reduce this type of response to a satisfactory value 
unless the frequency allocation is such that only dif¬ 
ference frequencies resulting from the fourth or 
higher-order harmonics of the two oscillators can yield 
the first intermediate frequency. This requirement 
means that the first intermediate frequency must be 
more than three times the highest frequency in the 
signal band, the precise ratio depending upon the 
selectivity available in the two i-f circuits and the fre¬ 
quency of the second i-f stage. Also the frequency of 
the first beating oscillator should be placed above the 
first intermediate frequency. 

Even when these conditions are fulfilled, careful 
design is necessary to minimize different types of un¬ 
desired response. Also there are two rides to be ob¬ 
served. First, to avoid spurious responses due to in¬ 
sufficient shielding or selectivity, it is desirable that 
no intermediate frequency or beating-oscillator fre¬ 
quency should fall within the input hand. Second, to 
avoid double response the location of the intermediate 
frequency should not be less than half of the width of 
the preceding i-f circuit as measured between atten¬ 
uating regions. This means that it is impracticable to 
go from a very high intermediate frequency to a very 
low one. Thus if the i-f amplifier operates at approxi¬ 
mately 1,000 me, and if the 3-to-l ratio mentioned 
above is preserved, an upper frequency of about 300 
me for the signal band is indicated. 

1224 Selectivity 

The characteristics of available selecting circuits 
play a large part in determining the location and num¬ 
ber of i-f stages in a panoramic receiver. There are 
several reasons for this. First, the requirements for 
the final selecting circuit, which determines the re- 


UONF1D FNT1AL 







66 


THE FUNDAMENTALS OF PANORAMIC RECEPTION 


ceiver resolution, aie much more rigorous than those 
for the final selecting circuit in an ordinary receiver. 
This exerts an important influence on the choice of 
the last intermediate frequency. Secondly, rapidity of 
attenuation rise in any i-f circuit affects the location 
of the succeeding intermediate frequency as noted 
above. Another rule is that the pass hands of all preced¬ 
ing circuits should he wider than the final selecting 



130 ± to mc too± 0.3 MC 


500 kc, which is a suitable but not necessarily pre¬ 
ferred location for the selecting filter. 

The allocation shown in Figure 2 is merely one out 
of a practically infinite number which would accom¬ 
plish the same result. For a wise choice, careful study 
of the capabilities of components and comparison of 
at least a moderate number of different allocations 
would he essential. 


0-5 MC 



13.3 mc 


Figure 2. Illustrative frequency allocation plan. 


circuit so as not to affect the performace of that circuit. 

Limitations of realizable selecting circuits therefore 
tend in the same direction as both sweep oscillator 
limitations and considerations of multiple response, 
namely, toward additional numbers of i-f stages. 
Hence triple or quadruple heterodyne technique may 
be desirable for panoramic reception. 

To illustrate the frequency allocation principles dis¬ 
cussed above, there is shown in Figure 2 a possible 
allocation diagram for a panoramic receiver intended 
to scan either the complete frequency range of 20 to 
40 mc, or, alternatively, a hand of 1 me anywhere in 
that range. Triple modulation is employed. To sim¬ 
plify the sweep oscillator design, two different sweep 
oscillators are provided, one for the wide-range sweep 
and one for the 1-nrc sweep. At any one time, one of 
these would be automatically varied while the other 
would remain fixed. The third heating oscillator is 
invariable. 

The first intermediate frequency is placed at 100 
mc, or slightly more than three times the top frequen¬ 
cy of the input band. The pass band of this i-f stage 
must be as wide as that of the succeeding sweep oscil¬ 
lator, i.e., 1 mc. The second intermediate frequency 
must be higher than half the width of the first i-f 
circuit between cutoff points but should preferably lie 
below the input band. Accordingly it is placed at 15 
mc. The pass band of this second i-f circuit must be 
wide enough not to affect the scanning filter charac¬ 
teristic. The final intermediate frequency is placed at 


123 AUTOMATIC CONTROL OF 
SIGNAL INTENSITY 

In panoramic reception, the input signals may range 
from one or more microvolts up to several millivolts, 
an intensity range (also referred to as volume range) 
of 60 db or more. The maximum intensity range which 
can he applied to types of indicators commonly used 
is from 5 to 20 db, depending on various factors. Ac¬ 
cordingly it is necessary to provide some method of 
automatically reducing the signal intensity range 
ahead of the indicator. This requirement can be met 
by a device which acts upon the signals after they 
have been selected in the scanning process. 

A more fundamental and more serious disadvantage 
of a wide intensity range is that a strong signal can¬ 
not he cut off sufficiently rapidly in the final selecting 
circuit to prevent some masking of a nearby weak 
signal, the result of which is a loss of resolution. As 
discussed below, no satisfactory way of getting around 
this difficulty is available. 

Devices for reducing the range of signal intensities 
have been used in the radio and telephone arts for 
many years. 12,13 Similar devices may he used to reduce 
the intensity range in panoramic receivers but the 
performance requirements are somewhat different. De¬ 
vices useful for panoramic reception are of two general 
types (see Figure 3). First, there is limiting or gain 
control arranged to maintain substantially constant 
output over a certain range of input and usually hav- 


1 CONFIDENTIAL 




























DESIGN AND PERFORMANCE OF SCANNING FILTER 


67 


ing constant gain below this range. Second, there is 
the compressor, whose input-output characteristic 
when plotted on a db scale is a straight line with a 
slope less than unity. With either device a threshold 
may be provided to suppress the output when the in¬ 
put is below a certain minimum value. Either one 
would consist of a circuit (sometimes referred to as a 
vario-losser) in which the loss or gain is changed in 
accordance with the amplitude of the signals, either 
directly or by means of an auxiliary control circuit. 


<D 

O 



input input db 

LIMITER OR GAIN CONTROL COMPRESSOR 

o) WITHOUT THRESHOLD 

b) WITH THRESHOLD 

Figure 3. Characteristics of devices for intensity control. 

Since there exists, prior to the final selection, no 
basis for obtaining a differential action as between 
signals, the intelligence necessary to operate the de¬ 
vice for controlling intensity range must be obtained 
after the desired signal has been selected from other 
signals. Moreover, any control device introduced ahead 
of the final selection would probably produce objec¬ 
tionable intermodulation between signals. A possible 
way of reducing the signal intensity range ahead of 
the selecting process is to employ tuned rejection cir¬ 
cuits to cut down signals that are especially high. 
Such an arrangement, however, involves a sacrifice in 
resolution, and becomes difficult and expensive where 
a number of stations with large signal strengths occur 
in the frequency band. 

The wanted signal delivered by the final selecting 
circuit consists of brief spurts of energy when signals 
are encountered. The intensity control device may op¬ 
erate either from the a-c signal or from the corre¬ 
sponding rectified signal. Distortion is of no conse¬ 
quence, since the only use made of the high-frequency 
pulses delivered by the selecting circuit is to rectify 
them to obtain a visual indication. Owing to the brief 
duration of each signal, it is generally desirable that 
the action of the device used to reduce the intensity 
range should be practically instantaneous both in 
attack and release. This makes it important to con¬ 
sider time constants carefully and may rule out de¬ 
vices in which a backward-acting circuit controls the 
vario-losser. 

While either limiting or compression may be used, 


compression is usually preferable, since it preserves 
some distinction between signals of different ampli¬ 
tudes. One advantage of such a distinction is that it 
facilitates differentiation between a weak signal and 
the transient response associated with a much stronger 
signal. A compressor in the form of a cascade limiter, 
comprising a series of interstages arranged so that suc¬ 
cessive interstages produce a limiting effect for in¬ 
creasing values of input, has been found useful. 2 

124 DESIGN AND PERFORMANCE OF 
SCANNING FILTER 

The circuit which selects in succession the differ¬ 
ent signals in the frequency band may be a single 
filter or tuned circuit, or it may consist of one or more 
tuned amplifier stages. In any case, it will be referred 
to below as the scanning filter. The resolution of a 
scanning receiver, that is, the minimum frequency 
separation for which it is possible to differentiate be¬ 
tween two signals of adjacent frequency, is dependent 
upon the design of this filter. Assuming a given filter 
design, the resolution obtainable depends upon the 
level difference between input signals. A further fac¬ 
tor in resolution is the capability of the indicator. 
However, it will be assumed in this section that the 
indicator is able to utilize the resolution obtainable 
with the scanning filter. The subject of indicator reso¬ 
lution will be taken up subsequently. 

When signals having broad spectra are to be re¬ 
solved, as for f-m transmission or transmission of 
short pulses, the design of the scanning filter and the 
resolution which it affords become of much less im¬ 
portance, since these types of signals are placed farther 
apart in frequency and since the boundaries of any 
one signal spectrum will lie less clearly defined. The 
following discussion therefore refers largely to sig¬ 
nals of fairly narrow spectra. 

124 1 Outline of Problems 

Because filters are made up of inductive and capac¬ 
itive elements which have appreciable time constants, 
and because the input frequencies are swept past the 
scanning filter, the design and performance of this 
filter must be considered in terms of transient re¬ 
sponse. The principles involved therefore differ from 
those for a filter or selective circuit for ordinary 
reception. 

The signals applied to the scanning filter are in the 


CONFIDENTIAL 










68 


THE FUNDAMENTALS OF PANORAMIC RECEPTION 


form of a succession of frequency-modulated waves 
whose rate of frequency change is the scanning speed. 
Assuming negligible retrace time in the frequency 
sweep, the scanning speed is equal to the product of 
the frequency band swept over and the repetition rate. 
When the scanning speed in radians per second per 
second is large in comparison with the time constant 
of the scanning filter, then the filter response is 
stretched out, which produces confusion between sig¬ 
nals of adjacent frequency and hence brings about loss 
of resolution. If, on the other hand, the rate of fre¬ 
quency change is very small compared to the filter 
time constant, then the filter is wider than it needs 
to be and the resolution consequently poorer. 

The width of the scanning filter and the scanning 
speed affect also the signal-to-noise ratio for the scan¬ 
ning receiver. In particular there exists for any given 
scanning speed a value of filter band width which 
yields optimum signal-to-noise ratio. 

Problems to be considered in connection with the 
scanning filter therefore include, first, determining the 
optimum design of scanning filter from the stand¬ 
point of (a) resolution and (b) signal-to-noise ratio; 
and second, the related but nevertheless distinct prob¬ 
lem of determining the performance obtained with a 
given design of filter under different operating condi¬ 
tions. While the basic theoretical approach to these 
problems is in terms of the transient response of the 
filter, the results can be expressed in terms of its 
steady-state response characteristics. 

The complete problem of scanning filter design is 
extremely complex. It was first studied a number of 
years ago. 14 Recently it has been treated more compre¬ 
hensively, but so many variables are involved that 
solutions thus far have had to be based on simplified 
and somewhat idealized assumptions. However, theo¬ 
retical work has been confirmed and supplemented by 
experimental results which have been particularly use¬ 
ful in furnishing quantitative relationships. 

12A.2 Analogy with Quasi-Stationary 
Filter Response 

A qualitative picture of the effect of scanning filter 
characteristics on resolution may be had by analogy 
with the more familiar problem of the response of a 
filter to an a-c pulse the fundamental frequency of 
which is the mid-frequency of the filter. Assume that 
such an a-c pulse is applied to a filter which is criti¬ 
cally damped, i.e., whose characteristics are such that 
the output rises quickly to its steady-state value but 


does not exceed it. Assume further that the filter has 
a discrimination of 12 db or more per octave of side¬ 
band frequency. It can be shown that for such a filter 
the build-up time approximates 1/B where B is the 
band width of the filter as measured between 6-db 
points. For this case, when the length of applied pulse 
is equal to or greater than the build-up time of the 
filter, then the duration of the response as measured 
between the half-amplitude points is the same as the 
length of applied pulse. If, however, the length of the 
applied pulse is less than the build-up time, the dura¬ 
tion of response is equal to the build-up time, so that 
the response is stretched out and reduced in amplitude. 

Now consider what happens in scanning. As an in¬ 
put frequency is swept past the scanning filter, the 
output response takes the form of an a-c pulse. The 
rectified response corresponding to this pulse is some¬ 
what similar to that obtained by turning on a mid¬ 
frequency wave when the scanned signal reaches the 
leading edge of the filter and turning this off as the 
scanned signal passes the trailing edge. The nominal 
duration of this assumed output pulse is 

r* = -, (i) 

7 

where B = width of scanning filter in cycles per sec¬ 
ond and y = scanning speed in cycles per second 
(y = nF where n = repetition rate and F — frequen¬ 
cy band swept over). The band width B is measured 
between points where the filter discrimination is 6 db. 

If the actual pulse is assumed to correspond to a 
mid-frequency pulse of duration T then the simple 
theory outlined above furnishes a rough criterion for 
determining the duration of the response. The dura¬ 
tion of response when multiplied by the scanning speed 
gives the apparent band width of response, which may 
be taken as an approximate indication of the resolu¬ 
tion obtainable between signals of equal amplitude. 

If the resolution is plotted against the filter band 
width B for a fixed scanning speed y, a minimum oc¬ 
curs when the filter is just Avide enough to transmit 
the nominal pulse of equation (1) without any sub¬ 
stantial decrease of amplitude and stretching out of 
response. This minimum determines the optimum 
band width for the assumed scanning speed. 

Consider next the law of variation of optimum 
band width with change of scanning speed y. Assume 
that the band width is left fixed and that y is increased 
above the optimum for that filter. This stretches out 
the response and decreases the pulse amplitude. To 
restore optimum design, it is evidently necessary to 
increase the band width. However, such increase gives 


( 0 N F UlSE NTJ A1. 





DESIGN AND PERFORMANCE OF SCANNING FILTER 


69 


a double benefit in that the length of pulse to be han¬ 
dled increases at the same rate as the ability of the 
filter to handle a given pulse length. The change in 
band width necessary to restore optimum design cor¬ 
responds to the change in the square root of scanning 
speed. That is, 

B 0 = K„\/y, (2) 

where B 0 = optimum width of scanning filter in 
cycles per second and K B may be termed the band¬ 
width factor. As before, filter band width is measured 
between 6-db points. 


12 4 3 Experimental ^ allies of Band Width 
and Resolution for Equal-Level Signals 

The foregoing discussion is based on simplified as¬ 
sumptions which account inadequately for the tran¬ 
sient phenomena associated with scanning. Experience 
with a number of actual filters has indicated how the 
simple theoretical relationships must be modified for 
practical use. 

Equation (2) for optimum filter band width has 
been found to apply reasonably well. However, the 



Figure 4. Resolution versus scanning speed. 


The minimum or best value of resolution occurs 
when the filter band width is about equal to yfy 
(K B — 1), the exact value depending on the filter 
damping. This value of scanning speed (y — B 2 ) 
may be termed the critical scanning speed. 

If now the filter band width B is left fixed while 
the scanning speed y is varied, a theoretical curve can 
be obtained for resolution as a function of scanning 
speed. Below the critical scanning speed the resolu¬ 
tion remains practically constant, while above the 
critical value the resolution becomes poorer directly 
as scanning speed increases. 


value of K B varies considerably, depending on the 
filter design. Since it is usually not convenient to vary 
the filter band width, the scanning speed may be 
varied instead and a curve obtained from which the 
value of K b may be computed. Values between 0.8 
and 2 have been observed. 

The optimum value of resolution S o for a given 
filter may be indicated by the following equation: 

s „= Tr B °- (3) 

Values of the resolution S for a given filter may be 










































70 


THE FUNDAMENTALS OF PANORAMIC RECEPTION 


determined by varying the scanning speed. A typical 
curve of S versus y, using equal-level signals, is shown 
at A in Figure 4. Above the critical scanning speed 
the resolution becomes much poorer, approaching di¬ 
rect proportionality to scanning speed, while below 
the critical speed it is more or less constant. This 
curve is intended for purposes of illustration and not 
for precise computation. More detailed experimental 
results are presented elsewhere. 1 Curve B shows the 
locus of best resolution when the filter band width is 
made optimum for each scanning speed. The point 
where the two curves are tangent represents the op¬ 
timum or critical scanning speed for the assumed 
filter band width. 

From such determination of resolution the value of 
K a in equation (3) may be determined. This value 
varies considerably, depending upon the type of filter, 
type of indicator, and the level difference between sig¬ 
nals being resolved. With equal-level signals and with 
an indicator affording maximum resolution, the spread 
of observed values of K s is from about 1.2 to 3. The 
value of 1.2 was obtained for a filter which also had 
a value of K B = 1.2, this being the best combination 
thus far obtained. 

A chart showing optimum filter band width or op¬ 
timum resolution as a function of scanning speed, 
assuming K B or K s = 1.2, appears in Figure 5. 


Filter band width or resolution for other values of 
K B or K s may be readily obtained by multiplying 
the values shown on the chart. A nomographic chart 
may be employed to derive optimum filter band width 
or optimum resolution directly from the three vari¬ 
ables: (1) scanning band, (2) repetition rate, and 
(3) I\ B or K s . 

12AA Effect of Level Difference 

Thus far consideration has been limited to the case 
where the two signals to be resolved are of equal level. 
For this condition the filter may be designed with a 
band width determined by assuming an appropriate 
value of K b and with reasonably flat delay between 
the 6-db points. The amount of cutoff required beyond 
the 6-db points will depend upon the type of indicator, 
but in general a moderate cutoff, of the order of 12 
db or more per octave of side-band frequency, would 
be satisfactory. 

If a large level difference exists between the two 
adjacent signals, then the performance of the filter in 
the cutoff region assumes greater importance. This 
is for the reason that if a strong signal is not cut off 
rapidly enough it will mask a nearby signal, produc¬ 
ing loss of resolution. Delay distortion in the region 
adjacent to the pass band of the filter may also give 



.1 .2 .4 .6 .8 I 2 4 6 8 10 20 40 60 

SCANNING SPEED MC/SEC 2 


Figure 5. Chart for determining filter band width and resolution as a function of scanning speed. 




































































DESIGN AND PERFORMANCE OF SCANNING FILTER 


71 


rise to substantial impairment of resolution. These 
effects are most pronounced when the larger signal is 
scanned first, so that the trailing edge of the response 
is unduly prolonged. Hence, in designing a filter to 
be used for large level difference, it is necessary to 
obtain rapid and sustained cutoff together with rea¬ 
sonably small delay distortion extending as far into 
the cutoff region as practicable. A cutoff which be¬ 
comes asymptotic to from 18 to 40 db per octave of 
side-band frequency, depending upon the level differ¬ 
ence, is desirable. 

Observations of resolution indicate that the op¬ 
timum scanning speed for any given filter design is 
about the same for relative signal strengths ranging 


noise ratio for a scanning receiver much poorer than 
that obtainable in a nonscanning receiver. Here it is 
interesting to consider the relation between scanning 
filter band width and signal-to-noise ratio. If the filter 
band width is chosen for best resolution, then the 
noise power varies inversely as the square root of the 
scanning speed. Hence, as low a scanning speed as 
practicable for best resolution is advantageous for 
signal-to-noise ratio as well. Looking at the filter 
purely from the standpoint of signal-to-noise ratio, 
two questions are of interest: (1) What is the opti- 
mum band width for a given scanning speed? (2) With 
a given band width, how does the signal-to-noise ratio 
vary as the scanning speed is changed ? 



SCANNING SPEED _ 

SCANNING SPEED FOR BEST RESOLUTION 


Figure 6. Signal-to-noise 

from 0 to 40 db, so that the filter band width infor¬ 
mation already presented for equal-level signals may 
be used in the case of unequal levels. 

With the best present technique, the value of K s 
increases with level difference somewhat as follows: 


Level difference (db) K , s 

0 1.2 

20 2.9 

40 5.3 

50 12.0 


With poorer filter design or poorer indicator resolu¬ 
tion the values of K s for large level differences are 
much higher. 

12 4 5 Signal-to-Noise Ratio 

© 

It has been pointed out that the band width required 
for the scanning filter tends to make the signal-to- 


ratio versus scanning speed. 

Experiments using a fixed filter have shown that 
the noise remains unchanged as the scanning velocity 
is varied over a wide range. However, as the scanning 
speed is increased above the critical resolution speed, 
the signal response decreases, so that the signal-to- 
noise ratio is reduced. Below the critical speed the sig¬ 
nal response remains substantially constant, so that 
the signal-to-noise ratio is also constant. In this range, 
however, it would be possible, by using a narrower 
filter, to obtain substantially the same signal response 
with less noise. The complete curve of relative signal- 
to-noise ratio versus relative scanning speed therefore 
takes the general shape shown in curve A of Figure 6, 
while curve B shows the locus of signal-to-noise ratios 
when the filter band width is optimum for each scan¬ 
ning speed. The difference indicates the impairment 
in signal-to-noise ratio incurred by using a filter at a 
scanning speed other than that for best resolution. 


CONFIDENTIAL 
































72 


THE FUNDAMENTALS OF PANORAMIC RECEPTION 


On the basis of experiments along this line, it appears 
that the best signal-to-noise ratio for a good scanning 
filter occurs at a scanning speed quite near the opti¬ 
mum for good resolution. 

12A.6 Use Q ne Filt er a t Two or More 
Scanning Speeds 

Under certain conditions it may be desirable, in 
the interest of apparatus economy, to operate a single 
scanning filter at two or more different scanning 
speeds. For such conditions the type of results to be 
expected as regards resolution and signal-to-noise ratio 
are indicated in Figures 4 and 6. These curves should 
be interpreted qualitatively since the actual results 
will differ considerably, depending on filter design, 
type of indicator, etc. It would appear from these 
curves that for best resolution with two different scan¬ 
ning speeds the filter band width should be designed 
for approximately the geometric mean between the 
two speeds. If signal-to-noise ratio should be of para¬ 
mount importance the best value would be somewhat 
less than the geometric mean. It appears from avail¬ 
able data that with good technique the resolution 
probably will not be degraded more than 10 per cent 
for scanning velocities ranging between 0.5 and 2 
times the optimum. The degradation in signal-to- 
noise ratio for a corresponding range should be only 
a few decibels. 

12 - 4 - 7 Location of Scanning Filter 

Determination of the preferred frequency location 
for obtaining the desired characteristics in the scan¬ 
ning filter depends upon available technique in dif¬ 
ferent parts of the frequency range and it is not pos¬ 
sible to lay down any definite rules. Generally the loca¬ 
tion must be a compromise between the desire to use a 
relatively low frequency position, where a narrow 
band can easily be secured, and a higher location which 
would avoid one or more additional steps of modula¬ 
tion to bring the signals down to the filter frequency. 

A location which would be advantageous purely 
from the standpoint of filter design is at zero fre¬ 
quency, since this would mean that the filter would 
need only one cutoff instead of two. However, if the 
signals were brought to zero frequency by rectification, 
it would be necessary first to separate them with a 
high-frequency filter affording a resolution substan¬ 


tially the same as the low-pass filter, so that no gain 
Avould result. It is possible that signals might be 
homodyned to zero frequency, but this has not been 
studied. 

12 - 4 - 8 Desirable Scanning Filter 
Characteristics 

The resolution 3 obtainable with scanning filters of 
various types was determined over a range of scan¬ 
ning velocities. The characteristics of the filter afford¬ 
ing the best resolution may be summarized as follows: 

1. The filter band width as measured between its 
6-db discrimination points should be equal to approxi¬ 
mately 1.2 times the square root of the scanning veloc¬ 
ity. That is, B = 1.2 y/nF where B — band width in 
cycles per second, n — the number of scans per second 
and F = the band scanned in cycles per second. 

2. The attenuation outside the pass band of the 
filter should become asymptotic to about 30 to 40 db 
per octave of side-band frequency. 

3. The response of the filter to suddenly applied 
pulses of a fundamental frequency equal to the mid¬ 
frequency of the filter should overshoot the steady- 
state value by 10 to 20 per cent. 

4. The delay distortion across the pass band should 
be as small as is consistent with the above require¬ 
ments. 

The relation between steady-state characteristics 
and the characteristic given in (3) above is complex, 
depending on filter configuration. For example, in a 
filter with selectivity of 4 or 5 times 6 db per octave 
of side-band frequency, a square wave response exhib¬ 
iting approximately 20 per cent overshoot is attained 
when the steady-state phase slope has a minimum at 
mid-band and maxima near the band edges, with a 
maximum-minimum ratio of 1.6. 

125 PANORAMIC INDICATORS 

The type of portrayal with which most of the dis¬ 
cussion in this section will be concerned is that of 
signal presence versus frequency. Another type of 
display is one of signal patterns versus frequency, 
that is to say, an arrangement side by side of a num¬ 
ber of facsimile patterns for different signals so as to 
permit simultaneous viewing. Other characteristics 
which may be shown are signal amplitude versus fre¬ 
quency, azimuth of signal source, etc. 

By far the most desirable and most versatile form 


CONFIDENTIAL 





PANORAMIC INDICATORS 


73 


of indicator for a panoramic receiver is a cathode-ray 
tube. Further discussion will therefore be predicated 
upon this device except for specific treatment of other 
types of indicators. 

12,51 Diagrams of Signal Presence versus 
Frequency 

The simplest way of diagramming signal presence 
versus frequency on a cathode-ray tube is to cause the 
cathode-ray beam to swing in fairly rapid succession 
over a path of chosen shape, distance along which in¬ 
dicates frequency, with signals indicated either (1) by 
deflecting the beam sidewise with respect to the fre¬ 
quency trace or (2) by modulating the intensity of 
the beam. 

Beam deflection affords a higher degree of resolu¬ 
tion, the amount of the difference depending princi¬ 
pally upon the design of the scanning filter and the 
level difference between input signals. On the other 
hand, beam deflection gives patterns which dance up 
and down because of signal modulation, static, etc., 
resulting in confusion and fatigue to the observer. 
Also if the frequency trace is folded on the tube face 
in any manner to obtain a longer frequency scale, 
beam deflection affords an opportunity for confusion 
between adjacent traces. 

Beam modulation results in a diagram which is 
pleasing to the observer and gives less confusion be¬ 
tween adjacent parts of a doubled-up frequency scale. 
The relative advantages of pleasing diagram as com¬ 
pared with superior resolution will depend on the 
circumstances involved. A disadvantage of beam mod¬ 
ulation is that during periods when severe static 
crashes occur at frequent intervals the entire fre¬ 
quency scale is illuminated by each crash and obser¬ 
vance of signals during periods between static crashes 
becomes quite difficult, especially if a persistent screen 
is employed. 

Single-Line Trace 

The simplest type of trace on which to show signal 
presence versus frequency is a single line produced by 
employing, for horizontal (or vertical) deflection, a 
wave corresponding to the frequency sweep. Such a 
single-line diagram is extremely satisfactory for many 
purposes. Sometimes, however, especially when scan¬ 
ning a wide band containing a large number of sig¬ 
nals, the length of frequency scale obtained with a 
single line may be inadequate. In such cases a longer 


scale can be obtained, at a sacrifice of simplicity, in 
any of a number of ways. With any such long-scale 
arrangement, either beam modulation or beam deflec¬ 
tion may be used to indicate signals, but in some in¬ 
stances beam deflection involves greater complication. 

When stations occur at close intervals it is readily 
possible with available technique to display an amount 
of information far greater than can be appreciated. 
Hence the practical length of scale is usually fixed by 
observer capability. This varies considerably as be¬ 
tween observers and depends upon the purpose for 
which the observations are made. Though no exact rule 
can be laid down, it appears that a single observer 
should not be required to monitor more than 15 or 
20 stations at one time. If, however, it is permissible 
for the observer to shift his attention at intervals from 
one group of stations to another, a considerably larger 
number might be displayed. 

Circular Trace 

One way to lengthen the frequency scale is to em¬ 
ploy a circular trace. This can be produced with an 
ordinary cathode-ray tube by applying to the vertical 
and horizontal deflecting plates or coils, respectively, 



TO FREQ 
SWEEP 


DEFLECTING WAVES 



Figure 7. Method of producing circular frequency diagram. 

two quadrature components of a sine wave whose fre¬ 
quency is the same as that of the frequency sweep. 
The arrangement is shown schematically in Figure 7. 
If it is desired to show signal presence by deflecting 


: CONFIDENTIAL 































74 


THE FUNDAMENTALS OF PANORAMIC RECEPTION 


the beam, this can be done by modulating the two 
quadrature components in accordance with rectified 
signal amplitude as indicated by the dotted lines. It 
is important in this case that well-balanced modulators 
with a minimum of unwanted intermodulation prod¬ 
ucts be employed. Deflection to indicate signals may 
be toward or away from the center. An alternative 
way of producing the circular trace is to use mechan¬ 
ical rotation of magnetic deflecting coils. 

Spiral Trace 

To produce a spiral trace, 1 the quadrature compo¬ 
nents which, if unmodulated, would give a circle are 
modulated by a sawtooth wave corresponding to the 
frequency sweep as shown in Figure 8. The number 


SIGNALS 



TO FREQ 
SWEEP 





Figure 8. Method of producing spiral frequency diagram. 

of turns in the spiral is determined by the ratio be¬ 
tween the frequency of the sine wave oscillator and 
that of the sawtooth wave. An alternative method of 
producing a spiral is to employ a physically rotating 
magnetic field, the intensity of which is varied in 
accordance with the sawtooth frequency sweep. 

Modulation of the quadrature currents by the sweep 
sawtooth wave involves some difficulty if a reasonable 
approach to a true spiral is wanted. The sawtooth wave 


may be considered as composed of a number of sine 
wave components corresponding to the sawtooth fre¬ 
quency and its harmonics. In the modulation process 
it is necessary to preserve the side frequencies repre¬ 
senting sawtooth components up to something like the 
tenth harmonic. Moreover, the nonlinear device which 
is used for the modulating process also generates har¬ 
monics of the quadrature sine wave frequencies. These 
harmonics lie in the same frequency band as the modu¬ 
lation products necessary for the transmission of the 
sawtooth wave and hence cannot be filtered out. Ac¬ 
cordingly, careful design and balance of the modulat¬ 
ing circuits are required. Balanced vacuum-tube mod¬ 
ulators have been found satisfactory. The beam may 
be deflected to show signals by adding the rectified 
signals to the sawtooth modulating wave. 

Parallel Traces 

A parallel-line diagram may be obtained by the 
method shown in Figure 9. In this case a sawtooth 
wave corresponding to the frequency sweep is applied 



TO FREQ 
SWEEP 



VERTICAL 


/VWt 


HORIZONTAL 


Figure 9. Method of producing parallel-line diagram. 


to the vertical deflecting plates, and a sawtooth wave 
having a frequency equal to that of the vertical wave 
multiplied by the number of lines is used for horizon¬ 
tal deflection. The lines obtained with this arrange- 














































PANORAMIC INDICATORS 


75 


merit, while parallel, are not exactly horizontal. To 
obtain horizontality the vertical deflecting wave must 
be a stepped wave as shown in Figure 10. To indicate 
signals by beam deflection the rectified signal wave 
is added to the vertical deflecting wave. 



the retrace time, to avoid any loss whatever of fre¬ 
quency range during the retrace interval. This adds 
complications, however, and might introduce some 
degradation in receiver performance near the retrace. 13 
See illustration No. ES-808183 in the C-36 final re¬ 
port, dated January 22, 1943, for a method of ‘‘back¬ 
ing up.” 

Retrace discontinuities during the scanning inter¬ 
val may be avoided by using a continuous zigzag scale, 

SIGNALS- m 



SAWTOOTH 
GEN 
NO. 2 


TO FREQ 
SWEEP 



TO FREQ 
SWEEP 



VERTICAL 



VERTICAL 



HORIZONTAL 


/V1/V1 


horizontal 


FREQ 
SWEEP 

Figure 10. Method of producing horizontal parallel lines. 

With a parallel-line diagram, the finite retrace time 
of the horizontal sweep means that signals may appear 
between parallel lines. If the retrace time is made 
sufficiently small, the retrace can be blanked out with¬ 
out entire loss of a signal, since any signal would ap¬ 
pear in part at one or both of the opposite ends of 
adjacent lines. Theoretically it would be possible, by 
stopping or backing up the scanning oscillator during 



Figure 11. Method of producing zigzag diagram. 

which may be produced by the method shown in Fig¬ 
ure 11. The sawtooth wave which controls the fre¬ 
quency sweep is used for vertical deflection, while an 
isosceles triangular wave serves for the horizontal de¬ 
flection. An objection to a zigzag form of diagram is 

b Cathode-ray tube indicators 1 having from 2 to 8 parallel-line 
traces on both low- and high-persistence screens were set up 
for rapid comparison with the spiral-trace indicator. Demon¬ 
strations were given for various members of NDRC, Army, 
and Navy. Personal preference varied, some preferring the 
spiral due to the absence of retraces in the scale, others pre¬ 
ferring the parallel lines as being more nearly like printed page 
representation. The general conclusion seemed to be that for 
most applications there was little to choose between the types 
of traces. The high-persistence screen was preferred. It seems 
likely that each type of representation and various degrees of 
persistence may have advantages for certain special uses of 
panoramic receivers. 


CONFIDENTIAL 











































76 


THE FUNDAMENTALS OF PANORAMIC RECEPTION 


that since the spacing between lines is not uniform, 
there would be difficulty in distinguishing signals near 
the apexes of the zigzag. 

12 52 Repetition Rates 

The slower the repetition rate used for a frequency 
diagram, the narrower can be the scanning filter and 
hence the better the resolution. Too slow a rate, how¬ 
ever, means less effective signal monitoring. 

In earlier panoramic work flicker was considered to 
be an important factor in repetition rate and for this 
reason rates upwards of 15 cycles per second were 
employed. However, if a persistent phosphor of the 
cascade type is employed in conjunction with an 
orange viewing filter it becomes possible to employ 
repetition rates of less than about 4 cycles per second. 
Rates between 4 and 15 cycles per second are still 
somewhat objectionable as regards flicker. 

The speed with which the presence of a signal can 
be detected is limited by the repetition rate. Require¬ 
ments in this respect will depend upon the type and 
duration of the signals and upon the use being made 
of the panoramic receiver. 

For keyed telegraph signals of the ordinary variety, 
the time is about equally divided between marking and 
spacing. In view of the short time required to pass a 
given frequency location the probability of encounter¬ 
ing a marking signal in any one scan is only about 50 
per cent. The probability of detecting a telegraph sig¬ 
nal which is present in the frequency band accordingly 
depends upon the interval of observation and the rep¬ 
etition rate, as indicated by the following approximate 
formula: 

P = 100(1 — 0.5 n/T ) , (4) 

where P = per cent probability of detecting telegraph 
signals during the observing period T (with T as¬ 
sumed to be ^ the period of one repetition) and n = 
repetition rate. Thus for T = 1 sec and n— 1 cycle 
per second, P equals 50 per cent. From this stand¬ 
point a repetition rate of less than 1 cycle per second 
may be undesirable. 

Another disadvantage of an extremely low repeti¬ 
tion rate is that even with a very long-persistence 
screen the complete signal diagram does not appear 
on the screen at one time. For certain types of work 
this might be objectionable. The minimum rate for 
which a complete diagram can be obtained with avail¬ 
able phosphors is of the order of 1 cycle per second. 


From all of the above it appears that the preferred 
range of repetition rates for frequency diagrams is 
from approximately 1 to 4 cycles per second, although 
far different rates may be used for special purposes if 
desired. 

12.5.3 Type of Cathode-Ray Screen 

Tests have indicated that a long-persistence screen 
is generally advantageous for frequency diagrams. 
Long persistence helps in keeping a signal of short 
duration on the screen long enough to permit observa¬ 
tion and frequency determination. It greatly reduces 
flicker effects and eye fatigue and hence permits lower 
scanning rates with their inherently greater resolution 
and improvement in signal-to-noise ratio. Some fur¬ 
ther improvement in signal-to-noise ratio is obtained 
with a long-persistence screen as a result of cumula¬ 
tion of signal indications occurring at the same place 
on the screen, whereas noise traces occur at random. 
A disadvantage of a long-persistence screen, particu¬ 
larly when signal presence is indicated by spot modula¬ 
tion, is spreading of the phosphorescent trace. When 
the entire background is frequently illuminated by 
static crashes, long persistence is disadvantageous 
since it tends to prevent observation of signals by 
means of the fluorescent patterns between crashes. 

In practice a cascade screen employing a P7 phos¬ 
phor with blue-white fluorescence and orange phos¬ 
phorescence has been found quite satisfactory. By 
providing both orange and blue viewing filters either 
the fluorescence or the phosphorescence, as desired, 
may be masked to a large extent. 

l2 - 5 - 4 Indicator Resolution 

The resolution obtainable in a frequency diagram 
can be no better than the fraction of the total scan¬ 
ning band which corresponds to the ratio of the re¬ 
solving power of the indicator to the length of fre¬ 
quency scale. The resolving power of a cathode-ray 
tube indicator is determined by the diameter of the 
spot. While extremely small spots can be realized 
under laboratory conditions, practical operating re¬ 
sults are generally poorer. With a long-persistence 
screen the spot tends to spread, thus reducing resolv¬ 
ing power. Size of spot increases less rapidly than 
tube size, so that there is an advantage in going to 
larger tubes. When intensity modulation of the spot 
is employed, the resolution is much poorer because of 





PANORAMIC INDICATORS 


77 


defocusing or blooming of the spot as the beam cur¬ 
rent becomes large. When beam-intensity modulation 
is used the size of spot depends on the maximum beam 
intensity. As the “blooming” point is approached, the 
size of spot increases very rapidly. 

If the frequency scale is long enough so that the 
resolving power of the indicator is not a direct limi¬ 
tation, the capability of the indicator for distinguish¬ 
ing between two nearby signals still affects the scan¬ 
ning resolution. With the beam-deflection type of 
indicator, the signal takes the form of an inverted V 
or U according to the filter characteristic. Adjacent 
signals are distinguished by observing a slight dip 
between two inverted V’s or U’s which are not quite 
superimposed. For certain repetition rates, particular¬ 
ly in the range 5 to 15 cycles per second, a more sensi¬ 
tive discrimination index can be found in a beat whicli 
occurs between the two signals at the repetition rate. 
Under practical conditions, however, particularly 
when keyed signals and noise are present, this beat 
method of discrimination is of little value. 

With intensity modulation of the cathode-ray beam, 
signals are distinguished by differences between the 
size and shape of adjacent spots. Discrimination in 
this case is considerably poorer than that with beam 
deflection. 

12.5.5 Facsimile-Type Diagrams 

An alternative to a diagram of signal presence 
versus frequency is one in which facsimile-type pat¬ 
terns for a number of signals of different frequencies 
are arranged side by side for simultaneous viewing. 
One way of producing such a diagram, 2 is to employ 
a special type of cathode-ray tube which is rotated so 
as to give a continuous motion of the fluorescent screen 
in relation to the scanning beam. The sawtooth wave 
which controls the frequency sweep is employed to 
deflect the cathode-ray beam in one dimension across 
the screen, the moving screen makes time the other 
dimension, and signal amplitude modulates the beam 
intensity. Signal patterns are seen as a result of screen 
persistence. 

Low brightness of aftertrace is a limitation in the 
moving-screen tubes thus far constructed. This makes 
it necessary to observe in an almost completely dark¬ 
ened room. However, considerable improvement in 
this respect is believed possible. 

With a continuously rotating cathode-ray screen, 
there is a possibility that the phosphorescent traces 


after completing one revolution will not have decayed 
sufficiently to avoid interference with the new traces. 
This might be avoided by accelerating the decay of the 
stored energy subsequent to its passage across the view¬ 
ing window. For this purpose infrared rays could be 
used, which would free the stored energy largely as 
heat but partly as accelerated light emission. Although 
it appears possible to develop a satisfactory arrange¬ 
ment for wiping out the phosphorescent trace in this 
way, this has not been found necessary with the par¬ 
ticular type of moving screen tube thus far employed. 

An alternative method of obtaining a panoramic 
diagram of the facsimile type is to substitute for the 
moving cathode-ray tube a moving web of phosphor¬ 
escent material activated by a light beam whose posi¬ 
tion is adjusted in accordance with the frequency scan 
and whose intensity is modulated by the input signals. 0 

1256 Recording Radio Telegraph Signals 

With a suitably designed moving-screen indicator 
observation of frequency patterns is possible. Duration 
of transmission can be observed, which may permit 
the pairing up of transmissions of two stations. Tele¬ 
graph code with hand sending can be reproduced and 
read from the screen. By using a very narrow fre¬ 
quency sweep the varying frequency limits and syl¬ 
labic intensity changes of a-m speech side bands can 
be observed. 

For observation of such signal patterns there is an 
optimum rate of travel of the moving screen, since 
too slow a rate does not afford adequate resolution and 
too rapid a rate makes it impossible for the eye to 
follow the pattern. Satisfactory results have been 
obtained with a rate of screen travel of about 1 in. per 
second. To resolve telegraph code it is necessary that 
the repetition rate of the frequency scanning be suffi¬ 
ciently high to provide several scannings during the 
interval of a single telegraph dot. A repetition rate of 
60 cycles per second has been found satisfactory for 
hand telegraph speeds. Repetition rates much higher 
than this would afford little advantage for continuous 
reading of high-speed telegraph in view of the limita¬ 
tions of vision. 

The keying speed 1 of the fastest transmission that 
is to be recorded determines the minimum scanning 

°In November 1945, the Bell Telephone Laboratories dem¬ 
onstrated such a moving-web system designed to aid deaf 
persons to learn to talk by viewing moving visible images 
corresponding to speech sounds. 


CONFIDENTIAL 






78 


THE FUNDAMENTALS OF PANORAMIC RECEPTION 


frequency for the receiver. If it is assumed that there 
should be 3 scans per dot, the number of scans per 
second (w) should be six times the keying fundamen¬ 
tal frequency of 2.4 times the number of words per 
minute. 

The primary relationship underlying scanning re¬ 
ceiver operation may be represented by 
8 = j 5T s V nF, 

where S = signal resolution or, in this case, minimum 
frequency difference between adjacent stations in 
cycles per second. 

K s = a constant. 
n == number of scans per second. 

F — band width scanned in cycles per second. 

This means that the band width that may be scanned 
varies inversely with the number of words per minute 
of the highest speed signal to be recorded. Also the 
minimum allowable frequency separation between sig¬ 
nals increases as the square root of either the number 
of words per minute or the band width scanned. 

In conditions where the stations are uniformly 
spaced, we can rewrite the above equation as S = 
K 2 s nx, where x equals the number of stations re¬ 
corded. In other words, the number of equally spaced 
stations that may be simultaneously recorded varies 
directly with their frequency separation and inversely 
with the number of words per minute transmitted 
by the fastest station. 

As the relative amplitude increases, K s increases 
in a complicated manner, for example, relative am¬ 
plitudes of 30 db may mean that K s is three times 
what it would be for equal signals, i.e., the number of 
equally spaced signals that could be recorded would 
be reduced ninefold. 


Another element controlling the value of K s is the 
type of indicator used in the receiver. The indicator 
resulting in the lowest values of K s at the present time 
is an amplitude deflection indicator which is not suit¬ 
able for recording. The present best indicator for re¬ 
cording (the so-called spot or intensity indicator) 
causes several-fold sacrifice in K s . K a is then contin¬ 
gent on the amplitude ratio between the weakest and 
the strongest signals in the band scanned, rather than 
merely on the ratio between adjacent stations. 

Table 1 illustrates the interrelationship of the fac¬ 
tors discussed above. K s for equal signals is assumed 
to be 1.25 and for signals of 30 db relative strength 
is assumed to be 3.6, although these values are some¬ 
what smaller than could probably be realized under 
service conditions. 

12 - 5 - 7 Other Types of Diagrams 

For some applications, as for example, in identify¬ 
ing the character of signals or in matching signal 
strengths for different purposes, a diagram of ampli¬ 
tude versus frequency of radio signals is useful. Such 
a diagram is afforded by a cathode-ray tube using 
beam deflection as already described, provided limit¬ 
ing is not introduced ahead of the indicator. It should 
be noted that the resolution obtainable in a diagram 
to be used for amplitude versus frequency determina¬ 
tion is many times poorer than that obtainable when 
the requirement is merely to indicate what frequencies 
are present. In the latter case the transient responses 
before and after the main response from a signal may 
blend with the response from another signal to alter 
greatly its apparent magnitude without preventing 


Table 1. Effect of variables on scanning-receiver recording performance. 


Minimum 

station 

separation (kc) 

Maximum words 
per minute 
per station 

No. of scans 
per sec (n) 

Max. band scanned in kc for 
relative signal strength 
indicated 

Number of equally spaced 
signals (x) for relative 
strength indicated 

0 db 

30 db 

Odb 

30 db 

5 

30 

72 

222 

27 

44 

5 


50 

120 

133 

16 

27 

3 


100 

240 

67 

8 

13 

2 


200 

480 

33 

4 

7 

1 


250 

600 

27 

3 

5 

1 

10 

30 

72 

890 

107 

89 

11 


50 

120 

534 

64 

53 

6 


100 

240 

267 

32 

27 

3 


200 

480 

133 

16 

13 

2 


250 

600 

107 

13 

11 

1 

20 

30 

72 

3,560 

427 

178 

21 


50 

120 

2,130 

256 

107 

13 


100 

240 

1,070 

128 

53 

6 


200 

480 

535 

64 

27 

3 


250 

600 

427 

51 

21 

3 


CONFIDENTIAL 


















PANORAMIC INDICATORS 


79 


observation of its presence. When amplitudes are to be 
depicted accurately the signal separation must be 
sufficiently large to prevent intermingling of tran¬ 
sient responses. 

The azimuth of a number of radio signals of differ¬ 
ent frequency can be displayed simultaneously by 
means of a panoramic receiver. Such a diagram might 
be in the form of a straight line, distance along which 
represents azimuth, but more conveniently would take 
the form of a circle on which azimuth is represented 
by angular position and signal presence is indicated 
either by beam modulation or beam deflection toward 
or away from the center. A method of obtaining such 
a diagram, using beam deflection, is shown schematic¬ 
ally in Figure 12. In addition to the frequency scan 
it is necessary that the receiving antenna scan azimuth 
in any of various well-known ways. One scanning rate 
must be considerably higher than the other for ade¬ 


quate resolution. A similar arrangement might be 
used to provide a diagram of azimuth of signal source 
versus frequency. This might be a polar diagram in 
which frequency is represented by radial distance and 
azimuth by angular position, using the schematic ar¬ 
rangement of Figure 13. 

12.5.8 Mechanical Forms of Indicators 

Mechanical counterparts can be devised for most 
of the types of diagrams which have been described. 
In general these are obtained by (a) providing a 
source of light which is modulated in accordance with 
the signal intensities delivered by the scanning filter 
and (b) viewing this source of light through an aper¬ 
ture the position of which is mechanically varied syn¬ 
chronously with the frequency scanning. The source 



CRT 


Figure 12. Method of producing diagram in which azimuth is represented by angular position on circle. 



Figure 13. Method of producing circular diagram in which frequency is shown by distance along radius, and azimuth by 
angular position. 














































































80 


THE FUNDAMENTALS OF PANORAMIC RECEPTION 


of light may be, for example, a neon tube. If this 
source is arranged so as to illuminate a line which is 
viewed through a rotating cylinder provided with a 
helical slit, a single-line diagram of signal presence 
versus frequency can be obtained. Similarly by illu¬ 
minating a circular path around which the viewing 
aperture moves, a circular frequency diagram can be 
obtained. By illuminating an area and using suitable 
viewing arrangements, spiral or other types of dia¬ 
grams can be obtained. 

Mechanical devices of this kind may be smaller than 
cathode-ray indicators but seem likely to involve con¬ 
siderable sacrifice in resolution. 

1259 Alarms—Automatic Stopping 

When the number of signals in the scanning range 
is not large, a convenient addition to a scanning re¬ 
ceiver may be an audible alarm which sounds either 
momentarily or continuously whenever a signal is 
encountered. In some instances it may be useful to 
provide means for automatically stopping the scan 
upon encountering a signal, in order to permit exam¬ 
ination of it, or to indicate its presence or other char¬ 
acteristics if it disappears quickly. 


If the scanning speed actually used differs from the 
optimum for which the scanning filter was designed, 
there will be a sacrifice in signal-to-noise ratio as dis¬ 
cussed above. 

It is believed that the capabilities of the eye in 
picking out signals in the presence of noise are some¬ 
what poorer than those of the ear. Signal-to-noise dis¬ 
crimination possible in visual reception varies consid¬ 
erably, depending upon the type of indicator and the 
type of diagram. If a persistent phosphor is used, 
cumulation may be obtained for successive signals 
while noise is distributed at random. From this stand¬ 
point beam modulation is somewhat superior to beam 
deflection. 

It is possible that a slight gain in signal-to-noise 
discrimination can be obtained with a diagram in 
which each signal is made to appear as a line trace, 
either by moving the screen with respect to the cath¬ 
ode-ray beam or vice versa. Thus a time pattern 
can be obtained with a fixed cathode-ray tube by mov¬ 
ing the beam so that the trace corresponding to any 
signal is a line which may be circular, vertical, or 
horizontal. A method of producing circular signal 
traces is illustrated in Figure 14. This employs a 
radial frequency sweep and circular time sweep, the 



Figure 14. Method of producing circular signal traces. 


12.6 SENSITIVITY 

The signal-to-noise ratio obtainable in a well-de¬ 
signed panoramic receiver is necessarily poorer than 
that obtainable with an ordinary aural receiver. This 
is because the width of the selecting filter as deter¬ 
mined by scanning limitations is greater than that 
needed for fixed tuning. In the case of telegraph the 
difference in signal-to-noise ratio may range from 10 
to 30 db, depending on the scanning speed. For a-m 
speech transmission the difference is smaller and for 
f-m transmission it may be practically negligible. 


radial scanning rate being a high multiple of the cir¬ 
cular sweep. A method for producing vertical line 
traces is shown in Figure 15. In this case the horizon¬ 
tal sweep corresponds to the frequency scan while the 
vertical sweep is much slower. With any of these 
schemes for producing line traces, it appears likely 
that there may be an optimum trace velocity which 
yields the best discrimination between signal and noise. 

127 FREQUENCY DETERMINATION 

Generally it is desirable in a panoramic receiver to 
be able to determine the frequency of observed signals 


CONFIDENTIAL 


































FREQUENCY DETERMINATION 


81 


with considerable accuracy. This can he done in a 
number of ways. 

Fixed frequency markings can be provided on the 
face of the tube and arrangements in the receiver to 
align the frequency scan with these markings. Usually 
it is not possible to obtain sufficient accuracy with this 
method. 



Figure 15 . Method of producing vertical line traces. 


A complete frequency scale may he produced by 
electrical markings on the screen. Even though the 
electrically produced scale markings are precise, it 
may still be difficult to obtain the requisite accuracy 
in reading the frequency of observed signals by inter¬ 
polation between markings. 

The frequency-marking method which has been 
found of greatest utility is to provide a signal gener¬ 
ator whose frequency may he adjusted over the scan¬ 
ning band. This makes it possible to produce on the 
indicator a marking signal whose frequency can be 


made to coincide with that of any received signal. 
Such a signal generator may he advantageously com¬ 
bined with an aural receiver arranged to monitor any 
signal received by the panoramic receiver. In this case 
the marking frequency is made the same as the fre¬ 
quency to which the aural receiver is tuned. This per¬ 
mits tuning the aural receiver quickly to any desired 
signal trace on the screen and reading the frequency 
of that signal on the receiver dial. 

To avoid confusion the marking frequency may be 
indicated in such a way as to differentiate it from the 
received signals, for example, by using upward deflec¬ 
tion for signals and downward deflection for the mark¬ 
er. However, the use of different directions of deflec¬ 
tion for signals and marker results in somewhat poorer 
scanning efficiency. Some of the scans (perhaps 25 per 
cent) must be devoted to scanning the marker alone. 
This means that the band scanned must be reduced 
(perhaps 25 per cent) if the same resolution is to be 
obtained.* 1 

With a receiver of the heterodyne type, the fre¬ 
quency of the beating oscillator differs from the input 
frequency by the intermediate frequency. Accordingly 


d Other testsi were made in which the marker, a manually 
controlled signal used to identify an unknown signal, was 
scanned alternately with the signals received from the an¬ 
tenna. This permitted the marker to be adjusted much closer 
to the exact frequency of the unidentified signal but was not 
an appreciable aid in tuning in the latter on the monitoring 
receiver. In many applications this arrangement has the ob¬ 
jection that it results in either fewer indications per second, 
narrower band scanned, or poorer resolution. In applications to 
frequency matching for “jammers” the additional accuracy 
far outweighs the objections. 



Figure 16 . Derivation of marker by double-modulation method. 




















































82 


THE FUNDAMENTALS OF PANORAMIC RECEPTION 


a frequency corresponding to the receiver tuning may 
be produced from the beating-oscillator frequency. 
There are two general methods: 

1. A separate oscillator ganged to the beating oscil¬ 
lator and differing in frequency by the intermediate 
frequency. The difficulty in this case is to obtain ac¬ 
curate tracking over the entire frequency range. 

2. Electric frequency derivation. If this is done by 
combining the beating oscillator and the intermediate 
frequency in a mixer, unwanted frequencies are pres¬ 
ent close to the desired frequency. Methods of elimi¬ 
nating or avoiding such unwanted frequencies are as 
follows: 

a. Ganged tuning. 

b. Use of double modulation as illustrated in 
Figure 16. 

c. Use of automatic frequency control as illus¬ 
trated, for example, in Figure 17. 

d. Some sort of single side-band modulation 
scheme using phase balance or a combination 
of amplitude and frequency modulation might 
be possible. 



L-J 


HETERODYNE 

RECEIVER 

Figure 17. Derivation of marker by automatic-frequency- 

control method. 

128 BLANKING OUT RECOGNIZED 
STATIONS 

When the number of signals appearing on a pano¬ 
ramic indicator is large, observation can be facilitated 
if identified signals are blanked out in some manner 
so that the sudden appearance of a new signal can be 
readily detected. There are several ways of doing this. 

Rejection or trap circuits, preferably adjustable, 
might be used ahead of the scanning oscillator to sup¬ 
press recognized signals. If a number of different sig¬ 
nals are to be suppressed, this scheme involves con¬ 
siderable complication. Also it would be difficult to 
obtain sufficiently sharp discrimination to avoid at 


least partial suppression of frequencies on either side 
of the desired signal. The scheme may be useful, how¬ 
ever, for suppressing a small number of high-level 
stations. 

Another way of blanking out recognized stations 
would be to generate a number of different blanking 
pulses, adjustable in time position and width, which 
could be applied directly to the cathode-ray tube. This 
likewise involves considerable complication. 

A simple scheme which has been used with the in¬ 
tensity type of indicator is to apply black paint to the 
face of the tube to cover up identified signals. With a 
suitable paint the screen can be wiped clean at inter¬ 
vals as desired. 

A scheme which has been suggested for blanking 
out is to record incoming signals on a continuous 
magnetic tape and utilize recorded signals to balance 
out incoming signals so that only a signal not pre¬ 
viously recorded would appear on the screen. This 
scheme appears to involve certain practical difficul¬ 
ties. An alternative scheme which might have advan¬ 
tages would be to use a magnetic tape on which blank¬ 
ing signals are produced by a local signal generator. 

Blanking schemes necessarily involve some sacrifice 
in resolution in the blanking region. A further disad¬ 
vantage of any blanking-out scheme is that an identi¬ 
fied signal may disappear and be replaced by an un¬ 
identified one without the observer being aware of this. 
Also, when blanking out is used a low-powered signal 
which is adjacent to a high-powered known signal may 
pass unobserved, whereas otherwise it might have been 
noted during idle periods of the larger signal. 

12.9 NARROW-BAND SCANNING — RANGE 
EXPANSION — COMBINED SCANNING 

A panoramic arrangement for scanning a relatively 
narrow band on either side of the tuning frequency 
has been found to be a useful attachment for an ordi¬ 
nary heterodyne receiver. 15 The arrangement is illus¬ 
trated schematically in Figure 18. The input to the 
panoramic unit is derived from the i-f output of the 
mixer, which should be reasonably flat over the sweep 
range. The signals from the mixer circuit are swept 
past a scanning filter by means of a beating oscillator 
which is swung through a range of about ±50 kc. 
The diagram normally is in the form of signal ampli¬ 
tude versus frequency. 

A narrow-band scanner of this kind may be com¬ 
bined with a wide-band panoramic receiver to obtain 
both a low-definition wide-band and a high-definition 































RECORDING 


83 



HETERODYNE 

RECEIVER 


panoramic 

ADAPTER 


Figure 18. Schematic of panoramic adapter. 


narrow-band diagram. The narrow band may be moved 
manually or automatically over the wide band. 

A combination wide- and narrow-band diagram may 
be provided on a single cathode-ray tube, the narrow 
band representing merely an expansion of a certain 
part of the wide band. When the resolving power of 
the c-athode-ray tube itself is the limiting factor, bet¬ 
ter resolution is obtained merely by expanding a part 
of the frequency scale, as shown, for example, in Fig¬ 
ure lb for a single-line frequency diagram. The wave 
used for horizontal deflection, instead of being merely 
a simple sawtooth, is a combination of a sawtooth and 
a stepped wave with a steep slope between steps. The 
position of this slope would be adjustable so that any 
part of the frequency range could be expanded. 

The above scheme is of little value if the resolution 
is limited, not by the indicator, but by the scanning 
filter. In this case it is necessary to obtain greater 
scanning resolution in the narrow band. The best way 
to get this is to use separate wide- and narrow-band 
scans. Results may be displayed as two diagrams, the 
wide- and narrow-band pictures may be combined on 
one screen. With the arrangement of Figure 20, alter¬ 
nate wide- and narrow-band scans may be shown on 
separate lines of the cathode-ray tube. Instead of using 
two separate scans, it would theoretically be possible to 
have a single wide-hand sweep with some arrangement 
for slowing down and introducing a narrower filter 
over a small part. The practical value of such a scheme 
is doubtful. 


One way of improving the resolution of a panoramic 
receiver is to provide a number of narrow-band scan¬ 
ners positioned side by side. The equivalent of this can 
be had, without complete duplication of equipment, 
by providing a number of filters spaced apart over a 
wide band and sweeping the wide band through a fre¬ 
quency interval corresponding to the spacing between 
filters. The outputs of the several filters would then be 
scanned at a rate which is a large multiple of the rate 
of input frequency swing. The method is illustrated 
schematically in Figure 21. To produce a single-line 
diagram of signal presence versus frequency a com¬ 
bination wave consisting of a sawtooth corresponding 
to the input swing and a stepped wave corresponding 
to the filter-output scan would be used for horizontal 
deflection of the cathode-ray beam. Signals might be 
indicated by beam deflection or beam modulation. As¬ 
suming a given total band, the increase in resolution 
obtained by dividing up with filters according to this 
method increases as the square root of the number of 
filters. On the other hand, if a given band can be scan¬ 
ned with a certain resolution using one filter, this band 
increases directly with the number of filters. 

12.10 RECORDING 

Strictly speaking, a recording receiver is not a pano¬ 
ramic receiver and therefore lies outside the scope of 
this report. Application of recording to receivers of 
the panoramic type will, however, be briefly con¬ 
sidered. 


















































84 


THE FUNDAMENTALS OF PANORAMIC RECEPTION 


SIGNALS 




70 FREQ 
SWEEP 




(<x) SAWTOOTH 


__/- 1 


(b) EXPANDED SWEEP 



(c) HOR DEFLECTING WAVE=(a)-KM 


Figure 19. Method of expanding part of frequency scale. 

Principal interest probably resides in facsimile-type 
recording where a number of signal patterns for sta¬ 
tions of different frequency are arranged side by side. 
For this it is necessary to have a moving record acti¬ 
vated by a light (or electronic) source whose intensity 
is modulated to indicate signals and which is moved 
transversely across the record in synchronism with the 
scanning of the frequency range. A suitably high-speed 
facsimile recording method (electrolytic, electrother¬ 
mal, photographic, etc.) may be employed. The receiv¬ 
ing equipment proper may be the same as for a pano¬ 
ramic receiver. 

Although a panoramic facsimile recorder avoids the 
limitations of the eye which have been noted for a 
facsimile type of indicating system, some important 
limitations remain. If, as is usually the case, it is de¬ 


Figure 20. Combined wide- and narrow-band diagrams. 

sired to resolve telegraph code, it is necessary to em¬ 
ploy a fairly high repetition rate for the scanning. The 
band width that may be scanned varies inversely with 
the number of words per minute transmitted in the 
highest speed signal to be recorded. Conversely, the 
minimum allowable frequency separation between sig¬ 
nals to be recorded increases as the square root of 
either the number of words per minute of the highest 
speed signal or the band width to be scanned. A re¬ 
petition rate of about 60 cycles is desirable for hand 
telegraph speeds and correspondingly higher rates for 
higher speeds. As already discussed, the use of a high 
repetition rate impairs frequency resolution. Further¬ 
more, the speed of the moving record is determined by 
the resolution desired along the time axis, which again 
is dependent on telegraph speed. A speed of the order 
of 1 in. per second is probably suitable for hand tele¬ 
graph speeds. One inch per second means I V3 miles of 
record per day, so that analysis of the record becomes 
difficult. If, in view of the limited frequency band that 
can be covered by a single recorder, a number of re¬ 
corders are employed for adjacent frequency bands, 
the problem of record analysis is correspondingly mul¬ 
tiplied. 

It has been proposed that length of record be re¬ 
duced by employing a combination of panoramic indi¬ 
cator and recorder, making a record only when signals 
of interest are observed on the indicator. This requires 
that the signals be stored long enough to permit a 


CONFIDENTIAL 









































PANORAMIC PULSE RECEPTION 


85 



CRT 


COMPONENTS OF HORIZONTAL DEFLECTING WAVE: 




Figure 21. System with multiple scanning filters. 


decision as to the desirability of recording. Such stor¬ 
age may be obtained by recording tire signals on a 
magnetic tape. 

12.11 panoramic pulse reception 

Panoramic reception of radar system pulses differs 
in certain respects from reception of ordinary signals. 
Tn the first place, the frequency spectrum of a high- 
frequency pulse is quite broad . 16 For practical pur¬ 
poses most of the energy of the pulsed high frequency 
may be assumed to lie within a band centered at the 
carrier frequency and having a width equal to twice 
the reciprocal of the pulse length. Thus the band for 
a 1 -p.sec pulse would he approximately 2 me. 

It is desirable that the scanning filter be as wide as 
the pulse signal band as roughly defined above. This 


is for two reasons: First, this affords good signal-to- 
noise ratio; and second, it permits stopping the re¬ 
ceiver and viewing pulse shape. Since the resolution is 
limited in any case by signal band width, no sacrifice 
of resolution results from making the filter band at 
least as wide as the signal band. 

If the width of the scanning filter is made equal to 
the pulse signal band, then it turns out that for prac¬ 
tical repetition rates and total frequency ranges the 
scanning speed is no longer a limitation on resolution. 
Thus, for example, with a 2 -mc filter the scanning 
speed (product of repetition rate and total frequency 
band) may he of the order of 10 12 cycles per second. 

A further factor in scanning reception of pulse sig¬ 
nals is the low duty cycle, i.e., the fact that the signals 
are present only a very small part of the time. In the 


CONFIDENTIAL 





































































86 


THE FUNDAMENTALS OF PANORAMIC RECEPTION 


processes of scanning reception a number of pulses are 
lost, the fraction received being equal to the ratio of 
tbe width of scanning filter to the total frequency hand 
covered. If, in addition to frequency scanning, the re¬ 
ceiver employs azimuth scanning or if the source of 
the pulse signals employs azimuth scanning, the frac¬ 
tion of the original pulses received is much further 
reduced. 

Pulses must be present during a sufficient percent¬ 
age of the time to actuate the panoramic indicator. 
The minimum band width for the scanning filter to 
satisfy this condition may be stated as 

( 5 ) 

where B is the hand width, F is the total frequency 
band, N is the pulsing rate, and K 7 is the number of 
pulses that must be received per second to actuate the 
indicator. If, for example, the total band to be swept 
were assumed to be 400 me, if the pulsing rate were 
400 per second and if 10 pulses per second were neces¬ 
sary to actuate the indicator, then the scanning filter 
should he 10 me wide. While a complete discussion of 
these factors is beyond the scope of this report, it is 
evident that if very wide frequency bands are to be 
scanned, it becomes necessary to make the scanning 
filter considerably wider than the band width of the 
pulse signal. Also it is clear that scanning reception 
in combination with azimuth scanning in the receiver 
or in the transmitter becomes extremely difficult. 

The signal-to-noise ratio, and hence the realizable 
sensitivity, of a scanning receiver for radar pulses is 
poorer than that of the same receiver for continuous 
narrow-band signals of the same field strength. There 
are two reasons for this. First, the pulse receiver is 
responsive to noise all the time while the pulses are 
present only a small part of the time. Secondly, the 
received noise power varies directly with band width, 
so that the noise is correspondingly greater in receiv¬ 
ing wide-hand pulse power than in receiving the same 
power concentrated in a narrow band. 

1212 PANORAMIC RECEIVERS WITHOUT 
FREQUENCY SWEEP 

Frequency sweep, which has formed the basis of a 
large part of the material in the foregoing sections, is 
not essential for panoramic reception. An alternative 
is to provide a selective device or devices whereby the 
input signal band is separated into small component 
bands which are applied in succession to a visual in¬ 


dicator. The selection may be accomplished by a series 
of filters having a common input and separate out¬ 
puts. The filter bands may be made adjacent, con¬ 
tiguous, or slightly overlapping as desired. These out¬ 
puts may he scanned in rapid succession and resultant 
signals used to provide the indicator diagram. A pos¬ 
sible arrangement is shown schematically in Figure 
22. A method of this kind has been used in studying 
the effect in a short-wave radio-telephone channel of 
changes in the transmission medium 17 and numerous 
subsequent applications have been made. 18 The same 
general method may be employed for facsimile or 
other kinds of recording. 

The outputs of the multiple selecting circuits may 
be scanned at practically any desired rate if amplifiers 
or other circuits which follow this scanning are made 
wide enough. Consequently, by eliminating the fre- 
quency sweep the problems inherent in the scanning 
filter are avoided. To provide a sufficient number of 
filters to cover the desired frequency band with ade¬ 
quate resolution may, however, involve considerable 
complexity of equipment. 

12.13 USES OF PANORAMIC RECEIVERS 

Listed below are possible types of uses, both mil¬ 
itary and nonmilitary, for panoramic receivers. AVhile 
considerable information is available regarding pos¬ 
sibilities of panoramic receivers for some of these ap¬ 
plications, much further study and experience will be 
necessary for a complete appraisal. 

Intercept. Panoramic reception may serve as an 
adjunct to aural reception in different types of inter¬ 
cept work, including particularly intelligence and traf¬ 
fic studies. It may supplement, or possibly avoid, con¬ 
tinuous aural searching. Wide- or narrow-band scan¬ 
ning may he used to provide information on signal 
presence, signal frequency, signal characteristics, dura¬ 
tion of transmission, etc. 

Communication. In communication channels or 
networks, panoramic reception may be used in adjust¬ 
ing the frequency of transmitters to assignments 
which are suitable from the standpoint of other signals 
and interference. Where transmissions are to be re¬ 
ceived on different frequencies over a wide band, a 
panoramic receiver could function as a call indicator. 
Narrow-band scanning would facilitate detection of, 
and receiver adjustment for, moderate variations in 
transmitter frequency. Panoramic reception might he 
used to detect interference or jamming by the enemy, 


1 


OXFIDENTIAL 


k 






USES OF PANORAMIC RECEIVERS 


87 



Figure 22. Multifilter system without frequency sweep. 


or other interference present in different parts of the 
spectrum. 

Jamming. Both broad- and narrow-hand scanning 
are useful in studying enemy signal channels and sig¬ 
nal characteristics and in the adjustment of jamming 
transmitters. 

Direction Finding. Panoramic reception permits 
simultaneous direction finding for a number of sig¬ 
nals of different frequency. 

Radio Navigation. Panoramic receivers may be used 
in a variety of ways for radio navigation . 8 A simple 
application would be to match the amplitudes of two 
frequencies constituting a range beacon. More elabo¬ 


rate schemes include the possibility of beacons varied 
simultaneously in azimuth and frequency with a pan¬ 
oramic indicator to show the azimuth of the beacon. 
By suitable proportioning of the azimuth-frequency, 
characteristic, navigation courses for moving craft 
can be charted in vertical, horizontal, or inclined 
planes. 

Instrument Landing. A panoramic receiver might 
perform the functions of runway localizing, runway 
marking, and glide-path indication . 8 

Transmission Studies. Panoramic reception may 
he used for studies of frequency spectra, radio noise, 
fading, etc. 


A 


.'OAPIDPNTIAL 


























Chapter 13 

PANORAMIC RECEIVER WITH MOVING-SCREEN INDICATOR 


Receiver utilizing a special cathode-ray tube in which the 
screen is rotated so that the information it conveys can be 
spread out in a three-dimensional pattern, with frequency 
shown vertically, time horizontally, and intensity by bril¬ 
liance of the trace. 

13-1 INTRODUCTION 

hex this project 21 was started there was a 
great deal of dissatisfaction with the panoramic 
reception methods then available and NDRC was 
asked (especially by the Navy) to investigate new 
methods of panoramic presentation. In the existing 
systems, the presentation was in the form of a rec¬ 
tangular coordinate pattern with frequency shown 
horizontally and intensity vertically. Radio signals on 
different frequencies appeared as vertical pips along 
the horizontal frequency scale. A serious limitation 
was the difficulty in detecting brief radio signals. 

The purpose of Project C-27 was to apply to the 
panoramic problem a new type of cathode-ray tube 
with a moving screen. 1 In this manner the information 
could be spread out in three dimensions with frequency 
shown vertically, time horizontally, and intensity by 
trace brilliance. Thus a telegraph signal would appear 
as an actual dot-and-dash trace across the screen and 
a-m telephone signals as a variable-intensity tone. 
The whole pattern of radio signals within a certain 
frequency interval and space of time would be spread 
out like the pattern on a strip of fabric coming off a 
loom. Brief signals would be seen as small details in 
this pattern and different kinds of signals should be 
easily recognizable. 

A model was built, all parts of which were developed 
specially under this project except the cathode-ray 
tube, work on which had begun for other purposes. 

13.2 RESULTS OBTAINED 

In the diagrams observed on the screen, different 
types of signals were clearly discernible. For telegraph 
stations sending 40 words per minute or less, the code 
was clearly reproduced and could be read from the 
screen. Fast machine-sent telegraph signals could be 

“Project C-27, Contract OEMsr-49, Bell Telephone Labora¬ 
tories, Inc., Western Electric Co., Inc. 


recognized as telegraph but could not be deciphered. 
Under good conditions it was possible to observe, in 
the case of speech-modulated signals, a trace having 
serrated edges indicative of the varying frequency 
limits of the double side band and light and dark 
striations corresponding to the syllabic modulation. 

The frequency resolution was such as to permit 
from 60 to 80 different stations with uniform spacing 
to be distinguished on the screen. By sweeping a nar¬ 
rower band, signals very close together in frequency 
could be spread apart. 

Thus, the moving-screen receiver overcame much of 
the difficulty experienced with existing conventional 
receivers where the picture on the screen was an in¬ 
stantaneous diagram of amplitude versus frequency, 
in which the vertical spikes representing different 
radio stations constantly move up and down because 
of modulation, fading, noise, etc. Such a diagram is 
very trying to monitor and signals of short duration 
are difficult to detect. 

13-3 GENERAL PRINCIPLES 

A block schematic of the receiver is shown in Fig¬ 
ure 1. Signals received from the antenna in the range 
2 to 10 me are separated by means of a band filter 
from signals in other parts of the spectrum. After two 
stages of r-f amplification, they are heterodyned to 
an intermediate frequency of 40 me by means of a 
sweep oscillator which is varied by a reactance tube. 
Unwanted products of the first modulator are elimi¬ 
nated by a filter having a pass band of 40 me ±100 
kc. The signals are then heterodyned with a fixed car¬ 
rier of 40.548 me to bring them to the second inter¬ 
mediate frequency of 548 kc where they are selected 
by a band filter. Four different widths of band filter, 
1, 5, 10, and 25 kc, are available on a switching basis. 
The signals then pass through a high-gain amplifier 
which compresses a wide range of input signal ampli¬ 
tudes to a very narrow range in the output. After 
further amplification, the signals are applied to the 
modulating grid of the cathode-ray tube which is 
biased so that the beam is turned on only when signals 
are present. 

The frequency of the first beating oscillator is varied 



CONFIDENTIAL 



88 






DESCRIPTION OF COMPONENTS 


89 


ANTENNA 



Figure 1. Block diagram of 2- to 10-mc panoramic scanning receiver with moving screen indicator. 


by means of a reactance tube control and sawtooth 
generator, with provision for setting the sawtooth fre¬ 
quency at 10, 20, 30, or 60 per second. In addition to 
regulating the f-m sweep, the sawtooth wave is also 
applied to a pair of coils which deflect the cathode- 
ray beam in a vertical direction across the screen. 

Rotation of the tube gives a diagram in which time 
is the horizontal dimension. Traces corresponding to 
the incoming signals are produced by persistence of 
the fluorescent screen. These traces appear at differ¬ 
ent vertical positions according to the frequency of the 
radio stations. 

In the model receiver the range swept over by the 
f-m oscillator could be adjusted in steps from 8 me 
down to 0.05 me and arrangements were available for 
positioning the sweep range at different locations in 
the 2- to 10-mc range. 

A Hammarlund 200-SPR receiver was incorporated 


in the model to permit aural reception of any train of 
signals appearing on the screen. In conjunction with 
this receiver there was provided an arrangement for 
producing on the screen a marking trace of known 
frequency. Thus the Hammarlund receiver could be 
quickly tuned to any desired trace and the frequency 
of that trace read on the receiver dial. 

13 4 DESCRIPTION OF COMPONENTS 

In the following paragraphs the individual compo¬ 
nents making up the model receiver will be described, 
together with their characteristics. 

Input Attenuator 

To take care of extremely high signal strengths at 
the input, a two-step 7r-type attenuator designed for 
a 75-ohm unbalanced circuit provided losses of 0, 
20, or 40 db. 


CONFIDENTIAL 



















































































90 


PANORAMIC RECEIVER WITH MOVING-SCREEN INDICATOR 


2- to 10-mc Filter 

This is an electric filter designed to provide not less 
than 35 db attenuation below 1,600 kc and above 
12,000 kc, with a mid-band loss of 0.3 db and less than 
1.5-db distortion over the band. The filter operates 
between 75-ohm unbalanced impedances. 

R-F and Modulator Circuits 

The r-f amplifier consists of two broad-band coupled 
stages using 6AC7 tubes. The output of the amplifier 
is applied to the control grid of the first modulator, 
which also uses a 6AC7 tube. The output of the first 
modulator is applied to the 40-mc filter, thence to the 
grid of the second modulator tube, also a 6AC7. This 
modulator operates into a balanced impedance of 125 
ohms. 

A buffer amplifier isolates the first modulator from 
its carrier oscillator. The circuit gain from the 2- to 
10-mc filter input to the second modulator output is 
17 ± 0.3 db. 

40-mc Band Filter 

To meet the requirements for this filter, two differ¬ 
ent models were built, the second of which is somewhat 
superior. The mid-band insertion loss at 40 me is less 
than 10 db and the distortion over the pass band of 
39.9 to 40.1 me is less than 5 db. Discrimination at 
frequencies of 40 ± 0.3 me is greater than 20 db and 
increases for frequencies farther from the pass band. 

The filter consists of small mica and air condensers 
and lengths of coaxial line which furnish the high-$ 
inductances required for the narrow pass band. 
Through an impedance transformation, the inductance 
values and hence the lengths of the line sections are 
made identical. 

First Oscillator 

The first oscillator delivers a carrier frequency which 
varies in accordance with the amplitude of the input 
sawtooth voltage. It consists essentially of a Hartley 
oscillator which is frequency-modulated by a reactance 
tube. Figure 2 shows the fundamental principle in¬ 
volved. The grid of the control tube T 1 is supplied 
with a voltage E RF which leads the tank voltage Et 
by 90 degrees. This voltage produces in the plate cir¬ 
cuit a current which leads the tank circuit voltage 
Et by approximately 90 degrees. Hence the control 
tube acts like a capacitor shunting the oscillator tank 
circuit. The capacitance depends on the plate current 
/,, which in turn is a function of the transconductance 


of the control tube. Thus by varying the control volt¬ 
age E c the frequency is varied. 

The input control voltage consists of two compo¬ 
nents whose amplitude can be varied independently. 
One of these is a d-c voltage obtained from a well- 



Fi gure 2. Reactar.ce tube rr.cdulator used as sweep oscillator. 

regulated power supply. The other is a sawtooth volt¬ 
age obtained from the sweep amplifier. Adjustable at¬ 
tenuators make it possible to change the width of the 
frequency band swept over and the mid-frequency 
position of this band. The frequency band covered can 
be adjusted to nominal values of 8, 4, 2, 1, 0.5, 0.2, 
0.1, or 0.05 me, and these bands can be positioned any¬ 
where in the range 30 to 38 me. 

The diagram of Figure 3 illustrates the operation 
of the f-m oscillator. The curve of frequency versus 
control voltage approximates a straight line. Thus 
if the input control voltage consists of a d-c compo¬ 
nent of 6 volts and a sawtooth component of 4 volts, 
then the frequency of the oscillator is swept through 
a band having a width of approximately 2 me and a 
mid-frequency of 35 me. 

The oscillator has a self-contained buffer amplifier 
which delivers approximately 1.5 volts rms to the grid 
of the buffer amplifier. Over the frequency range from 
30 to 38 me the output level is flat to within about 
1 db. 

Second Oscillator 

This oscillator employs two 6AG7 tubes. The fre¬ 
quency is controlled by a bridge which is balanced for 
all frequencies except that of a 40.548-mc crystal, so 
that feedback for oscillation is available at that fre¬ 
quency. A buffer amplifier delivers about 2 volts to 
the grid of the second modulator tube. 

548-kc Band-Pass Filters 

By means of a switch, selection may be made of 
any one of four filters having band widths of 1, 5, 10, 
and 25 kc, respectively, all centered at 548 kc. The 
two narrowest filters employ crystal elements, while 


CONFIDENTIAL 

























DESCRIPTION OF COMPONENTS 


91 



the other two use electric elements only. The terminat¬ 
ing impedance of the filters is 125 ohms balanced to 
ground. Distortion over the filter band in each case 
is not greater than 3 db and the mid-band loss does not 
exceed 6 db. In the attenuating regions the loss rises 
rapidly to a discrimination not less than 30 db at 548 
kc ± 2f b where f b = band width. A shielded unbal¬ 
anced 125 :125-ohm transformer is connected between 
the balanced filters and the unbalanced amplifier 
which follows. 

548-kc Compression Amplifier 

The compression amplifier consists of three stages 
made up of a 6AC7 (or a 6SJ7) and a 6116 rectifier. 
It receives signals from the 548-kc band filter which 
may vary in amplitude from 86 to 26 db below 1 volt. 
Its output is delivered across a 200-ohm load to the 
modulating amplifier. 


maximum compression, an input level range of 60 
db may be compressed to an output level range of 
4 db. Both the gain and the compression are variable, 
the gain over a range of about 20 db and the compres¬ 
sion over a range of about 60 db. 

Band-pass interstage impedances for the first three 
stages are about 25,000 ohms at low levels. When the 
level across each interstage increases to a value that 
makes the biased diodes conduct, the interstage im¬ 
pedance drops to about 1,000 ohms (the conductance 
of the diodes). The limiting action produced by the 
diodes successfully cutting in as the level across the 
interstage increases will be understood from reference 
to Figure 4. The levels shown in Figure 5 are for a 
typical setting of gain and compression. 

The compression is instantaneous. Whenever the 
voltage across any interstage exceeds the diode cutoff 
voltage, the wave can no longer rise in amplitude. 
Both positive and negative halves of the wave are 


This amplifier has a maximum gain of 80 db. With 

[ CONFIDENTIAL 






















92 


PANORAMIC RECEIVER WITH MOVING-SCREEN INDICATOR 




in the cathode-ray tube. The bias is normally adjusted 
to approximately this value by the intensity control. 
The modulating amplifier is resistance-capacity cou¬ 
pled to the cathode-ray grid and its action varies 
the grid voltage below and above the cutoff value. To 
provide ample margin for discrimination against 
noise, the modulating amplifier is designed to produce 
approximately 140 volts peak. 

Two high-gain pentode stages are used. The first 
is an inductance - compensated resistance - capacity 
coupled stage using a 6SJ7 tube. The second stage 
employs a 6AG7 high-£r m beam tube with a broadly 
tuned resonant output circuit which is essentially 
flat over the widest band received from the 548-kc 
filter. Cathode and screen by-passing and the inter¬ 
stages are designed to reduce the gain sharply outside 
the pass band. 

Cathode-Kay Tube 


D E 

Figure 4. A, simplified schematic of clipping stage; B, 
input wave; C, output wave without clipping; D, output 
with clipping; E, output after clipping and amplification 
through band-pass amplifier. 

clipped by the double-diode arrangement. After being 
distorted by clipping, the wave becomes sinusoidal 
again when it is amplified by following band-pass 
amplifier stages. 


The appearance of the cathode-ray tube is shown 
in Figure 6. The electron beam is emitted vertically 
from a gun in the lower neck of the tube, passing a 
control grid which regulates the intensity of the beam. 
The spherical part of the tube is girdled by a fluores¬ 
cent horizontal band 5 in. wide. The upper and lower 
portions of the sphere are covered by Aquadag to 
which is connected the anode potential of 5,000 volts. 
Rv means of beam-bending coils a stationary horizon- 



40 08 GAIN MAX 


20 08 GAIN MAX 


I I 

-20 08 GAIN MAX -t^t-6 08 GAIN** 





1 

AMP 1 


LIMIT 1 


AMP XI 


limit n 


amp m 


LI MIT HI 


AMP IX 

' + 

r 

6AC7 


6H6 

1 

6SJ7 


6H6 

T 

i 

€SJ7 


6H6 

T 

1 

6 AC7 


H 


200 

OHMS 


86 TO 

-46 TO 

66 DB 

-26 08 

66 TO 

-26 TO 

46 08 

- 6 08 

46 TO 

-6 TO 

26 08 

-4 DB 


-26 TO 

-6 TO 

- 6 DB 

-4 DB 

-6 TO 

-4 TO 

-4 OB 

-3 08 

-4 TO 

-3 TO 

-3 08 

-2 08 


Figure 5. Operation of compression amplifier in schematic form. 


Modulating Amplified 

The modulating amplifier increases the output of 
the compression amplifier to a value sufficient to com¬ 
pletely modulate the electron beam of the cathode-ray 
tube. It receives from the compression amplifier ap¬ 
proximately 0.75 volt peak across an input impedance 
of 0.5 megohm. A grid bias of approximately — 90 
volts is required to completely cut off beam current 


tal electromagnetic field bends the electron beam into 
an approximately horizontal direction so that it falls 
on the fluorescent screen. A pair of deflection coils 
arranged in the form of a yoke is mounted around the 
upper end of the lower neck of the tube. When a saw¬ 
tooth wave with a frequency of 10, 20, 30, or 60 cycles 
is passed through the deflection coils, the beam is 
caused to swing back and forth in the tube neck. This 
results in a vertical trace on the fluorescent screen. 


CONFIDENTIAL 





















































DESCRIPTION OF COMPONENTS 


93 


The operating potentials are delivered to the tube 
through slip rings. 



Figure 6. View of moving screen receiver showing 
respective positions of cathode-ray tube and screen. 


Since the beam-bending coils and the deflection 
coils are held stationary, the electron beam always 
lies in a fixed vertical plane while the tube is rotated. 
Thus a continuously changing screen surface is pro¬ 
vided to serve as a time axis. Since the deflection of 
the beam is in synchronism with the sweep of the in¬ 
coming signals, a spot is obtained at each point in the 


vertical plane of the screen for which a signal occurs 
at that instant in the corresponding position in the 
frequency spectrum. The next trace of the sawtooth 
oscillator finds a fresh surface on the screen, while the 
original mark persists. 

The tube has a diameter of 10 in. It is rotated at a 
rate of 1 revolution in 30 seconds, thus producing ap¬ 
proximately 1 in. of horizontal movement of the screen 
per second. The persistence of the screen is sufficient 
so that the decay during the 5 seconds corresponding 
to the viewing aperture is not marked. While observa¬ 
tions have been made in a very dark room of signals 
persisting on the screen for as many as 5 to 10 revo¬ 
lutions, i.e., 2 V 2 to 5 minutes, the decay characteristic 
is such that signals coming around after one revolu¬ 
tion of the tube do not interfere appreciably with the 
observance of the signal diagram. 

Sawtooth Generator and Associated Amplifiers 

The sawtooth generator produces a sawtooth wave 
of 10, 20, 30, or 60 cycles which is amplified and ap¬ 
plied through the deflecting coils of the cathode-ray 
tube. The sawtooth voltage is also used to control 
the frequency of the first carrier oscillator. 

The sawtooth generator employs two 6AC7 tubes in 
a direct-coupled trigger-tube circuit. One tube charges 
a parallel EC circuit which discharges. When it has 
discharged sufficiently it actuates the trigger tube 
which delivers a sharp pulse to the charging tube 
which brings the condenser back to its original volt¬ 
age. The constants of the RC circuit are chosen so as 
to provide constant voltage output and maximum 
sweep linearity. A small amount of 60-cycle voltage 
is injected into the trigger-tube circuit to synchronize 
the sweep with the power-line frequency. 

Adjustment of sweep rate is accomplished by a 
switch. A fine frequency control adjusts the EC dis¬ 
charge circuit to synchronization. 

The output of the sawtooth generator is applied to 
a buffer amplifier which consists of a single 6SJ7 
stage. This amplifies the sweep output voltage of five 
volts to approximately 20 volts peak to peak. 

The buffer amplifier energizes a 6AG7 cathode fol¬ 
lower which delivers 15 volts peak to peak to a 1,000- 
ohm load which supplies the sawtooth wave to control 
the first carrier oscillator. 

The output of the buffer amplifier also energizes 
the current amplifier which supplies the sawtooth wave 
to the deflection coils. A current swing of 100 ma is 
required for full sweep across the screen of the catli- 


(OXFIDENTIAI. 






































94 


PANORAMIC RECEIVER WITH MOVING-SCREEN INDICATOR 


ode-ray tube. The deflection coils are direct coupled 
to the 6L6 output stage, the d-c component being 
balanced out. The plate load consists of inductance 
and resistance whose time constant is made equal to 
that of the deflection coils. Vernier controls are pro¬ 
vided to adjust deflection yoke bias and the time con¬ 
stant of the plate circuit. 

Marking Arrangement 

In the marking arrangement used on the model, the 
heterodyne frequency provided by a Hammarlund 
200-SPR receiver, which is 465 kc above the fre¬ 
quency to which the receiver is tuned, is modulated 
with a carrier of 465 kc. The output of the modulator 


tion is within ±0.05 volt. Over the usual working 
range of input voltage and load, the regulation is of 
the order of ±0.02 volt. The circuits consist of the 
usual type of rectifier with a capacitance input filter. 
The unusual degree of regulation is achieved by a spe¬ 
cial feedback amplifier working into two or more 
paralleled current valves or control tubes, the number 
depending on the requirements. Any small change in 
output voltage produced by a change in input voltage 
is amplified by the 6SF5 feedback amplifier tubes 
shown in Figure 7. The resulting amplified variation 
applied to the grids of the 6Y6G current valves varies 
their impedance, thus compensating the voltage change 
in the output. 



Figure 7. Circuit diagram of voltage-regulated power supply for 250-volt, 300-ma output. 


is applied to the input of the 2- to 10-mc filter, thus 
producing on the screen three marking frequencies, 
the lowest one of which is the one to which the Ham- 
marlund receiver is tuned. When this lowest marking 
line is made to coincide on the screen with any ob¬ 
served signal train, the frequency of that train is 
determined and the signals are received aurally. A 
different marking circuit was developed subsequently 
and is described below. 

Power Supplies 

All the 300-volt anode supplies are closely regulated 
to compensate changes in input voltage and variations 
in the load. Over a range of 105 to 125 volts a-c input 
and a 30 per cent range of output current, the regula¬ 


rs RESEARCH WORK 

Scanning Oscillator 

Considerable research was required to develop the 
scanning oscillator, since no previous knowledge was 
available regarding the design of an electronically 
controlled oscillator with such a Avide percentage sweep 
at such a high frequency. Extremely meticulous de¬ 
sign was required to obtain satisfactory stability. 

In the course of work on the scanning-oscillator 
problem an alternative involving the use of an RC 
oscillator was studied. In this the frequency would 
be varied by changing the bias of a diode to vary the 
resistance of the oscillator circuit. It appeared that 









































































FURTHER WORK 


95 


an oscillator of this type might be used at a frequency 
higher than previous RC oscillators but in view of 
the constants involved in operation at 30 to 38 me 
it was planned to use a lower fundamental frequency 
swing in the range 15 to 20 me and select out the 
second harmonic for the desired variable carrier. A 
circuit was designed which appeared capable of provid¬ 
ing a wider percentage frequency swing than the re¬ 
actance-controlled oscillator. Difficulties were experi¬ 
enced, however, in obtaining stability, and although 
it appeared that these could be solved this approach 
was abandoned when the reactance - tube scheme 
proved successful. 

Some study was also made of the application of 
feedback to the present sweep oscillator but this was 
not carried to completion. 

Compression Amplifier 

Study was required of different methods of handling 
a wide range of input signals. This indicated that the 
preferred method would be to use a compression ampli¬ 
fier. Since an amplifier of this type with such a high 
compression range had not previously been built, re¬ 
search was required to determine the best circuit 
design. 

Infrared Wiping Out 

Originally it appeared that it would be desirable to 
wipe out the traces on the screen after passing the 
viewing window, so that these traces after completing 
one revolution would not interfere with the new traces. 
Some work was done with an arrangement employing 
infrared rays to obtain such wiping out. The infrared 
was obtained from ordinary lamps. With the filtering 
used, the amount of light in the lower part of the vis¬ 
ible spectrum appearing on the screen seemed to be 
somewhat detrimental to observation. Although it 
appeared that a satisfactory wiping-out arrangement 
could be secured, further work was abandoned when 
it was found that the traces persisting after one revo¬ 
lution did not interfere appreciably with observation. 

13.6 FURTHER WORK 

After demonstrating the preliminary model, cer¬ 
tain improvements and modifications were suggested, 
so that studies of microsecond pulses in the frequency 
range 500 to 600 me could be made. 

ddie frequency-marking arrangement, whereby the 
receiver is tuned to any signal appearing on the 


screen, was modified so that only one marking fre¬ 
quency appears on the screen. The new arrangement 
employs an additional tuning unit of the same type 
as that used in the Hammarlund receiver. This unit is 
modified so as to produce a frequency which differs 
by 465 kc from the heterodyne frequency of the re¬ 
ceiver and therefore is the same as the frequency to 
which the receiver is tuned. The marking unit is 
coupled mechanically to the receiver by means of a 
sprocket and chain which drives the band-changing 
switch and by a phosphor bronze cable and pulley 
which drive the tuning condenser. In this way, fairly 
close tracking is obtained. With the marking frequen¬ 
cy in operation, a trace moves across the screen as the 
tuning dial of the receiver is turned. When this trace 
is superposed on a signal trace, that signal is heard. 
The frequency of the station may then be determined 
from the tuning dial. Evidently it would be possible to 
connect a transmitter to the frequency marker and 
thus jam the signals in question. 

Study of Scanning Problem 

During the development work, certain fundamental 
investigations were necessary. For example, in a scan¬ 
ning receiver the build-up time of the selecting filter 
limits the rate at which the given frequency band can 
be scanned, or, if the scanning rate is fixed, limits the 
width of frequency band covered. Conversely,, if the 
frequency band and scanning rate are fixed, then the 
width of the selecting filter and the consequent degree 
of resolution are limited. This problem had been 
studied by various investigators prior to the inception 
of this project. (See Chapter 12.) To make it possible 
to study this relationship as applied to the moving- 
screen type of scanning receiver and to permit differ¬ 
ent degrees of resolution of the frequency band as 
desired, provision was incorporated in the experimen¬ 
tal model for (a) different widths of selecting filter, 
namely 1, 5, 10, and 25 kc, (b) different scanning 
rates, namely 10, 20, 30, and 60 cycles per second, 
and (c) different frequency ranges from 8 me to 
0.05 me. Diagrams employing sweep rates of 30 cycles 
or less have been found to be unsatisfactory due to 
the noticeable gaps between successive marks for dif¬ 
ferent scanning cycles. Accordingly the 60-cycle sweep 
is generally to be preferred. With this sweep the entire 
8 -mc band can be covered using a 10-kc filter. For 
narrower bands greater resolution can be obtained 
with a narrower filter. 


CONFIDENTIAL 




96 


PANORAMIC RECEIVER WITH MOVING-SCREEN INDICATOR 


Pulse Reception 

Preliminary study of pulse reception in the range 
of several hundred megacycles indicates that with 
pulses which are on for only a very small percentage 
of the time and with a total signaling period which 
may be very brief, the method of scanning the fre¬ 
quency range as employed in the present receiver may 
be undesirable because the signals might be missed. 
On the other hand, because the pulses are extremely 


same amplitude and then through a discriminator 
which makes the amplitude of the pulses proportional 
to their frequency. The output of the discriminator 
could then be applied to a moving-screen cathode-ray 
tube so as to produce a diagram in which there would 
appear for each pulse train a horizontal line whose 
vertical position would indicate the radio frequency 
and whose length would correspond to the duration 
of the pulse train. The general method is illustrated 
in Figure 8A. 



I_J 


A 



stanoaro 

C-R TUBE 


B 

Figure 8. Proposed method of receiving short pulses on panoramic receiver. 


short compared to the period between them, there is a 
small probability that pulses from more than one sta¬ 
tion will occur simultaneously. Hence it is believed 
possible to receive a broad band of frequencies and to 
sort out the pulses which are found. Such sorting out 
might be accomplished by passing the received signal 
hand through a limiter which reduces all pulses to the 


Instead of the above, the output of the discrimina¬ 
tor might be connected to the vertical deflecting plates 
of a cathode-ray tube of the usual type as indicated in 
Figure 8B. Thus, whenever pulses were present, the 
vertical position of the electron beam would corre¬ 
spond to the radio frequency of the transmitting sta¬ 
tion. Now if a low-frequency sawtooth wave were used 


(CONFIDENTIAL 




































































FURTHER WORK 


97 


as a horizontal sweep, the pulses would move across 
the screen at a rate sufficiently rapid to form continu¬ 
ous lines. With flat-topped pulses, these lines would 
stand out in comparison with the traces produced by 
the sides of the pulses. Moreover, by applying the con¬ 
stant-amplitude pulses in the output of the limiter to 
the modulating grid of the tube, the traces during the 
intervals corresponding to the sides of the pulses could 
be largely blanked out. Thus the device would detect 


the presence of pulses and indicate their radio fre¬ 
quency but not the duration of the pulse train. If the 
sawtooth frequency were varied until a frequency is 
reached which is a submultiple of the pulse frequency 
of a particular pulse train, those pulses would be 
stopped on the screen and the pulse rate could be de¬ 
termined. With either arrangement shown in Figure 
8 it might be desirable to introduce a marking fre¬ 
quency as a simple means of frequency determination. 


'confidential 






Chapter 14 

IMPROVED PANORAMIC RECEIVER 


Further studies on panoramic reception, development of an 
improved long-scale model for the 3- to 10-mc range, study 
of multiline indicators, methods of indicating marker fre¬ 
quency. 

141 INTRODUCTION 

xder Project C-27 a panoramic receiver was con¬ 
structed which scanned the approximate frequen¬ 
cy range of 3 to 10 me or any of various narrower 
bands located anywhere in that range. The indicator 
employed was a special cathode-ray tube rotated to 
give a continuous motion of the screen in relation to 
the scanning beam. The sawtooth wave which con¬ 
trolled the f-m sweep deflected the beam in the catli- 
ode-ray tube in a vertical direction representing fre¬ 
quency while rotation of the tube gave a diagram in 
which time was the horizontal dimension. The equip¬ 
ment was built for laboratory tests and, therefore, 
had many more variables that would be useful in a 
field receiver. 

Tests indicated that the length of the frequency 
scale, approximately 4 in., was a limitation, particii- 
larlv when scanning a wide band in which a large 
number of signals were present. It was also desirable 
to extend the frequency range from 10 to 30 me. 
Numerous other factors needed further investigation. 
Accordingly, under Project C-36 a a long-scale 3- to 
10-mc model 1 was constructed, studies were made on 
scanning filter design, multiline indicators were ex¬ 
amined, methods of indicating marker frequency were 
developed, and techniques were worked out for pano¬ 
ramic reception in the range from 0.1 to 30 me. 

142 LONG-SCALE 3- TO 10-MC RECEIVER 

Before actual work could proceed on the lower-fre¬ 
quency receiver built under this project, several factors 
had to be determined. They are outlined in the para¬ 
graphs which follow. 

14 - 21 Length of Scale 

In studying methods of providing a long-scale in¬ 
dicator, theoretical and experimental investigations 

“Project C-36, Contract No. OEMsr-357, Bell Telephone Lab¬ 
oratories, Inc.. Western Electric Co., Inc. 


were made on length of scale, type of diagram, type of 
indication, scanning rate, type of cathode-ray tube 
and other factors. As a result a diagram in the form 
of a spiral on a 7-in. tube was chosen. 

Assuming no limitation in the resolution capabili¬ 
ties of the equipment, the amount of information that 
can be portrayed on the screen increases directly with 
length of scale. However, it is readily possible with 
available technique to display an amount of informa¬ 
tion far exceeding the capabilities of the observer. 
Thus for example, if radio stations were located at 5- 
kc intervals throughout a band of 8 me and the scale 
length associated with each station were held to 0.2 
in., the scale length would be greater than 300 in. 
Even though many stations were missing it would be 
quite impossible for an observer to monitor such a 300- 
in. scale. The length of scale was therefore determined 
by the capabilities of the observer rather than those 
of the scanning receiver. 

Observer capability will vary widely with observers 
and with the purposes for which observations are 
made. However, television experience indicates that 
screens 5 in. square are about right for arm’s length 
viewing of entertainment. This seems to indicate that 
the screen of a 7-in. cathode-ray tube is at least ade¬ 
quate for presenting all the panoramic receiver indica¬ 
tions that an observer can consistently use. The length 
of trace may be varied at the discretion of the observer 
and experience indicates that 40 in. is a good value 
for most conditions. 

14 - 2 - 2 Shape of Scale 

Ease of viewing requires that the scale be com¬ 
pressed into an area not much wider than its height. 
Accordingly, the scale should be doubled up in some 
manner. 

A retrace discontinuity during the scanning inter¬ 
val is avoided by bending the trace into a spiral. This 
also affords a substantially uniform distance between 
adjacent lines. The spiral may be adjusted to any de¬ 
sired length of trace and to occupy any desired per 
cent of the screen area by changing the number of 
turns of the spiral and adjusting the distance between 
turns. For normal operation the spiral in the 3- to 



CONFI 



98 




LONG-SCALE 3- TO 10-MC RECEIVER 


99 


10 -mc model is operated with 4 turns spaced about 
V‘i in. apart. The frequency band scanned during each 
turn is approximately the same. The progression of 
the scale is from high frequencies nearest the center 
to low frequencies at the outer end. Radio stations 
tend to have less frequency separation at the lower- 
frequency end of the band, so the outer or longer turns 
of the spiral are used for the low frequencies. 

14.2.3 Method of Obtaining Spiral Trace 

The electron beam of the cathode-ray tube may be 
caused to follow an elliptical path by applying alter¬ 
nating currents in quadrature phase relationship to 
the vertical and horizontal deflection coils. If the 
magnetic fields set up by the quadrature currents have 
the same value and are single-frequency sinusoids, the 
trace becomes a circle whose radius is determined by 
the peak magnitude of the magnetic field. If the mag¬ 
nitude of the alternating current in each set of coils 
is increased simultaneously at a rate linear with time, 
the trace will become spiral. 

In the 3- to 10-mc receiver the output of a variable 
frequency oscillator creates the quadrature currents. 
The currents of each phase are then modulated in 
magnitude by the same sawtooth pulse that controls 
the reactance tube of the sweep oscillator circuit in 
the scanning receiver. Accordingly the spiral is ini¬ 
tiated at the beginning of each sweep of the scanning 
receiver and each discrete frequency in the band 
scanned is represented by a particular position on the 
spiral. The sweep circuit is synchronized with a vari¬ 
able-frequency oscillator rather than with the power- 
supply frequency as in the case when the moving- 
screen indicator is used. 

The number of turns in the spiral is determined by 
the ratio between the frequency of the variable-fre¬ 
quency oscillator (and hence of the rotating field) 
and that of the sweep circuit. The usual settings are 
60 cycles per second for the variable frequency oscil¬ 
lator and 15 cycles per second for the sweep circuit. 
This gives a 4-turn spiral whose size is determined by 
the magnitude of the output of the variable frequency 
oscillator. The distance between turns is determined 
by the magnitude of the sweep power used to modu¬ 
late the quadrature currents. 


and produces a minimum amount of confusion be¬ 
tween adjacent turns of the spiral. Experience with 
the model indicates that there is a sacrifice in resolu¬ 
tion due to this method of indication. How much of 
this sacrifice might be eliminated by a different design 
of the scanning filters and of the compression arrange¬ 
ment has not been determined. Also it is not known 
which is the more important, improved resolution or 
a more pleasing type of indication. 

By swinging the electron beam at right angles to 
its sweep path when the signal is received, more detail 
and hence better resolution are obtained. This also 
permits more distortion in the scanning filters. How¬ 
ever, the indicator pattern resulting from modulated 
signals, static, etc., dances violently. It may be that 
the resultant confusion and observer fatigue would 
outweigh the improved resolution. While sidewise 
deflection can be used with the spiral trace, a parallel - 
line diagram is better adapted for this purpose. 

1425 Scanning Filters 

The resolution obtainable with different scanning 
filters at various scanning rates and band widths has 
been studied experimentally and compared with the 
optimum theoretically obtainable. An important cause 
of degradation is delay distortion in the filter, i.e., 
different transmission times for different frequencies. 
The signal response is prolonged due to the delay dis¬ 
tortion and a weak signal located near a strong one 
may be completely masked. Another difficulty is that 
when a compression amplifier follows the filter, the 
effect of sloping cutoff of the filter is to widen the pass 
band as the input signal strength becomes greater. 
This tends to mass signals at adjacent frequencies. 
Unfortunately the tendency in normal filter design 
is for the delay distortion to increase as the sides of 
the attenuation characteristics are made steeper. Ac¬ 
cordingly, obtaining optimum characteristics for the 
scanning filters is a very complicated matter. 

Perhaps one of the most important results of this 
research project was that concerned with improvement 
of resolution by reduction of distortion in the scan¬ 
ning filter. Unless special attention is given to this 
factor in the panoramic receiver design, the resolution 
may suffer materially. 


14.2.4 Methods of Indicating Signals 

Intensity modulation, which is used in the model, 
results in a diagram which is pleasing to the observer 


14.2.6 


Type of Cathode-Ray Tube 

Tests of long- and short-persistence screens indi¬ 
cated that long persistence has a definite advantage for 


('OXElHENTfAL 






100 


IMPROVED PANORAMIC RECEIVER 


this type of work. Accordingly an 1813-P7 tube hav¬ 
ing a cascade screen was selected. The fluorescent color 
was blue-white and the phosphorescent color was or¬ 
ange. An orange filter was provided so that the fluo¬ 
rescence could be eliminated if desired. 

Blanking Out Recognized Stations 

A simple blanking out was obtained by painting 
over the spots produced by identified signals. A slow- 
drying black paint was used so that the screen could 
be wiped clean at frequent intervals. (See Section 
12 . 8 .) 

14 - 2 - 7 Rate of Scanning 

The model is arranged for scanning rates of 4, 15, 
30, and 60 per second. With the long-persistence 
screen and orange mask, flicker is not objectionable 
at the higher rates and not serious even at the 4-per- 
second rate. The greater resolution obtained with the 
lower rate is distinctly worthwhile if the desired sta¬ 
tions are closely grouped in frequency or differ great¬ 
ly in magnitude. However, the 4-per-second rate has a 
very definite effect on the speed with which the marker 
can be adjusted to an unknown signal in order that 
the monitoring receiver may be tuned to it for iden¬ 
tification purposes. With the slow scan, rapid move¬ 
ment of the tuning control of the monitoring receiver 
causes the marker spot to move in jerks and become 
confused with spots associated with modulated signals 
such as telegraph signals. This is because the fre¬ 
quency rate of change of the marker becomes compar¬ 
able to that of the scanning oscillator. 

u.2.8 Reactions on Scanning-Receiver 
Design 

The use of the long-scale indicator imposes more 
severe requirements on the scanning-receiver design 
than does the use of the moving-screen indicator. 

1. The increased scale affords greater resolution and 
hence imperfections such as small nonlinearity in the 
sawtooth scanning wave become more noticeable. 

2. The sweep oscillator is no longer synchronized 
with the 60-cycle power supply. It is now synchronized 
with the variable frequency oscillator used to create 
the spiral trace. This means that the sweep-control 
circuits must be carefully protected against interfer¬ 
ence from the 60-cycle power circuits. Otherwise the 
phasing in and out of the interference and the vari¬ 
able-frequency oscillator output will cause erratic 


sweep times and distortion of the pattern. At times, 
60-cycle commercial power was used in place of the 
output of the variable-frequency oscillator. This is 
satisfactory if the commercial power is stable in volt¬ 
age and frequency and has exceptionally pure wave 
form. Otherwise, as is the usual case, the indicator 
pattern is distorted and the same signal does not ap¬ 
pear in the exact same spot during successive scans. 

The scanning control circuit wiring of the original 
model (Project C-27) was changed to reduce coupling 
with the commercial power circuits. Special care in 
grounding arrangements is necessary in each operat¬ 
ing location in order further to reduce 60-cycle 
coupling. 

D3 MODEL EQUIPMENT WITH 
SPIRAL-TRACE INDICATOR 

The 3- to 10-mc spiral-trace indicator was used 
with the same h-f equipment constructed for the mov¬ 
ing-screen indicator '(Project C-27). Components 
shown within the dashed lines of Figure 1 comprise 
parts of the original equipment. 

Referring to Figure 1, the operation of the h-f 
equipment used with the spiral-trace indicator is as 
follows. Incoming signals in the 3- to 10-mc range, 
after selection and amplification, are heterodyned to 
an i-f frequency of 40 me by a beat oscillator whose 
frequency is controlled by a sawtooth wave through 
a reactance tube. The range swept over by the beating 
oscillator can be adjusted either to 7 me or to any of 
a number of narrower bands which may be positioned 
anywhere in the 3- to 10-mc range. The signals from 
the 40-mc i-f stage are heterodyned with a fixed car¬ 
rier to bring them to a second intermediate frequency 
of 548 kc. At this point a relatively narrow band filter 
is used to separate out the different incoming signals 
in succession. Several different widths of band filter 
are available for use with different rates of sweeping 
over the frequency band and different widths of band 
swept over. The signals from the 548-kc filter are 
passed through a high-gain amplifier which compresses 
a wide range of input amplitudes to a very narrow out¬ 
put range. The signals then pass to the long-scale 
indicating equipment. 

14,31 Operation of Indicating Equipment 

The operation of the long-scale indicating equip¬ 
ment is as follows. Signals from the compression am- 


CONFIDENTIAL 




MODEL EQUIPMENT WITH SPIRAL-TRACE INDICATOR 


101 



Figure 1. Block diagram of long-scale 3- to 10-mc spiral-trace receiver. 


plifier are passed through a modulating amplifier to 
obtain a voltage sufficient to modulate completely the 
electron beam of the cathode-ray tube. The spiral trace 
is obtained by deriving two quadrature components of 
a sine wave and modulating these with a sawtooth 
wave whose frequency is a submultiple of the sine 
wave. The two resultant waves are applied to the ver¬ 
tical and horizontal deflecting coils, producing a 
spiral for which the number of turns is equal to the 
ratio of the sine wave frequency to the sawtooth fre¬ 
quency. 

Each received signal is indicated by a change in the 
intensity of the beam, giving a bright spot on the tube. 
Since the same sawtooth frequency is employed both 
for controlling the frequency of the f-m oscillator and 
for producing the spiral, the frequency of the signal 
is indicated approximately by the position of its spot 


on the spiral. The highest frequency signals appear 
near the center and the lowest frequency signals near 
the outer end. 

The spots for a given frequency signal are superim¬ 
posed during successive scannings. This permits in¬ 
creased signal resolution as compared to a moving- 
screen indicator because the scanning rate may be 
lower and the screen distance between signals may be 
larger. The liigh-persistence screen reduces flicker in¬ 
herent to low scanning rates. 

The model is arranged so that different numbers of 
turns in the spiral and different sizes of spiral can be 
obtained. This affords a wide range of scale lengths. 
Experience has indicated a preference for a 4-turn 
spiral which affords a total scale length of about 40 in. 
A longer scale or larger number of turns appears to 
confuse the observer. 


CONFIDENTI 


AL 











































































102 


IMPROVED PANORAMIC RECEIVER 


IN OUT 



Figure 2. Circuit of amplifier for modulating beam of 
cathode-ray tube. 

Incorporated in the model receiver was a Hammar- 
lund receiver with which signals of any frequency 
within the band being scanned could be aurally mon¬ 
itored. In conjunction with this receiver there was 


provided an arrangement for producing on the screen 
a marking trace of the same frequency as that to which 
the Hammarlund was tuned as a means of determin¬ 
ing the radio frequency of the signal being observed. 

14 3 2 Modulating Amplifier 

The circuit of the modulating amplifier is shown in 
Figure 2. This amplifier increases the output of the 
compression amplifier to a value sufficient to modu¬ 
late the electron beam of the cathode-ray tube over the 
useful range of intensity. 

The modulating amplifier receives from the com¬ 
pression amplifier approximately 0.75 volt peak. The 
grid bias is normally adjusted by the intensity con¬ 
trol to a value near —50 volts, which is required for 
substantially complete cutoff of the beam current. The 
modulating amplifier varies the grid voltage below and 
above the cutoff value. To provide ample margin for 
discrimination against noise, the amplifier is designed 
to produce approximately 1-10 volts peak. The second 
stage has a broadly tuned resonant output circuit 
which is essentially flat over the widest band received 
from the 548-kc filter with sharp reduction in gain 



Figure 3. Circuit of balanced modulator and deflection amplifier, 3- to 10-mc receiver. 


COXFIJ)FXT.TAl 


















































































































MODEL EQUIPMENT WITH SPIRAL-TRACE INDICATOR 


103 


HORIZONTAL MODULATOR 



60 Oj INPUT 


15 OJ SAWTOOTH INPUT 



OUTPUT 




15 OJ SAWTOOTH INPUT 




RESULTING SPIRAL ON CATHODE 
RAY TUBE 


OUTPUT 


Figure 4. Production of spiral trace for 3- to 10-mc receiver. 


outside the pass band. The gain control has approxi¬ 
mately a 6-db range. 

14 - 3 - 3 Balanced Modulators 

The circuit of the balanced modulator is shown in 
Figure 3. This circuit consists of two push-pull stages 
with resistance coupling between stages. A 60-cycle 
sine-wave voltage is fed to the two grids of the first 
stage in phase opposition. A 15-cycle sawtooth voltage 
is fed in phase to the same grids. By means of sepa¬ 
rate bias controls on each of the input tubes, the 
modulator may be closely balanced so that the saw¬ 
tooth wave is balanced out at the plates of the output 
stage where one of the deflection coils is directly 
coupled. The output current is then a 60-cycle sine 
wave, amplitude modulated by the sawtooth wave. 
A control on the sawtooth input voltage allows ad¬ 
justment for the desired percentage of modulation. 
The plate load consists of inductance and resistance 
whose time constant is made equal to that of the de¬ 
flection coil. This reduces distortion of the output 
wave. A potentiometer is provided in the plate load 
to provide for a small amount of unbalance of the d-c 
plate voltages, thus allowing direct current to flow 


through the deflection coil. This adjustment is used 
for centering the trace on the screen. 

One of these balanced modulators was connected 
to the horizontal deflection coil and another to the 
vertical deflection coil. The only difference in the 
operation of the two circuits is that the 60-cycle in¬ 
puts are in quadrature. This produces two output 
waves as shown in Figure 4. With these two waves 
applied to the two deflection coils, the resultant trace 
will be the four-looped spiral shown. 

The modulators are fed with the same sawtooth 
wave that controls the frequency of the f-m oscillator. 
The output of the modulators can swing up to about 
100 ma, which is required for full-scale deflection on 
the cathode-ray tube. 

1434 Phase Shifter 

The circuit of the phase shifter is shown in Figure 
5. This is a balanced circuit which takes a single 60- 
cycle sine-wave input and converts it to two outputs 
that are in quadrature. A variable attenuator is pro¬ 
vided in the nonshifted output to insert loss in this 
branch equal to the loss of the phase-shifting network, 
so that the levels applied to the vertical and horizontal 
modulators are equal. 


CONFIDENTIAL 
























104 


IMPROVED PANORAMIC RECEIVER 



Figure 5. Phase shift circuit for producing two quadra¬ 
ture 60-cycle outputs. 


14.3.5 Synchronization Amplifier 

The synchronization amplifier is a single-tube am¬ 
plifier with gain control for adjusting the amplitude 
of the 60-cycle synchronizing voltage (obtained from 
a Hewlett-Packard 200D oscillator) that is fed back 
to the sawtooth generator. 

144 GENERAL PLAN FOR 0.1- TO 30-MC 
SCANNING RECEIVER 

Following the work described above, plans were 
completed for a proposed 0.1- to 30-mc scanning re¬ 
ceiver, a block diagram of which is shown in Figure 
6. b The objective was to uncover any fundamental 
limitation which might be encountered in building 
a scanning receiver in this frequency range. Accord¬ 
ingly the plan was to minimize chances for spurious 
signals resulting from modulation products without 
resorting to unduly high intermediate frequencies 
and to use existing designs wherever feasible. Designs 
already completed for the earlier scanning receiver 

b Reports on some of the research which took place during the 
development of the two receivers described herein have been 
included in Chapter 12, thus concentrating the present report 
on apparatus actually designed and built. 


were to be used for the f-m oscillator and all circuits 
following the second modulator. 

The range of the proposed receiver was extended 
down to 0.1 me when it was found that this could be 
done without any major change other than the addi¬ 
tion of one input filter and associated beat oscillator. 
This would provide a universal-type receiver which 
could be adjusted to monitor any frequency between 
0.1 and 30 me, thereby eliminating the necessity for a 
separate receiver covering the frequencies between 0.1 
and 10 me. 

Experience in laboratory and field demonstrations 
indicated that a band of 10 me was probably the 
largest that could be viewed advantageously at one 
time. Consequently, the total range covered by this re¬ 
ceiver was divided into three bands of about 10 me 
each, any one of which could be obtained by selecting 
the proper input filter and associated fixed frequency 
oscillator. Circuit design advantages resulting from 
this division of frequencies are discussed below. 

The proposed receiver was a triple heterodyne cir¬ 
cuit in which the incoming band of frequencies was 
converted by means of a fixed crystal oscillator and 
modulator to an intermediate band of 55 to 65 me, 
which in turn was converted in a second modulator to 
an f-m band having a mean frequency of 40 me. This 
conversion was accomplished by means of an f-m wave 
varying between 95 and 105 me obtained by tripling 
the output of an f-m oscillator varying between 31.66 
and 35 me. 

The output of the second modulator passed through 
a narrow-band 40-mc filter to a third modulator where 
it was combined with a frequency of 40.548 me. The 
resulting lower side band was then filtered, amplified 
and applied to the cathode-ray-tube indicator. 

Aural monitoring of radio stations was provided by 
means of a receiver responding to either a-m or f-m 
signals and operating from the 55- to 65-mc inter¬ 
mediate frequency, where minimum tuning effort is 
required to cover all input frequencies. A marking 
oscillator adjusted to track with the frequency to 
which the monitoring receiver was tuned was fed into 
the 55- to 65-mc amplifier circuit as an aid in tuning 
the monitoring receiver to any station appearing on 
the cathode-ray tube. 

1441 Components for the Receiver 

The main components of the proposed receiver are 
discussed in this section. The progress which had been 


CONFIDENTIAL 



































GENERAL PLAN FOR 0.1- TO 30-MC SCANNING RECEIVER 


105 



L 


Figure 6. Block diagram of the 0.1- to 30-mc scanning receiver. 


made on each when it was decided that further work 
was unnecessary is indicated. Detailed information 0 

c This information includes insertion-loss characteristics of the 
input filters, circuit and gain frequency characteristics of the 
r-f amplifier and [first modulator, circuit of the 35-, 45-, and 
55-mc oscillator, insertion loss of the 55- to 65-mc band-pass 
filter, details of interstage transformers and circuit for the 
55- to 65-mc band-pass amplifier and second modulator, circuit 
of the f-m oscillator, circuit of the f-m tripler amplifier, and 
insertion loss of the 40-mc band-pass filters. 


pertaining to these components not included in this 
summary, will be found in the contractor’s final 
report. 1 

Input Filters 

Input filters connected between the antenna and the 
r-f stages limit the range of frequencies connected to 
the vacuum tubes at any one time to a band width of 


CONFIDENTIAL 


































































106 


IMPROVED PANORAMIC RECEIVER 


approximately 10 me. This not only eliminates image 
frequency responses but also simplifies the circuit con¬ 
ditions necessary for restricting the modulation prod¬ 
ucts which might produce false images on the screen. 
Limiting the band scanned to 10 me further simplifies 
circuit design and operation by reducing the frequency 
band swept by the f-m oscillator. 

The three input filters were substantially completed. 

R-F Amplifier and First Modulator 

Experience gained with the 3- to 10-mc scanning 
receiver indicated a need for greater sensitivity, so an 
r-f amplifier was provided with a gain of 23.5 ± 1 db 
over the range of 0.1 to 30 me. This is only about 7 db 
more than that in the present scanning receiver, but 
additional gain was provided in the 55- to 65-mc i-f 
amplifier so that an overall additional gain of about 
25 db was obtained. 

The first modulator was arranged for cathode injec¬ 
tion of the beat oscillator signal in order to lessen the 
shunting effect caused by mixing the signals in a pen¬ 
tode. Although the modulator was built with the r-f 
amplifier, it was tested. The construction is such that 
conversion to grid injection may be made if necessary 
with only minor circuit modifications. The output of 
the modulator goes directly into the 55- to 65-mc 
filter which constitutes a part of the plate circuit of 
the modulator. 

35-, 45-, and 55-mc Oscillator 

It was planned to have the aural monitoring receiver 
operate from the first i-f frequency, so it was necessary 
that the beat oscillator frequency remain constant in 
order that the monitoring receiver might be calibrated 
in terms of input frequency to the scanning receiver. 
Consequently crystal-controlled oscillators were chosen 
to supply the beat frequencies for the first modulator. 
Crystals for frequencies below 10 me were more readily 
obtainable than those for higher frequencies, so crys¬ 
tals of 3.89, 5, and 6.11 me were chosen. The oscillator 
output was to be tripled twice, in cascade, to obtain the 
desired modulating frequencies. 

A single oscillator and tripler circuit was contem¬ 
plated. The frequency desired was to be obtained by 
switch selection of the proper crystal and associated 
tuning condensers in the filter circuits. Sharply tuned 
filter circuits are necessary in order to obtain large 
discrimination in favor of the desired frequency. This 
oscillator was not built. 


55- to 65-mc Band-Pass Filter 

This is a coaxial-line type filter having low loss in 
the pass band and sharp cutoffs. It had an insertion 
loss of about 2.5 db over the pass band and was down 
10 db at 55 and 67 me and 30 db at 52 and 70.5 me. 

55- to 65-mc Band-Pass Amplifier and 
Second Modulator 

The input circuit of this amplifier constitutes a part 
of the preceding filter, so that the filter may operate 
most effectively. High gain per stage and discrimina¬ 
tion in favor of the 55- to 65-mc band, in addition to 
that obtained by the band-pass filter, was obtained by 
the use of interstage transformers designed to match 
the impedances of the associated output and input 
circuits. 

In the second modulator the 10-mc band of frequen¬ 
cies is frequency modulated with 95 to 105 me. The 
output of this modulator passes through a 40 ± 0.1- 
mc filter which is sufficiently narrow to suppress image 
frequencies in the third modulator. Effectively, the 
f-m modulating voltage sweeps the input band of fre¬ 
quencies past a filter having a 0.2-mc width. Signals 
in this band will pass through the filter when the 
modulation products fall within the limits of the filter. 

F-M Oscillator 

The f-m oscillator was designed to vary from 31.6 
to 35 me. Its output was tripled to produce the desired 
modulating frequencies of 95 to 105 me. This arrange¬ 
ment was chosen in preference to designing an oscil¬ 
lator to work directly at 95 to 105 me, because it per¬ 
mitted use of the design employed in the earlier 3- to 
10-mc scanning receiver. Stability and linearity of 
operation were enhanced by limiting the f-m sweep to 
about one-half the maximum obtainable with the cir¬ 
cuit used. It was intended that this oscillator should be 
provided with means for controlling the band width 
swept and for centering the sweep about any desired 
frequency in the band, similar to those in use in the 
3- to 10-mc receiver. 

The f-m oscillator was similar to but smaller and 
more compact than the one used in the 3- to 10-mc 
receiver. Preliminary tests showed that its perform¬ 
ance was at least as good as that of the earlier model. 

F-M Tripler and Amplifier 

This is a conventional multiplier-amplifier circuit 
having the multiplier tube biased beyond cutoff and 


| (OXFIDEXTIAL 





GENERAL PLAN FOR 0.1- TO 30-MC SCANNING RECEIVER 


107 


the multiplier interstage transformers tuned to the 
third harmonic of the input frequency, which in this 
case is a band from 95 to 105 me. One amplifier stage 
precedes the multiplier stage to insure adequate input 
level. The f-m tripler and amplifier was built and 
tested. 

Monitoring Receiver and Marker 

A receiver providing adequate coverage of the range 
from 0.1 to 30 me would need to be divided into five 
or six separate tuning bands which would necessitate 
frequent band shifts when operating the scanning re¬ 
ceiver. To avoid this frequent shifting, the monitoring 
receiver was to be operated from the 55- to 65-mc in¬ 
termediate frequency. The receiver proposed for this 
use was a special Hallicrafter having an expanded 
range of 55 to 65 me and three separate frequency 
scales of 10 me each and calibrated in terms of input 
frequency to the scanning receiver. 

A marker oscillator geared to coincide with the fre¬ 
quency to which the receiver is tuned was planned to 
furnish a signal for injection into the i-f (55- to 65- 
mc) amplifier as a means for quickly selecting and 
identifying any signal appearing on the cathode-ray 
tube indicator. 

Miscellaneous 

The circuits for the remaining components were to 
be essentially duplicates of those in the 3- to 10-mc 
receiver except that the apparatus was to be rearranged 
to meet the requirements of the new equipment layout. 
Considerable attention was given to providing proper 
layout, shielding, and filtering between circuits to 
eliminate spurious signals such as have been observed 
in the present model of the scanning receiver. 

The 40-mc band-pass filter was built and tested. 
Two scanning filters were built and tested as part of 
the study of desirable scanning filter characteristics 
and for use in the 0.1- to 30-mc receiver. The W-76099 
filter consisted of two crystal sections separated by a 
resistance pad. The W-76132 filter consisted of four 


crystal sections separated by vacuum tubes or resist¬ 
ance pads. 

144 2 Further Work 

Many problems in connection with panoramic re¬ 
ception remained to be solved but these were beyond 
the scope of Projects C-27 and C-36. The importance 
of many of these problems is contingent on the pur¬ 
poses for which the receiver is desired and the operat¬ 
ing conditions. 

Some of these problems are listed below. No attempt 
has been made to evaluate their relative importance. 

1 . Radio-frequency tuning ahead of the first modu¬ 
lator varied in synchronism with the sweep wave to 
reduce intermodulation products and make scanning- 
receiver performance more nearly comparable to that 
of fixed-tuned receivers. 

2 . Division of the band swept into fractions for re¬ 
ception, amplification, and modulation and then re¬ 
combination to permit rapid scanning of bands of par¬ 
ticular interest without requiring continuous scan¬ 
ning of the entire band. 

3. Scanning receivers to indicate primarily relative 
amplitudes rather than the frequencies of the signals 
scanned. It seems likely that radically different scan¬ 
ning-filter and indicator characteristics may be de¬ 
sirable. 

4. Mechanical versus cathode-ray tube indicators. 

5. Facsimile and magnetic-tape recorders. 

6 . Blanking devices for reducing operator distrac¬ 
tion due to the presence of identified signals. 

7. Electronically controlled f-m oscillators for use 
at higher frequencies and to afford larger percentage 
swing. 

8 . Investigation of detector characteristics and their 
effect on receiver resolution, particularly resolution be¬ 
tween signals of widely different amplitude. 

9. Simplified methods of combining wide- and nar¬ 
row-band scanning. (Present technique involves sepa¬ 
rate equipment and indicator.) 

10 . Electronic switching of broad-band circuits 
without the present large signal-to-noise ratio penalty. 


| CONFIDENTIAL 





Chapter 15 


RECEIVER FOR PULSE SIGNALS 


Research leading to the design and construction of a scan¬ 
ning receiver covering the range of 350 to 750 me, indicating 
when radio position-finding pulses are directed at the receiver, 
arranged to scan automatically, to stop and sound an alarm 
on reception of a signal, and to indicate azimuth of received 
signal. Information from this project contributed to better 
understanding of interception problems and was used in the 
development of the AN/ARQ-9 radio set. 

15-1 INTRODUCTION 

U nder projects C-27 and C-36 the principles of 
panoramic reception had been explored quite 
thoroughly but nothing had been developed for the 
band width and frequency range contemplated under 
Project C-39. a After a study of the principles involved 
and the problems, a model receiver was constructed 
capable of automatically scanning the band of 350 to 
750 me, stopping on reception of a signal, or alterna¬ 
tively continuously scanning the band and sounding 
an alarm on reception of a signal. Features were in¬ 
cluded to determine the repetition frequency and pulse 
length of the incoming signals and arrangements were 
worked out whereby the azimuth from which the sig¬ 
nals came could be determined. 

152 GENERAL PRINCIPLES 

In studying the problem of reception of radar pulses 
with a scanning receiver, it was necessary to consider 
the types of information desired and the differences 
between the scanning reception of radar pulses and 
that of ordinary signals. 

Information which might be portrayed would in¬ 
clude existence of pulses, frequency, azimuth, range, 
pulse rate, pulse shape, and length of pulsing period. 
These types of information differ both in difficulty of 
determination and usefulness after being determined. 
The first four are of greater use tactically; the latter 
three are of interest more from the research point of 
view. 

The existence of pulses and their carrier frequencies 
are probably of primary interest, since their very pres¬ 
ence indicates the presence of radar equipment. De¬ 
termination of frequency indicates whether or not the 

“Project C-39, Contract No. OEMsr-311, Bell Telephone 
Laboratories, Inc., Western Electric Co., Inc. 


source is friendly. Azimuth indication would be of 
considerable tactical importance on a ship or aircraft, 
since it would enable the craft to approach the source 
of signals. 

It was apparent after some study that it would be 
an enormous undertaking to construct a receiver which 
would furnish complete information concerning all 
variables for all types of transmission. Accordingly, it 
was decided to construct a receiver of fairly simple 
design which would indicate the presence and fre¬ 
quency of both pulse signals and other types of trans¬ 
mission. 

153 SCANNING RECEPTION OF 
PULSE SIGNALS 

There are a number of differences between scanning 
reception of radar pulses and that of ordinary signals. 
In the first place the frequency spectrum of a h-f pulse 
is quite broad. For practical purposes most of the en¬ 
ergy of the pulsed high frequency may be assumed to 
lie within a band centered at the carrier frequency and 
having a width equal to twice the reciprocal of the 
pulse length. This band for a 1-fisec pulse would be 
approximately 2 me. 

It is desirable that the scanning filter be at least as 
wide as the pulse signal band for two reasons. First, 
this affords a good signal-to-noise ratio and, second, 
it permits stopping the receiver and viewing the pulse 
shape. Since the resolution is limited in any case by 
the signal band width, no sacrifice of resolution results 
from making the filter band at least as wide as the sig¬ 
nal band. 

In a panoramic or scanning receiver there is a fun¬ 
damental relation between the width of the scanning 
filter and the scanning speed, i.e., the rate at which 
frequencies are swept past the filter. To obtain satisfac¬ 
tory transient response from the scanning filter, the 
band width should not be less than that indicated by 
the following formula: 

B = K B \/nF, 

where B = width of scanning filter in cycles per sec¬ 
ond measured between 6 db points, nF = scanning 
speed in cycles per second per second (i.e., the product 
of repetition rate n and total frequency band F), and 


CONFIDENTIAL 


108 





SCANNING RECEPTION OF PULSE SIGNALS 


109 


K B is a proportionality factor. For good design and 
small level difference between adjacent signals which 
are to be resolved, the value of A” g may be taken as 1.5. 

Although the above relationship constitutes a very 
important limitation in the reception of narrow-band 
signals it turns out in a scanning receiver for pulse 
signals that with practical repetition rates and fre¬ 
quency sweeps, if the width of the scanning filter is 
made equal to the pulse signal band, the scanning 
speed is no longer a limitation. Thus, for example, 
with a 2-mc filter the scanning speed may be of the 
order of 10 12 cycles per second per second. 

Duty Cycle 

A further factor in scanning reception of pulse sig¬ 
nals is the low duty cycle, i.e., the fact that the signals 
are present only a small part of the time. In the proc¬ 
ess of scanning reception a number of pulses are lost, 
the fraction received being equal to the ratio of the 
width of scanning filter to the total frequency band 
covered. If, in addition to frequency scanning, the re¬ 
ceiver employs azimuth scanning or if the source of 
the pulse signals employs azimuth scanning, the frac¬ 
tion of the original pulses received is much further 
reduced. 

Pulses must be present during a sufficient percent¬ 
age of the time to actuate the indicator associated 
with the scanning receiver. The minimum band width 
for the scanning filter to satisfy this condition may be 
stated as 

B = K t FT, 

where B is the band width, F is the total frequency 
band, T is the period of the pulsed signal or the in¬ 
verse of the pulsing rate, and K r is the number of 
pulses that must be received per second to actuate the 
indicator. If we assume the total band to be swept is 
400 me and that 10 pulses per second are necessary to 
actuate the indicator, then the scanning filter should 
be 10 me wide for a 400-cycle pulsing rate. Further¬ 
more it is apparent that scanning reception in com¬ 
bination with azimuth scanning either in the receiver 
or in the transmitter presents an extremely difficult 
problem. 

Signal-to-Noise Patio 

The signal-to-noise ratio of a scanning receiver for 
radar pulses is poorer than that of the same receiver 
for continuous narrow-band signals of the same total 
field strength. There are two reasons for this. First, 
the pulse receiver is responsive to noise all the time, 


while the pulses are present only a small part of the 
time. The loss in signal-to-noise ratio on this account 
corresponds directly to the duty cycle of the radar 
pulses. Second, the received noise power varies directly 
with band width so that the noise is correspondingly 
greater for wide-band pulse power than for a signal 
of the same power concentrated in a narrow band. 
Consequently the sensitivity of a receiver for pulses 
is poorer than that for a receiver for continuous nar¬ 
row-band signals in direct proportion to their relative 
band widths. 

In addition it was necessary to consider the band 
width required for stopping the tuning mechanism 
while the signal was still tuned in. These considera¬ 
tions led to the choice of a scanning-filter band width 
of approximately 10 me. The total band covered, 350 
to 750 me, was determined by the sweep range realiz¬ 
able in the beating oscillator. The scanning rate of 1 
sweep per second was chosen to provide a suitable in¬ 
terval between audible alarms for different signals ex¬ 
pected to be in the range and to allow time for the 
mechanism to operate. This sweep rate was also satis¬ 
factory for later addition to the receiver, if desired, 
of a visual indicator to show simultaneously all signals 
present in the frequency band. 

15.3.1 Alternative Arrangements Studied 

In addition to the design finally selected, other al¬ 
ternative arrangements were studied. The first scheme 
considered for indicating the existence and frequency 
of pulse signals portrayed the signals by a vertical de¬ 
flection of the cathode-ray beam, the height of the de¬ 
flection being proportional to the frequency of the re¬ 
ceived pulse. A horizontal time scale would be pro¬ 
vided by rotating the tube or by means of a sawtooth 
sweep circuit. The unfavorable signal-to-noise ratio 
caused by the fact that the whole band must be am¬ 
plified and not just that portion occupied by the single 
pulse being received forced the abandonment of the 
scheme. 

Another arrangement considered would indicate azi¬ 
muth of the signals as well as their existence and fre¬ 
quency. Analysis indicated that very few signals would 
be received because of the fact that both receiving and 
transmitting antennas would be rotating, that the re¬ 
ceiving antenna would have high directivity, that the 
receiver band width would be a small part of the whole 
frequency band swept through. This scheme was aban¬ 
doned. 


CONFIDENTIAL 





110 


RECEIVER FOR PULSE SIGNALS 


153 2 Azimuth Indicating System 

The model receiver actually constructed was de¬ 
signed so that it could be used as the basis of a four- 
channel azimuth scheme in which four antennas with 
circular directivity patterns directed to the four com¬ 
pass points fed voltages to the four deflecting plates 
of the cathode-ray tube. The radial deflection of the 
beam indicated the direction from which the signals 
came. 


350 - 750 MC SECOND I*F 



Figure 1. Block schematic of the 350- to 750-mc pano¬ 
ramic receiver for pulse signals. 


15 4 DETAILS OF MODEL RECEIVER 

A block schematic diagram of the receiver is shown 
in Figure 1. The heterodyne oscillator is controlled 
either by the motor and clutch arrangement shown or 
manually. In addition to the automatic volume con¬ 
trol, a manual gain control has been provided to ad¬ 
just the threshold of sensitivity of the receiver. Part 
of the signal from the video amplifier is connected to 
the alarm and control circuit which actuates a bell 
when a signal is received and is also capable of stop¬ 
ping the sweep oscillator when a signal is received. 


Part of the output signal is fed to an a-f monitoring 
amplifier. Part of the audio frequency from this am¬ 
plifier is connected to a linear sweep circuit to syn¬ 
chronize the sweep rate with the pulsing rate of an in¬ 
coming signal. 

Antenna 

The antenna provided with the receiver is mounted 
in the middle of the top of the cabinet and consists of 
a quarter-wave vertical stub resonant at approximately 
the middle of the band, 550 me. The antenna was made 
with a length-to-thickness ratio of about 6 to provide 
reasonable performance over the 350- to 750-mc band 
of the receiver. 

Converter 

The converter circuit is shown schematically in 
Figure 2. The converter itself is a Western Electric 


ANTENNA TRANSFORMER 



Figure 2. Silicon converter schematic. 

fixed silicon crystal. A transmission-line stub trans¬ 
former is connected between the antenna jack and the 
crystal to match the impedance of the antenna ap¬ 
proximately to that of the crystal. The transformer 
was designed for an impedance ratio of 72 to 275 
ohms. This transformer also serves as a broadly tuned 
input filter to attenuate the signals outside the 350- 
to 750-mc band. A 20-ju/xf capacitance is part of the 
mounting block of the crystal converter. This capaci- 


(’ONFIDENTIAll % 


















































































DETAILS OF MODEL RECEIVER 


111 



TO ALARM AND CONTROL 
CKT 


Figure 3. Oscillator using GL-44G tube tuning over range from 420 to 090 me. 


tor is effectively a by-pass for the signal frequencies. 
The 60-mc i-f output from the converter is obtained 
across this capacitor. Another stub transformer, this 
time at 60 me, is used to match the 275-ohm imped¬ 
ance of the crystal to the 72-ohm impedance of the 
first i-f amplifier. 


Oscillator 

A schematic of the oscillator is shown in Figure 3 
and a cross section in Figure 4. The GL-446 tube is 
mounted in one end of a cylindrical cavity which is 
effectively a short-circuited coaxial line about A/4 long 

























































































































































112 


RECEIVER FOR PULSE SIGNALS 


aoi>*f 



ALARM a MOTOR 
CONTROL CIRCUIT 


Figure 5. CRO control circuits and gas tube circuit for alarm and motor control. 


at the highest frequency of oscillation, 690 me. This 
coaxial line is connected between the plate terminal 
and cathode of the tube. Grid excitation for maintain¬ 
ing oscillations is obtained from two thin metallic 
fingers which project from the end of the cavity and 


make contact with the grid connection on the tube. 
These fingers effectively form pickup loops in the os¬ 
cillating field at the high-voltage end of the cavity. 

The oscillator is tuned over a frequency range from 
690 to 410 me by a variable air capacitor connected 


CONFIDENTIAL 


| 



















































































































DETAILS OF MODEL RECEIVER 


113 


across the cavity at its high impedance point, that is, 
as near to the tube as possible. The capacitor has two 
semicircular plates and in 180 degrees of rotation its 
capacitance varies about 5 fi/xi. This capacitor may he 
operated manually or driven by a motor. 

The plate voltage for the oscillator is supplied 
through the inside of the coaxial inner conductor. The 
plug which fills the end of this hollow inner conductor 
has a capacitance to the inner conductor of about 100 
/jl/jlI acting as a high-frequency by-pass from the plate 
to the cavity. Additional filtering is obtained by vir¬ 
tue of the inductance of the plate lead wire and an¬ 
other by-pass capacitor near the other end of the inner 
conductor. The output is obtained by means of a small 
pickup loop projecting into the cavity at the high- 
current or short-circuited end. 

Intermediate-Frequency Amplifiers 

The two i-f amplifiers are much the same in con¬ 
struction. The first amplifier has two stages, the sec¬ 
ond four, 713A tubes being used throughout. Both 
amplifiers have 72-ohm input and output impedances. 
The first amplifier has a gain of about 30 db and the 
second a maximum gain of 65 db. The gain of the sec¬ 
ond amplifier is controlled by means of the grid bias 
voltage on the second and third stages. This voltage 
is furnished by the a-v-c circuit or the manual gain 
control which is a potentiometer across a 9-volt bat¬ 
tery. The combined gain of the two amplifiers may be 
varied between 88 db and substantially 0 gain by a 
range of 9 volts grid bias. 

Detector 

The detector is of the so-called infinite impedance 
type using a 713A tube. The envelope voltage appears 
across a 5,000-ohm resistance in the cathode circuit 
of the tube. The input impedance to the circuit is 72 
ohms, similar to that of the input to the i-f amplifiers. 
A choke in the output lead is provided to block the 60- 
mc intermediate frequency. 

Video Amplifier 

The envelope of the received signal is applied to the 
input of the video amplifier which is a 6AC7 tube. The 
proportion of the signal which is connected is con¬ 
trolled by means of the video gain controls in the grid 
circuit of the 6AC7 tubes. The cathode bias resistor 
for the two 6AC7 tubes is not by-passed and provides 
phase inversion for the applied signal. The plate-cir¬ 
cuit stages for the two 6V6 output tubes are induct¬ 


ance-compensated resistance networks to extend the 
gain characteristic of the amplifier. 

The plate voltages and cathode biases of the tubes 
in the video amplifier are all connected to a rotary 
switch through suitable dividing resistances so that a 
voltage indication may be obtained on a 0- to 2-ma 
meter on the face of the panel. 

A-V-C Circuit 

Output signal pulses of positive polarity from the 
video amplifier are connected through the two halves 
of a 6H6 tube V13 and applied to the grid of the 
6V6 tube V15 as shown in Figure 5. The proportion 
of the voltage applied to the 6V6 tube is controlled 
by means of the a-v-c potentiometer. The amplified 
impulse is rectified by V16 and applied to the grids 
of the second and third stages of i-f amplifier No. 2. 
The amplifier gain is thus controlled by the amplitude 
of the received signal. The i-f gain potentiometer con¬ 
trols the fixed bias and hence the maximum gain of 
these stages. 


B 



Figure 6. Relay circuits employed in alarm and motor 
control system. 


Alarm and Motor Control 

A gas discharge tube and a relay sequence permit 
continuous scanning of the frequency range, sound- 


CONFIDENTIAL 
































































114 


RECEIVER FOR PULSE SIGNALS 


ing an alarm whenever a signal is swept through or, 
to permit interrupted scanning, sounding an alarm 
and automatically stopping the scanning mechanism 
on the receipt of a signal. 

Positive pulses from either side of the output of 
the video amplifier are connected through a 6116 and 
applied to the grid of the 885 gas discharge tube VI7 
as shown in Figure 5. The breakdown point of this 
tube is controlled by the “aim sens” potentiometer. 
Figure 6 shows the alarm and motor control circuits. 

With the scan key in the “cont” position for contin¬ 
uous operation, the discharging of the gas discharge 
tube by a signal causes the C relay to operate. This, 
in turn, operates the D relay, which completes the 
alarm circuit. The D relay also operates the B relay, 
restoring the gas discharge tube and releasing the 


relays by opening the plate battery lead. Thus, the 
alarm has sounded and the trigger circuit restored to 
normal for a subsequent signal response. 

With the scan key in the “int” position for inter¬ 
rupted operation, the receipt of a signal causes the 
gas tube to discharge, the C and D relays to operate, 
and the alarm to sound as described for continuous 
operation. In this case, however, the C relay also op¬ 
erates the .1 relay, which releases a clutch mechanism 
disengaging the motor from the frequency scanning 
capacitor of the oscillator. The total sequence of oper¬ 
ation is sufficiently rapid to permit the scanning os¬ 
cillator to remain tuned to the incoming signal. Tc 
release the circuit to resume the scanning for othei 
responses, the reset button is pressed, opening the 
plate circuit of the gas tube. 


CAVITY OSCILLATOR 



I* iGURE t. Oscillator and video chassis of the 350- to 750-mc pulse receiver. 

\ CONFIDENTIAL 












































FUTURE WORK 


Operating the “aim” key to the off position opens 
the grid lead of the gas tube disabling the alarm and 
motor and control circuit. When operating the receiver 
manually, the tune key is operated to the “man” posi¬ 
tion. This disengages the motor clutch mechanism and 
disables the alarm and relay circuit. Operating the 
manual frequency control also opens the circuit to the 
clutch mechanism. 

Monitor and Sweep 

A monitor and sweep circuit are provided to assist 
in the identification of signals and the determination 
of the type of transmission. Connection to this circuit 
is made from the cathode of the video tube Y3. The 
signal is amplified by a 6AC7 tube and then is applied 
to the grid of a gas tube sweep circuit, the output of 
which is applied to the horizontal deflecting plates. 

Test Oscillator 

Pulse-modulated signals are supplied from a Wien 
or PC oscillator for testing the receiver. Sine-wave 
frequencies of 400, 1,000, 1,600, 2,000, and 4,000 
cycles from the oscillator are selected by a switch and 
are amplitude-controlled by a lamp in the cathode 
circuit of the oscillator. The bridge output signals are 
peak-chopped and are then applied to parallel reso¬ 
nant circuits tuned to 1, 0.5, 0.33, and 0.2 me respec¬ 
tively. A diode damps out all but the first positive half 
cycle of the transient so that the width of the result¬ 
ing pulses is 0.5, 1.0. 1.5, or 2.5 /xsec. These pulses are 
amplified and shaped and then modulate the plate 
voltage of a cavity oscillator, permitting it to oscillate 
only during the pulse interval. The frequency range of 
this oscillator, which is similar to the oscillator in the 
receiver, is 550 to 730 me. 

The oscillator build-up time is just less than 0.5 
/xsec so that the pulses resulting are that much shorter 
than the impressed modulating pulse. Operating the 
pulse-cont key to the “cont” position removes the modu¬ 
lating signal, permitting the cavity oscillator to sup¬ 
ply a continuous output. In this condition the oscil¬ 
lator will cover the range of 450 to 730 me. A quarter- 
wave stub antenna, similar to that used with the re¬ 
ceiver, may be connected for transmitting. 

Receiver Characteristics 

The sensitivity of the receiver to pulse-modulated 
signals is such that about 50 pv are required for a 
y 2 -m. deflection. The band width of the i-f amplifier 


115 



Figure 8. Converter assembly for panoramic receiver 
for pulse signals. 

is 9.5 me at the —3 db point, frequencies 10 me away 
from the 60-mc mid-point being 70 db down. At a puls¬ 
ing rate of 4,000 cycles and a pulse width of 2 ^sec, 
a 56-db variation in input signal produces a 16-db 
variation in cathode-ray tube deflection. 

15 5 FUTURE WORK 

There are a number of possibilities for further work 
along lines related to this project. 1 Some of these rep¬ 
resent improvements which might be made on the 
equipment as it now stands and others represent ex¬ 
tension of the work in different directions. 

A panoramic indicator might be attached to the 


CONFIDENTIAL 


















116 


RECEIVER FOR PULSE SIGNALS 


present receiver to give an instantaneous cross section 
of the frequency band covered. This would be in the 
usual form of a vertical deflection for each received 
signal on a horizontal base line to which a frequency 
scale could be applied. The circuits necessary to do 
this have been designed. The scan would involve a 
360-degree potentiometer which would be driven by 
the oscillator motor drive mechanism. From this po¬ 
tentiometer a horizontal pyramidal sweep wave would 
be generated. This would be applied to the horizontal 
deflecting system of the cathode-ray tube instead of 
the sawtooth sweep which is used for examining sig¬ 
nals. The received signals would be applied to the ver¬ 
tical deflecting plates as at present. Because of the 


image response, the frequency scale for the abscissa 
would be exactly the same as the frequency scales on 
the upper half of the frequency dial of the receiver. 
That is, there would be a frequency scale for signals 
higher in frequency than the beating oscillator and 
another scale for signals lower in frequency than the 
beating oscillator. These two scales differ from each 
other by 120 me or twice the intermediate frequency. 
This double frequency response makes the panoramic 
indication of doubtful value in the model receiver. 

The equipment layout and arrangements of this re¬ 
ceiver were designed with a view towards the possibil¬ 
ity of constructing a 4-channel receiver for azimuth 
indication. 


f CONFIDENTIAL 





PART V 


INTERFERENCE GENERATION 


T he extreme importance of proper communication 
between units and between commanders and their 
units in any form of warfare, is paralleled by the 
equivalent importance of detecting enemy communi¬ 
cations and, if possible, of making them ineffective. 
Thus, observers of World War II before our entry into 
it were struck by the effectiveness of the liaison be¬ 
tween tanks and planes by which German tanks were 
piloted by German planes flying over the battlefields 
and were warned of hazards and obstacles in their 
path. 

If this team of plane and tank could be broken up 
by making it difficult or impossible for one to com¬ 
municate with the other, the value of the team could 
be decreased considerably. 

Several studies of this problem were made in Divi¬ 


sion 13 and resulted in at least two jammers in addi¬ 
tion to a fundamental examination of the most effec¬ 
tive means of disrupting communication by speech or 
code on amplitude or frequency modulation by fac¬ 
simile or pulse transmission. 

Three Division 13 projects were concerned directly 
with the interference production problem, C-25, C-26, 
and C-56. The first two were devoted essentially to 
production of interference generators covering the 
bands from 2 to 20 and from 15 to 30 me respectively, 
although C-26 also covers a great deal of basic re¬ 
search, while C-56 dealt entirely with the fundamental 
aspects of the means for effective jamming. In the 
summaries that follow the work is presented in a logi¬ 
cal sequence of interest rather than in the chrono¬ 
logical order in which it was done. 


confident: 


117 



V 



















Chapter 16 

STUDY OF INTERFERENCE GENERATION 


Research to determine the most effective types of inter¬ 
ference signals for jamming radio-telephone and telegraph 
communication channels. This summary contains all of the 
technical data available in the contractor’s final report. 1 
Later, more extensive work was done by Division 15. 

161 INTRODUCTION 

This summary presents in condensed form the data 
and conclusions from the work on this project,* in¬ 
cluding those from some preliminary experiments 
which preceded the actual authorization of the project. 
The experimental work covered rather completely the 
question of the effectiveness of various a-f noises in 
rendering speech and telegraph signals unintelligible. 

162 SPEECH TRANSMISSION 

Four general types of a-f interference were tested: 2 
resistance noise, speech, noise of the scanning type 
whose fundamental frequency is modulated so as to 
sweep through the voice hand in various manners, and 
noise consisting of trains of impulses. In addition, the 
noise generated in the generator developed in Project 
C-26 was studied. The data were obtained in terms 
of the per cent of discrete sentences rendered unin¬ 
telligible by the noise for various noise-to-signal ratios 
expressed in decibels, and were supplemented by spec¬ 
trograms of typical conditions, showing visual evi¬ 
dence of the degree of masking. The data from the in¬ 
telligibility tests are summarized in Table 1. 

The general conclusion from the work is that for a 
noise to have maximum effectiveness its spectrum 
must he continuous both in time and in frequency 
over the hand. Any appreciable gaps in either dimen¬ 
sion permit unmasked fragments of speech to be heard 
which convey considerable intelligence. A supplemen¬ 
tary conclusion is that visual inspection of its spec¬ 
trogram will usually indicate the general effectiveness 
of a given noise. At least it can he stated positively 
that if the noise spectrogram is of an open nature 
so that the pattern of superposed speech would be 
readily visible through it, the noise will be relatively 
ineffective in suppressing speech intelligence. 

“Project C-56, Contract OEMsr-626, Bell Telephone Labora¬ 
tories, Inc., Western Electric Co., Inc. 


The effective noises of those tested were resistance 
noise and mixtures of speech from two to twelve 
voices. These were about equally effective and required 
that the interference exceed the desired speech signal 
by about 10 db, as read by a volume indicator, for 
complete suppression of intelligence under the quiet 
listening conditions of the experiments. The curves 
were quite steep. If the noise was lowered 10 db below 
the above maximum, about nine out of ten sentences 
could be understood. Impulse and scanning types of 
noise were all much less effective, requiring from 7 
to 29 db higher noise levels than the above for equal 
loss of intelligibility. 


Table 1. Noise-to-signal ratios necessary to jam speech. 


Type of noise 

N/S ratios* in db for 
sentence errors of 
50% 90% 

Peak 
factort 
(db) 

Resistance noise 

4 

8.5 

10 

Speech interference 

Babble of 12 voices 

4.5 

9 


Mixture of 2 voices 

2 

7 


Scanning types of noises 

Stepped tones 

19-27 



Sawtooth scanning! 

Impulses 

16 

(23) 


Sharp impulses 

51 per sec 

33 


19 

118 per sec 

17 

25 

15 

Buzzers 

180 per sec 

11 

16 

10 

325 per sec 

15 

(25) 

9 

350 per sec 

15 


6 

Resistance noise interrupted 
at rate of 12 per sec§ 

0% of period 

4 

8.5 

10 

25% 

11 

(17) 

11 

50% 

30 

36 

13 

75% 

33 

37 

16 

90% 

28 

32 

20 


*N /S ratio refers to the difference in db between the rms noise power 
referred to one milliwatt and the speech volume read on a standard volume 
indicator in the standard manner. Figures in parentheses were derived by 
considerable extrapolation beyond the experimental data. 

tPeak factor is the height of the instantaneous peaks in db above the 
rms value of the wave. For comparison, the peak factor of speech is about 
13 db and of a sine wave is 3 db. If peak rather than rms values of the 
noise are of interest, the given N/S ratios should be increased by the 
peak factor. 

^Consists of a tone rich in harmonics the fundamental of which is 
periodically swept nearly linearly from 2,000 to 400 cycles w'ith instant 
return. Figures hold for scanning rates of 5 to 45 cycles per second. 

§The interruption was not quite complete but consisted of the sudden 
introduction of a loss of 30 to 38 db. The background noise, rather than 
the pulses, was controlling for interruptions greater than 50 per cent of 
the period. 


' CONFIDENT[AL 


119 











120 


STUDY OF INTERFERENCE GENERATION 


If the noise spectrum is continuous, the loss of in¬ 
telligibility is due to a true masking phenomenon de¬ 
pendent only on the hearing mechanism of the ear. 
That is, the listeners actually fail to hear the speech 
sounds at certain rather critical superposed noise 
levels. This can be explained by the accepted theory 
of hearing, for such a noise continuously stimulates 
all of the nerve endings on the basilar membrane in 
the cochlea and therefore interferes with the trans¬ 
mission to the brain over any nerve path of impulses 
recognizable as speech. The threshold of understand- 
ability of speech in the presence of this noise is there¬ 
fore about the same for all normal persons. On the 
other hand, if the noise is discontinuous or does not 
possess energy at all frequencies, some of the nerves 
at least some of the time are unstimulated by the noise 
and are free to transmit speech signals to the brain. 
The energy level of such noise must therefore be con¬ 
siderably greater to interfere with intelligibility, and 
moreover the listeners are aware that they can still 
hear the speech sounds even when they no longer 
can understand them. It was found that observers 
differed greatly in their ability to understand speech 
through such noise, since it depended largely on their 
psychological ability to disregard loud noise and to 
listen to and interpret the weak speech sounds that, 
being unmasked, can be heard through the noise. Ex¬ 
perience may greatly improve a listener’s ability to 
hear through this kind of noise. 

In the tests involving the use of noise consisting of 
speech from two voices, an attempt was made to add 
a psychological handicap to the observers by using for 
one of the voices the same voice (from a recording) 
that later spoke the sentences for the intelligibility 
tests and, moreover, repeating a similar list of sen¬ 
tences. In spite of this, the speech interference was not 
noticeably more effective than resistance noise, in 
terms of the peak powers required as measured by a 
volume indicator which integrates over about 0.3 sec¬ 
ond. A single voice was less effective than resistance 
noise because of the gaps between syllables, words, and 
sentences.The two- and twelve-voice interferences were 
equally effective within about 2 db and this doubtless 
would apply to any other intermediate number of 
voices. 

A further comment on impulsive noises is war¬ 
ranted. If a noise with a continuous spectrum, such 
as resistance noise, is interrupted for small percentages 
of the time, it rapidly loses in effectiveness. For ex¬ 
ample, Table 2 shows the ratio by which the average 
power of periodically interrupted resistance noise must 


be increased over that of uninterrupted noise for the 
same degree of jamming. The second column indi¬ 
cates the required power averaged over the interrup¬ 
tions and the last column the corresponding power in 
the pulses or, in other words, before interruption. 

Table 2. Comparison of interrupted and uninterrupted 
resistance noise. 


Required factor by which 
power must be multiplied 

% of total time Power averaged Power during 
Noise is interrupted* over long time pulse 


0 

1 

1 

12.5 

2.25 

2.6 

25 

5 

7 

50 

400 

800 

75 

1,000 

4,000 

90 

1,000 

10,000 


*The rate of interruption was four times per second for the 12.5% case, 
and 12 times per second for the others. 


Two important conclusions are apparent. 

1. If it is necessary to interrupt the interference in 
order to observe the wave to be jammed, the interrup¬ 
tions should be for as small a per cent of the total 
time as practicable. 

2. Nothing can be gained by causing a given band 
of interference to sweep across several channels so as 
to hit any one only part of the time (assuming that 
time constants of limiters, a-v-e circuits, etc., in the 
receivers do not affect the results). For example, if 
the noise is swept across four channels, it is on any 
one only 25 per cent of the time and is interrupted 
on each 75 per cent of the time. The figures for 75 
per cent interruption in Table 2 show that if the 
noise is distributed in this way among four channels, 
the total power must be multiplied by over 4,000 as 
compared with steady noise on one channel. If, in¬ 
stead, it were continuously applied to all four bands, 
the required total power would be increased only pro¬ 
portionally to the band covered, or by a factor of only 
4. The latter method is evidently the more efficient 
one by a ratio of more than 1,000 to 1. 

16.3 TELEGRAPH TRANSMISSION 

Two types of interference were tested for their ef¬ 
fectiveness in preventing experienced operators from 
reading hand-sent telegraph signals of a pitch of about 
935 cycles. 3,4 One interference consisted of resistance 
noise and the other was a sine wave whose frequency 
suddenly changed at intervals of 0.1 second among 
nine values lying between 580 and 1,500 cycles, one 
of which differed from the telegraph frequency by 










RESISTANCE NOISE 


121 


only 5 cycles. In the case of resistance noise, reception 
was completely interfered with when the noise inten¬ 
sity or power per cycle was 25 db below the marking 
telegraph signals, regardless of the band width, pro¬ 
vided it was greater than 120 cycles. If the band were 
limited to only 120 cycles, complete loss of intelligi¬ 
bility could be obtained with a total noise power 
which is about 4 db weaker than the marking tele¬ 
graph signal. The stepped frequency, when listened to 
through a 3,000-cycle band, completely interfered with 
the signals when its power was about 1 db stronger 
than the marking signals. When listened to through 
a 120-cycle band, which removed some of the inter¬ 
fering frequencies, the interference power had to be 
increased about 10 db for equal effectiveness. 

Telegraph signals were plainly visible in spectro¬ 
graphs through resistance noise when the noise power 
in a 3,000-cycle band exceeded the marking telegraph 
signal power by 5 and by 6.5 db, at which values the 
errors were 20 and 60 per cent, respectively. At 10 db, 
however, where the errors were 100 per cent, the signal 
is invisible. In the case of stepped frequency inter¬ 
ference, the signals were still visible at a level where 
they could not be read. This indicates that failure in 
this case was due to the mental confusion created 
by the interference rather than to a masking phe¬ 
nomenon. 

Of the two interferences tested, the resistance noise 
was appreciably superior in effectiveness when the 
band was narrow. It, of course, has the advantage that 
if it can be assured that the noise falls somewhere on 
the telegraph signals, it is immaterial what part of the 
noise band is close to the signal. The masking is 
caused by the noise within 50 cycles or so of the tele¬ 
graph frequency and this has the same character at 
all parts of the spectrum. It is possible that another 
telegraph signal located within a very few cycles of 
the signal to be jammed would be more effective than 


either of the noises tested. The practical possibility 
of achieving this depends upon other characteristics 
of the system, however, and it was felt that further 
tests should be made on a radio rather than an audio 
basis. 

16 - 4 RESISTANCE NOISE 

If resistance noise should be adopted as the inter¬ 
ference signal, the question of its generation and trans¬ 
mission would become important. One method which 
seems promising is to generate it at voice frequency 
(which is readily done by means of a gas-tube circuit) 
and to apply this to modulate an f-m transmitter. 
It is obvious that this would be a very effective inter¬ 
ference against f-m channels, since it would have a 
maximum effect on the limiters in the receivers and 
since the noise heard in the receiver outputs would be 
pure resistance noise. To obtain an idea of its probable 
effectiveness against a-m channels, the radio-frequen¬ 
cy spectrum of the interference signal was determined 
theoretically. 5 It was found that the relative energy 
per cycle versus frequency is substantially a normal 
distribution curve centered about the unmodulated 
carrier frequency. This curve would have a standard 
deviation corresponding to the frequency deviation 
which would be caused by impressing on the trans¬ 
mitter a steady (d-c) current of the same value as 
the mis value of the impressed noise. It appears that 
this noise would therefore be quite effective against 
a-m if the standard deviation is adjusted to corre¬ 
spond to the channel band width. 

From the practical standpoint, this method of gen¬ 
erating the interference would be very flexible, as the 
band width can be readily widened or narrowed to fit 
particular channels to be jammed by merely chang¬ 
ing the gain in the a-f circuits of the interference 
transmitter, thus changing the degree of modulation. 


CONFIDENTIAL 





Chapter 17 

RADIO INTERFERENCE GENERATORS 


Design and development of interference generators'* 
(jammers) covering the 2- to 20-mc and 15- to 30-mc bands. 
This summary covers work performed on Project C-25 1 prior 
to May 20, 1942, and on Project C-26 2 prior to June 29, 1942, 
after which respective dates both projects were transferred to 
Division 15 where more extensive work was done on this 
problem. 

Hi FUNDAMENTAL STUDIES 

N the book Speech and Hearing 3 the masking of 

pure tones by other pure tones, as well as the mask¬ 
ing of the former by complex sounds, is discussed. 
As is generally the case in this type of subject, no 
simple all-inclusive rules can be given. Certain gen¬ 
eralizations, however, can be made. 

1. A low-frequency tone can obliterate a high-fre¬ 
quency tone if the former is at a sufficient level (say 
100-db sensation level). 

2. A high-frequency tone in general cannot mask 
a low-frequency tone. 

3. The threshold shift for a masked tone (required 
increase in level above original threshold value in order 
that it may again be perceived) is greatest when the 
masking tone is close to it in frequency. In this special 
case, it is possible for a higher-pitched tone to mask one 
of lower pitch, a contradiction or exception to 2 above. 

The reason advanced for the greater effectiveness 
for masking of a low-frequency tone is that if it is 
of sufficient intensity it is able to produce subjective 
overtones in the ear due to the nonlinearity of the 
latter, and one or more of these overtones will be close 
to the high-frequency tones to be masked, and hence 
(by 3) very effective in doing so. The reason ad¬ 
vanced for 3 is that the masking tone has stimulated 
the particular nerve fibers of those terminating in the 
basilar membrane that normally respond to the fre¬ 
quency to be masked, and caused them to discharge 
their unit loads, so that they can no longer respond 
to the frequency to be masked. 

4. A complex sound produces further masking by 
virtue of the cross-modulation effects (summation and 
difference beat frequencies) as well as by means of 
the harmonic overtones which it creates subjectively 
in the ear. 

“Project C-25, Contract OEMsr-89, Farnsworth Television & 
Radio Corp.; and Project C-26, Contract OEMsr-285; Federal 
Telephone & Radio Corp. 


No conclusions are presented in the above reference 
as to the relative effectiveness for masking of single 
tones and complex sounds. However, the loudness of 
a sound whose energy is distributed over the spectrum 
appears greater than that of a sound of equal inten¬ 
sity whose energy is concentrated in a narrow band, 
provided the hearing mechanism is thereby subjected 
to sound levels of 40 db or higher. 4 This might indi¬ 
cate that a complex tone is more efficacious for mask¬ 
ing than a single tone where such masking is desired 
for a range in the spectrum rather than for one par¬ 
ticular frequency. 

1711 Masking by Clicks and Pulses 

Isolated and recurring clicks and pulses appear to 
gain more rapidly in loudness with increase in level 
than a steady 1,000-cycle tone. 4 This would indicate 
that for effective masking, the interfering sound 
should be relatively high in intensity. More quantita¬ 
tive data will be presented below. Measurements of the 
effect of disturbing noises and single-frequency tones 
upon a telephone headset which has a fairly flat 
frequency response indicate that for pure tones, the 
1,000-cycle tone has the most disturbing effect, al¬ 
though for the telephone set employed frequencies 
from 330 to 3,500 cycles per second are within 10 db 
of the 1,000-cycle tone in disturbing effect. 

1712 Masking by Noise 

Noise in the frequency range containing the trans¬ 
mitted speech most important to understanding in¬ 
terferes with understanding more than noise in some 
other frequency range, although it is true that certain 
noises produce a harmful effect through annoyance 
rather than through masking. 5 

Experimental tests 3 indicate that the threshold shift 
of speech sounds produced by noise is about the same 
as the average shifts produced on a 500-, 1,000-, and 
2,000-cycle tone. In another paper, 6 it is stated that in 
the region of 1,000 cycles a tone of one frequency can 
just be perceived in the presence of thermal noise when 
its intensity is about 60 times the noise intensity per 


122 




CONFIDENTIAL 


i 





FUNDAMENTAL STUDIES 


123 


cycle for noise bands no narrower than 60 cycles. 
Thus, for example, for a 6-kc band, the signal need 
be only about 1/100 of the total noise intensity, i.e., 
20 db below the latter. 

17,1-3 Masking of Code Signals 

An unpublished memorandum 7 throws considerable 
light on the interference of telegraph code signals. 
This report indicates that for a 3,000-cycle band 
width, thermal noise must be 10 db stronger than the 
signal to cause a sharp break in the ability to receive 
the message. Complete obliteration of the signal re¬ 
quired that the noise be 17 db stronger than the signal. 
An alternative form of disturbance employed was that 
of nine tones, 580, 1,010, 1,320, 810, 1,100, 940, 1,260, 
650, and 1,500 cycles, produced in sequence at a rate 
of approximately one such group per second. This 
“jumping” tone for the 3,000-cycle band width, had 
to be but 2 db higher than the signal to disrupt the 
transmission. The above frequencies require only 
about one-half the band width that the thermal noise 
was permitted to occupy. 

The telegraph pitch w - as 935 cycles. According to 3, 
most effective masking would be accomplished by fre¬ 
quency components close to this in value. This seems 
to be borne out by a test in which a 120-cycle band¬ 
pass filter (935 cycles mid-frequency) was used in 
cascade with the rest of the system. Actually this filter 
was employed to simulate the sharp tuning capabil¬ 
ities of a receiver, but the results indicated that the 
narrower 120-cycle band of thermal noise was as effec¬ 
tive as the original 3,000-cycle band in masking the 
signals when it was 4 db below the signal in intensity. 
This indicates, as stated above, that only the compo¬ 
nents close to 935 cycles are effective in masking, 
and that those remote from this frequency represent 
waste energy as far as interference is concerned. 

1714 Further Discussion 

The band-pass filter mentioned above was also em¬ 
ployed with the jumping tones. The input to the filter 
in this case had to be increased from the original ±2 
db to ±11 db. This is probably due to the fact that 
only the 940-cycle tone came through with any appre¬ 
ciable intensity, although some of the other tones 
could be heard. While some interesting conclusions 
might be assumed from this report, it must be remem¬ 


bered that it is only a preliminary one and is subject 
to further verification. For instance, if only the 940- 
cycle tone came through the 120-cycle filter with any 
appreciable intensity, then the effective interfering 
energy was essentially in pulses of 1/9-second dura¬ 
tion, coming at a repetition rate of one per second. 
The average energy would be 1/9 of the peak. The 
increase in required energy when the filter is employed 
is 9 db, which is eight times the energy. This is close 
to nine times the energy required during 1/9 of a sec¬ 
ond to give the same average energy as when the filter 
was not employed. Thus it might be argued that the 
increased level for the jumping tone was due simply 
to the fact that the band-pass filter cut down the 
energy transmitted. On the other hand, many of the 
tones were remote from the telegraph signal in fre¬ 
quency and hence probably not very effective in mask¬ 
ing the latter. We may be permitted to conclude, how¬ 
ever, that the only virtue of noise over any other type 
of interference is that no matter how limited the re¬ 
ceiving channel is made in frequency range, there will 
be available noise components in that range, since 
noise is a random signal whose spectrum consists of 
components of all frequencies and all of equal magni¬ 
tude. 


17,1,5 Criteria for Interference 

All the references cited, except the last, discuss the 
complete obliteration or masking of tones. For suc¬ 
cessful interference it is not necessary to mask com¬ 
pletely, but only to reduce the intelligibility of the 
transmission below an acceptable limit. In commercial 
practice, this limit is 60 per cent successful transmis¬ 
sion. Probably a lower limit is acceptable in military 
practice, but the exact value does not seem to be knowm 
yet. The interference level for this purpose will be less 
than that for complete obliteration, how much less 
depends upon whether the test is one for disconnected 
words (discrete word intelligibility) or for discon¬ 
nected sentence (discrete sentence intelligibility). 8 If 
complete masking is obtained, it is of course obvious 
that no intelligence can be transmitted. 

In connection with these facts it should be noted 
that the interference curves are quite steep. For ex¬ 
ample, the range from 50 per cent to substantially 0 
per cent successful transmission is covered by a change 
in the level of the interfering resistance noise of only 
7 or 8 db for speech and about half that for telegraph. 


CONFIDENTIAL 






124 


RADIO INTERFERENCE GENERATORS 


The exact value of the limit chosen as acceptable is 
therefore not very important practically speaking. 
Whether one chooses 60 per cent or 40 per cent suc¬ 
cessful transmission as the limit affects the required 
interference by only about 1 db, which is small com¬ 
pared to the other uncertainties involved. 

1716 Work Done under Project C-26 

One conclusion which had been drawn early was that 
the form of modulation of the interference generator 
was of secondary importance. At first pulses of essen¬ 
tially rectangular form and varying width were em¬ 
ployed, interrupted periodically to permit monitoring 
of the enemy signal. This form of modulation was 
subsequently changed to one in which the pulses varied 
in pitch at a low rate, say one every 2 seconds. The 
width of the pulses was adjustable, as was the rate of 
“warbling.” 

Articulation tests indicated that discrete word intel¬ 
ligibility was only about 30 per cent for the latter 
form of modulation as compared to about 50 per 
cent for the former type, which was an appreciable, 
but not outstanding, improvement. The interfering 
sound in these tests was about 6 db above the speech 
sounds. Tests at Fort Monmouth gave similar results 
after the tuning adjustment was improved. 

The final form of modulation described above oc¬ 
cupies the region of speech frequencies, is of complex 
form and sufficiently irregular to be considerably dis¬ 
tracting and hence annoying. It thus essentially meets 
the requirements for interference detailed above. 
Moreover, it is produced by a simple circuit employing 
two thyratron tubes, and, as stated previously, can be 
adjusted as to pitch and as to rate of warbling. Fur¬ 
ther development along this line did not appear jus¬ 
tified at the time. 

1717 Military Considerations 

Of greater importance than the foregoing are the 
military considerations involved. Among these are 
the questions as to whether several communications 
are to be jammed simultaneously or only one at a time, 
whether the link consists of two parties or more than 
two, whether the frequencies involved are fixed (crys¬ 
tal-controlled transmitters) or continuously adjust¬ 
able, and also what they are, whether switching from 
one frequency to another is possible or not, and 
whether a-m, f-m, c-w, or i-c-w signals are to be 
jammed. 


17.1.8 Requirements Met by Project C-26 

Interference Generator 

Since the project was fundamentally exploratory in 
nature, equipment was built to meet the following 
requirements: monitor a band of frequencies, about 
3.0 me wide at one time, cover a range of 15 to 30 me 
in two bands, jam a-m, f-m, c-w, or i-c-w signals, and 
handle communication systems in which switching 
from one frequency to another is employed by the 
enemy to avoid interference. 

17.1.9 Electrical Considerations 

The discussion on the masking of tones dealt with 
the effects of interfering sounds upon a signal after 
these had been conveyed to the ears of the observer. 
There are, however, electric circuits to be traversed be¬ 
fore the composite of signal and noise reaches the ears, 
and it is in this portion of the system that a great 
amount of ingenuity has been exercised in the past 
to obviate interference. The precautionary measures 
employed must now be circumvented. 


17.1.10 Tuning Requirements 

For interference to be effective, the jammer must be 
accurately tuned to the enemy frequency. Tests made 
in the laboratory and confirmed by those performed 
at Fort Monmouth indicated that even in the case of 
a-m speech reception, accurate frequency alignment 
was necessary, since otherwise very little interference 
energy would get into the system unless the impinging 
energy was prohibitively great. The alignment had to 
be within 1 kc or better. 


17.1.11 Original Method of Tuning 

The original method of frequency alignment was 
that of having a spectrum-scanning receiver, which 
portrayed during one scan the frequency to be jammed 
(among many others that might be on the air) and 
during the next scan the frequency of the interference 
generator. Alignment was deemed adequate when the 
two signals (resonance curves) were placed in juxta¬ 
position. To facilitate this, a narrower scan was em¬ 
ployed for a final adjustment. 


CONFIDENTIAL 





FUNDAMENTAL STUDIES 


125 


17.1.12 Present Method of Tuning 

This, however, was found to be inferior to a method 
subsequently developed. The wide scanning was re¬ 
tained and gave the operator a visual indication of the 
stations on the air and told him whether or not the 
enemy signal to be jammed had shifted to a new fre¬ 
quency to elude interference. For fine tuning, the hor¬ 
izontal sweep on the oscilloscope screen was retained 
and serves as a linear time base during those periodic 
intervals of time when scanning occurs. The receiver, 
however, is “fix-tuned” during this time to the enemy 
signal by a finer adjustment of the tuning capacitor. 
The result, during fine tuning, is that the enemy car¬ 
rier appears periodically on the screen, and the nature 
of the signal, whether a-m, speech, or c-w, can be 
easily determined. 

Apparently, the reason why the scanning method is 
not sufficiently accurate is that the interference signal 
is in the form of pulses unrelated in timing to the 
small interval when the scanning capacitor is tuning 
through the interference carrier frequency. As a result, 
sometimes two pulses, sometimes three, occur during 
such an interval, and hence the resonance peak 
changes in amplitude in an erratic, jumpy manner, 
which makes the alignment of it with the center cross 
hair a difficult matter. On the other hand, a much 
larger number of pulses appear during one entire 
scan (1/50 second) and hence the picture in the tun¬ 
ing position is steadier. 

17113 Appearance of Signals on Screen 

The transmitter portion of the interference gener¬ 
ator is ganged with the receiver with a vernier control 
to permit it to be more accurately tuned to the re¬ 
ceiver, which has already been fix-tuned to the enemy 
signal. The interference signal also appears on the 
oscilloscope screen in alternate scans to those present¬ 
ing the enemy signal. All this is possible because the 
transmitter is keyed on and off periodically. During 
the time it is on, it swamps out the enemy signal in 
the receiver. (The receiver is blocked by a negative 
voltage during this time, so that it will not be over¬ 
loaded by the transmitter.) During the time the latter 
is off, the enemy signal has a chance to come through, 
especially since the sensitivity of the receiver has been 
restored by the removal of the blocking voltage. 

The two signals appear superimposed upon the 


screen. Complete alignment is attained when both 
signals appear at maximum amplitude. The method 
is simple and accurate but unfortunately gives no in¬ 
dication as to the direction in which the enemy may 
have changed frequency to elude interference nor his 
new position in the spectrum. Hence the spectrum 
scanning is normally used to monitor the enemy sig¬ 
nal, and fine tuning used momentarily after his sig¬ 
nal has been located by the former method. 

171,14 Interference for Frequency 
Modulation 

In the case of frequency modulation, a further re¬ 
duction in interference is possible if the desired signal 
is at least 6 db higher than the interference. On the 
other hand, if the latter is 6 db higher than the former, 
it will take control of the limiter and tend to obliter¬ 
ate the signal. Tests performed with a fixed carrier of 
fixed amplitude seemed to corroborate this, but if the 
carrier pulsated in amplitude as described above, the 
control was insufficient, particularly on the Army f-m 
transceivers. 

The transceivers have a 10-^sec release time on the 
limiters. If the interfering carrier is amplitude-mod¬ 
ulated in that it is keyed on and off, then the short 
receiver release time will permit another signal, such 
as that of the enemy, to take control of the receiver 
during the intervals when the interfering carrier is 
off. The enemy signal can thus come through with, 
at most, a flutter, but nevertheless intelligible. 

By wobbling the interference carrier frequency dur¬ 
ing the on periods, jamming was practically complete. 
Apparently the carrier, when on, not only obliterated 
the signal, but produced a tone as well, which was 
sufficiently distracting to mask the enemy signal dur¬ 
ing the times that the latter was on and the tone was 
off. In this way, amplitude modulation could be 
jammed without wobbling the frequency and fre¬ 
quency modulation merely by turning on, in addition, 
a motor-driven rotating capacitor, thus wobbling the 
frequency. The number of operations to be performed 
could thus be kept at a minimum. 

It was undesirable to wobble the frequency for a-m 
interference, since the energy was spread out over an 
unnecessarily wide portion of the spectrum, and hence 
less was available at the frequency of the a-m carrier 
to be jammed. For a frequency deviation of about 16 
kc, the reduction in power at the carrier or average 
frequency was found to be about 15 db. 


CONFIDENTIAL , 





126 


RADIO INTERFERENCE GENERATORS 


17115 Effect of Noise Limiter 

A preliminary investigation pointed to the desira¬ 
bility of operating the transmitter power oscillator so 
as to obtain momentary peak powers greatly exceeding 
the normal output of the tube. This could be accom¬ 
plished by periodically keying the tube with pulses. As 
first constructed, the tube was keyed for about 1/50 
second and off for 3/50 second, during which time 
scanning and monitoring of the enemy could be 
performed. 

This, however, proved to be an incorrect attack on 
the problem because of the electrical characteristics 
of receivers designed to circumvent such disturbances 
(ordinarily static pulses), that is, noise- or crash- 
limiter circuits. These may be of the form of a diode, 
biased by the average value of the carrier so that it 
is inoperative until the carrier exceeds twice its aver¬ 
age value. Static pulses, which are momentary pulses 
of high amplitude, will operate the diode, which shorts 
them out. Such a circuit will also short out high in¬ 
terference pulses of the type described above without 
affecting the gain of the receiver during the off peri¬ 
ods of the transmitter. If these latter periods are each 
of 3/50-second duration, sufficient enemy signal can 
come through during these times to nullify the mask¬ 
ing tone during the other 1/50 second, since it has 
been greatly reduced in amplitude by the noise-limiter 
circuit. Such was found to be the case in tests per¬ 
formed at Fort Monmouth. 

If no noise-limiter circuit is employed but the AYC 
has a sufficiently fast release time, then the receiver 
may recover its gain during the 3/50-second period 
to permit a sizable enemy signal to come through too. 

In any event, the solution is to reduce the peak 
power for a given average power and permissable plate 
dissipation by prolonging the on period for the trans¬ 
mitter and reducing the modulation voltage. This 
was done: the on period was made 3/50 second, and 
the off period 1/50 second, which is sufficient time 
for monitoring. The interference was much more effec¬ 
tive after this change was made. 

17.1.16 Degree of Modulation 

Finally, it is evident that maximum interference 
sound output is produced in the receiver if the latter 
is operated at maximum effect, that is, if the interfer¬ 
ence carrier is modulated 100 per cent in the case of 
amplitude modulation or over the maximum devia¬ 


tion capabilities of the receiver in the case of frequen¬ 
cy modulation. (More than 100 per cent amplitude 
modulation will be removed by the noise limiter.) 
Such modulation characteristics are now more nearly 
met by the 3/50 second on and 1/50 second off char¬ 
acteristic. 

17.2 GENERAL DESCRIPTION OF THE 
C-26 JAMMER 

The interference generator developed under Proj¬ 
ect C-26 consists essentially of a tunable transmitter 
and scanning receiver powered by an inverter fed from 
a 24-volt battery, with transmitter and receiver ganged 
together mechanically and synchronized in time so as 
to afford an interval for jamming and one for mon¬ 
itoring the enemy signal. 

The inverter drives a scanning capacitor and a 
switch in the r-f tuner so as to generate synchronizing, 
blanking, and sawtooth waves in addition to tuning 
the receiver cyclically over approximately a 3-mc 
range in the spectrum. 

The sawtooth wave furnishes suitable horizontal 
deflection for an oscilloscope. The blanking signal 
blanks out the return trace on the oscilloscope screen. 
The synchronizing signal locks a multivibrator in the 
modulator unit at half its frequency for keying and 
modulating the transmitter unit. 

The output of the modulator actuates the trans¬ 
mitter so that it is on for one period of time and off for 
a shorter period, during which the enemy signal may 
be observed. A portion of the transmitter output is 
picked up by the receiver to afford observation of the 
interference signal on the oscilloscope screen as well 
as the enemy signal, but in alternate sequence. To pre¬ 
vent overloading of the receiver when the interference 
signal is on, the gain of the receiver is reduced at such 
times by a blocking voltage, also obtained from the 
modulator unit. 

17 21 Results Secured 

Field tests indicated that for reasonable distances 
between jammer and enemy receiver an interference 
power of about 25 watts is required for jamming the 
usual field transmitters with an output of about 5 
watts. This power is confined to a band width not ex¬ 
ceeding 5 kc and much less than this for CW. Thus 
the power required for jamming a single channel is 
not very great. 





WORK DONE UNDER PROJECT C-25 


127 


If, however, a wider band is to be jammed, for 
example, 0.5 me, simultaneously, then the power 
must be 2,500 watts; for a 1-mc range it must he 
5 kw; and if the enemy shifted frequency as much as 
5 me the power required to disrupt his communica¬ 
tions in spite of his frequency shifts would have to be 
25 kw, which would be too great for mobile operation. 

If the energy is spread by employing narrow modu¬ 
lation peaks (pulses) then a noise limiter in the re¬ 
ceiver will partially or perhaps completely vitiate the 
effectiveness of the jammer. The use of a spark trans¬ 
mitter with its well-known wide hand was suggested 2 
as an interesting line of development that might yield 
profitable results. 

The final report on Project C-26 2 contains other 
discussion of such matters as narrow-band interfer¬ 
ence, the advantages of separation of keying and mod¬ 
ulation, of the use of a common antenna, of the vir¬ 
tues of using two receivers, one for scanning and one 
for tuning, of using motors to eliminate mechanical 
coupling of receiver and transmitter and other per¬ 
tinent factors leading to the design of effective jam¬ 
mers. 

17.3 WORK DONE UNDER PROJECT C-25 

The purposes of Project C-25 were as follows. 

1. To investigate the possibility of locating an en¬ 
emy transmission accurately and quickly within a 
band approximately 20 me wide. 

2. To jam the enemy within a relatively narrow 
band (3 kc) without interfering with other commu¬ 
nications. 

3. To investigate the type of transmission which 
would most effectively jam i-c-w communications. 

4. To accomplish this within a range of a few miles 
with minimum power. 

5. To construct complete airborne equipment in the 
minimum space and weight required. 

1731 Accomplishments 

A panoramic receiver was used to spot the enemy. 
This receiver was suitable over the 18-mc range and 


three bands. Various widths of spectrum could be 
examined by expanding the sweep. Spotting was by 
visual indication but phones were provided for listen¬ 
ing to determine the advisability of jamming. 

Having spotted the enemy, a transmitter was tuned 
to the required frequency and was energized by a 
trigger. The transmitter signal appeared on the pan¬ 
oramic receiver opposite that of the enemy so that the 
transmitter could be tuned accurately before pressing 
the trigger. 

A “step c-w” type of modulation was employed in 
which the transmitter frequency was shifted within 
4 kc at a random rate. It was found, however, that a 
random tone, modulated and applied to the step trans¬ 
mitter signal, was more effective. 

Normal communication signals could be jammed 
at a distance of approximately 10 miles from a plane 
5,000 ft in the air. 

The complete equipment weighed approximately 
100 lb and occupied about 5 cu ft. 

17 3 2 The End Results 

The effectiveness of jammers of the types developed 
under Projects C-25 and C-26 may be made evident by 
the following excerpts from reports on their use dur¬ 
ing field maneuvers in Louisiana. “A complete mech¬ 
anized division of the Army was delayed two hours in 
starting operations because its communications were 
jammed by the equipment. It was necessary to resort 
to motorcycles for communication before the opera¬ 
tion could get under way.” From another communica¬ 
tion, “... the presence of the equipment was not 
known to anybody except the highest ranking officers. 
It proved to be brilliantly successful and disrupted 
maneuvers so completely that the men in tanks had 
to stop operations and instead got out of their vehicles 
and ate their lunch!” 

After these tests in this country, some 50 of the sets, 
covering various frequency ranges, were ordered for 
use at once on the battlefields, even though the dem¬ 
onstrations indicated that improvements of consider¬ 
able value could be made. 


I 


CONFIDENTIAL 


ft 























































PART VI 


RADIO TRANSMISSION FORECASTING 


confide: 























































•• V v. 














Chapter 18 

IONOSPHERE STUDIES 


F ive laboratories participated in a cooperative 
project on radio transmission conditions, measur¬ 
ing the field intensities of numerous transmitting sta¬ 
tions on various frequencies from 660 to 15,355 kc, 
and continuing measurements on conditions in the 
ionosphere. At the completion of individual contracts, 
work was continued under the Interservice Radio 
Propagation Laboratory [IRPL], set up by the Joint 
IT. S. Communications Board to furnish comprehen¬ 
sive radio propagation service to the Armed Services. 

The following summary is written from the final 
reports of the individual projects, the bibliography 
showing the relation between the several contracts and 
their extensions. Related work will he found in Part 
II of Volume 1, Division 13, on direction finding and 
antennas. 

i8i INTRODUCTION 

At the time the projects a were started there had 
been no systematic attempt to forecast the radio and 
ionospheric storms or disturbances which interrupt 
long-distance h-f radio communication. Sufficient 
background of experience had been acquired, especi¬ 
ally at National Bureau of Standards [NBS] and at 
the Department of Terrestrial Magnetism, Carnegie 
Institution of Washington [CIW] to indicate the 
promising nature of a broad study of the type de¬ 
scribed below. Some success had already been achieved 
in short-time forecasting of radio disturbances, using 
very limited data. 

Two general types of data were collected under the 
several projects, one dealing with geomagnetic phe¬ 
nomena (magnetic activity) and solar phenomena and 
the other with ionosphere characteristics and field- 
intensity measurements. The solar data were collected 
by a chain of solar observatories which submitted to 
the Department of Terrestrial Magnetism daily re¬ 
ports which were correlated with available magnetic 

a C-9, no contract, National Bureau of Standards [NBS]; 
C-13, no contract, NBS; C-14, Contract OEMsr-200, Carnegie 
Institution of Washington [CIW]; C-20, Contract OEMsr-227, 
Stanford University [SU]; C-22, Contract OEMsr-378, 
Louisiana State University [LSU]; C-44, no contract, NBS; 
C-45, Contract OEMsr-632, University of Puerto Rico; C-46, 
Contract OEMsr-573, LSU; C-47, Contract OEMsr-590, 
SU; C-48, Contract OEMsr-558, CIW; C-49, no contract, 
NBS; C-53, Contract OEMsr-594, CIW. 


information (Project C-53). For the ionosphere and 
radio data, NBS, under Projects C2-13 and C2-49, 
acted as the centralizing agency. 

182 ACCOMPLISHMENTS 

A brief summary of the work accomplished under 
the several projects will he found below but the entire 
effort may he summarized as follows: Improvements 
were made in methods of determining maximum 
usable frequencies, lowest usable frequencies, skip dis¬ 
tances, and distance ranges. A method of calculation 
of the absorption index over a path was devised. A 
new service of short-time forecasting of radio trans¬ 
mission conditions was developed, providing forecasts 
for a week ahead. World charts showing predicted 
variation of maximum usable frequencies based on ob¬ 
servations at all of the cooperating stations were pre¬ 
pared. 

The methods and practices developed were turned 
over for use of the IRPL, set up by the Joint U. S. 
Communications Board to furnish comprehensive 
radio propagation service to the Armed Forces. At the 
end of the fiscal years of the projects, the work was 
fully under way and NDRC was able to close out its 
work relating to radio propagation predictions as a 
completed phase of the research, sufficient to permit 
the beginning of an operational program. 

Cooperating Laboratories 

The laboratories participating in the program of 
the ionosphere and radio field-intensity observations 
were National Bureau of Standards (centralizing lab¬ 
oratory), Washington, D. C.; Carnegie Institution of 
Washington, College (near Fairbanks), Alaska; Lou¬ 
isiana State University, Baton Rouge, Louisiana; 
Stanford University, Palo Alto, California; University 
of Puerto Rico, San Juan, Puerto Rico. 

These stations observed field strengths and iono¬ 
sphere conditions of the following transmitting sta¬ 
tions : 


Station 

Frequency in kc 

Location 


Recorded at Baton Rouge, La. 

wwv 

5,000 

Beltsville, Md. 

W8XAL 

6,080 

Mason, O. 

COCH 

9,435 

Havana, Cuba 

XEWW 

9,500 

Mexico City, Mex. 

WWV 

15,000 

Beltsville, Md. 




132 


IONOSPHERE STUDIES 


Recorded at College, Alaska 


KGEI 

7,250 

San Francisco, Calif. 

KEP 

9,4S0 

San Francisco, Calif. 

KGEI 

9,550 

San Francisco, Calif. 

WLWO 

9,590 

Mason, O. 

JAP 

9,595 

Tokyo, Japan 

DJW 

9,650 

Zeesen, Ger. 

WRCA 

9,670 

Bound Brook, N. J. 

JAP 

9,672 

Tokyo, Japan 

DJX 

9,675 

Zeesen, Ger. 

WNBI 

9,690 

New York, N. Y. 

WRUW 

9,700 

Boston, Mass. 

GSD 

11,750 

Daventry, Eng. 

WRITE 

11,790 

Boston, Mass. 

JVZK 

11,815 

Tokyo, Japan 

WBOS 

11,870 

Boston, Mass. 

GSI 

15,260 

London, Eng. 

KWID 

15,290 

San Francisco, Calif. 

KGEI 

15,330 

San Francisco, Calif. 

WGEA 

15,330 

Schenectady, N. Y. 

KWU 

15.355 

San Francisco, Calif. 


Recorded at San Juan, Puerto Rico 

WJZ 

770 

New York, N. Y. 

WWV 

5,000 

Beltsville, Md. 

COCH 

9,435 

Havana, Cuba 

GLH 

13,525 

Dorchester, Eng. 


Recorded at Pa 

llo Alto, Calif. 

WWV 

5,000 

Beltsville, Md. 

CBRX 

6,160 

Vancouver, B. C. 

COCH 

9,435 

Havana, Cuba 

XEWW 

9,500 

Mexico City, Mex. 

GSD 

11,750 

London, Eng. 

WRUL 

11,790 

Boston, Mass. 

JZJ 

11,800 

Tokyo, Japan 


Recorded at Washington, D. C. 

WEAR 

660 

New York, N. Y. 

WLW 

700 

Mason, O. 

WBBM 

780 

Glenview, Ill. 

WCKY 

1,530 

Covington, Ivy. 

WQXR 

1,560 

New York, N. Y. 

WWV 

5,000 

Beltsville, Md. 

W8XAL 

6,0S0 

Mason, O. 

Data secured from such 

radio field-intensity meas- 

urements 

furnish a means 

of checking, from actual 

operating 

conditions, the computations of maximum 

usable frequencies and are 

: useful in confirming the 

existence of ionosphere storms and in indicating their 

intensities 

and the areas 

and frequencies affected. 

They are 

the chief source of detailed information 

about sudden ionosphere 

disturbances and indicate 

the time of their beginning, their intensity, and their 

duration. They can be correlated with solar and mag- 


netic data to provide additional information for the 
prediction of radio transmission conditions. They are 
used to determine the amount of absorption of sky- 
wave transmission for different latitudes, frequencies, 
and lengths of paths, so that the decrease in field in¬ 


tensity in addition to the inverse distance factor may 
he estimated. 15 

The final report on Project C-49 11 gives the math¬ 
ematical background for calculating additional atten¬ 
uation of radio waves by absorption, for determining 
maximum usable frequencies (see also the final report, 
Project C-9, Radio Transmission Handbook), 1 lowest 
usable high frequencies and distance ranges. Certain 
of this analysis is over-simplified and many techniques 
have been greatly improved and knowledge expanded 
since the dates of these projects. This analysis, how¬ 
ever, gives a clear, concise, and readable picture of the 
ionosphere and of the manner in which, at the time 
of writing, ionosphere and field-intensity data could be 
usefully applied to radio communication predictions. 

1821 Apparatus Employed 

In general, the equipment used to obtain field-in¬ 
tensity records consisted of a communications receiver 
in conjunction with a continuous automatic recorder. 
The arrangement is essentially a recording r-f volt¬ 
meter, the voltages in question being those at the in¬ 
put to the receiving set. In particular, the receiver 
was usually an HRO or an HQ-120-X, the recording 
mechanism being of the Micromax line-drawing type 
of recorder or a Brown instrument. 

Ionosphere data were collected by means of fixed- 
frequency pulse transmitters or by means of sweep- 
frequency pulse transmitters, the returning echoes be¬ 
ing displayed on a cathode-ray tube screen, photo¬ 
graphs of which were taken at regular intervals. The 
set up at Stanford University is typical. “Each of 
the transmitters produced single side-band pulses 
at the rate of 20 per second. A balanced modulator 
combined the output of a variable oscillator with that 
of a 460-kc i-f oscillator and excited a final amplifier 
which developed a peak power of approximately 4,000 
watts. A complete sweep of the range was photo¬ 
graphed every half-hour on the screen of a cathode- 
ray tube which was intensity-modulated by the signal 
from the receiver. Virtual heights up to about 700 
km were recorded.” 4,9 

The recorders in the field-intensity work operated 
from a bridge circuit in the a-v-c circuit of the re- 

b For a description of the part played by ionosphere pre¬ 
diction and information making possible the choice of best 
frequencies for radio sky wave propagation over known dis¬ 
tances and paths, particularly in setting up the Loran system of 
long-range navigation aid, see reference 13 of the bibliography. 


CONFIDENTIAL 






CONCLUSION 


133 


ceiver. Antennas were dipoles properly matched to the 
transmission lines connecting them to the receiver and 
oriented to secure maximum signal from the trans¬ 
mitters being recorded. The bridge circuit connecting 
the HEO receiver to the Micromax recorder (Figure 
1) is taken from the final report 7 of Project C-45. 


18,000 



Figure 1. Bridge circuit connecting HRO receiver to 
Micromax recorder. 


More detail on the methods of making echo-height 
measurements will be found in reference 14 of the 
bibliography. At the University of Puerto Rico 7 iono¬ 
sphere recording antennas were vertically directed 
rhombics, one having 170-ft legs and another having 
115-ft legs, the two being connected in series to give 
reasonably uniform impedance characteristics over the 
3- to 12-me range covered in the measurements. Rliom- 
hics were employed in receiving from the stations 
selected to be monitored for field intensities. 

18-3 SOLAR AND GEOMAGNETIC 
OBSERVATIONS 

Under Project C-53 12 a correlation of solar and 
geomagnetic observations with conditions of the iono¬ 
sphere was carried out. Weekly reports of magnetic 
activity were received from the stations of the U. S. 
Coast and Geodetic Survey at Cheltenham, Maryland; 


Tucson, Arizona; Sitka, Alaska; Honolulu, Hawaii, 
and San Juan, Puerto Rico; and from the magnetic 
observatories of the Department of Terrestrial Mag¬ 
netism at Watheroo, Western Australia; Huancayo, 
Peru, and College, Alaska. 

Solar observations were conducted for this service 
at the following places: Climax, Colorado, U. S. Naval, 
McMath-Hulbert, and Mount Wilson observatories. 

A study of the progress of sun-spot cycles and of 
solar-activity cycles, magnetic disturbances, coronal 
intensities, sporadic E-region ionization, radio fade- 
outs, aurora displays, and radio conditions during the 
partial solar eclipse of February 4, 1943, at College, 
Alaska, was made under the Projects conducted by the 
Department of Terrestrial Magnetism. 3,10,12 

184 CONCLUSION 

The advantages to the Military of having predic¬ 
tions of radio communication effectiveness in plan¬ 
ning campaigns are obvious. In 1942 it became im¬ 
portant to have certain forecasts some weeks in ad¬ 
vance and other types a few days in advance. Radio 
and ionosphere forecasts of the first kind were pre¬ 
pared and distributed by IRPL. Those of the second 
kind were prepared for a few months by CIW, and in 
October 1942, this forecasting function was trans¬ 
ferred to IRPL, while the function of centralizing 
solar and cosmic data for this purpose was retained 
by the CIW. The final report, Project C2-49, 11 gives 
an example of the weekly forecast and also of the type 
of information that was distributed by telephone to 
proper services. As an example of special warnings, on 
October 28, 1942, an ionosphere storm was forecast 
and the information distributed by telephone a few 
hours in advance, thus warning communications ser¬ 
vices that trouble might be experienced. 

Projects C-9 and C-44, NBS, resulted in radio 
transmission handbooks for frequencies of 1,000 to 
30,000 kc and show conditions under which the differ¬ 
ent radio frequencies would be useful for the time 
periods covered, the winter of 1941-42 and the sum¬ 
mer of 1942. 


CONFIDENTIAL 


L 








































PART VII 


APPARATUS DESIGN 


CONFIDENTIAL 










































Chapter 19 

U-H-F FIELD-INTENSITY MEASURING EQUIPMENT 


Development of two field-measuring sets, including receiv¬ 
ers, signal generators, and antennas, for the regions 300 to 
1,000 and 1,000 to 3,000 me. The first units of the kind cover¬ 
ing these frequencies were developed under this project. 

i9.i INTRODUCTION 

he object of the research conducted under this 
project 1 * was to develop field-strength measuring 
equipment for the u-h-f and microwave regions which 
were coming to be of the utmost importance for radio 
communication and radar work beginning to be used 
at the time, early 1941. 

At the time there was no commercial field-strength 
equipment of any kind for use at frequencies above 
50 me. Some experimental instruments had been 
built for higher frequencies but each design was for 
a very limited band. The aim of this development was 
to provide in a single equipment, or in not more than 
two equipments, the means for making precise field- 
strength measurements in the range from 300 to 3,000 
me. The equipment had to be as simple as possible 
for field operation considering the precision with 
which the measurements were to be made. 

The problem was complicated by the fact that at 
that time there was no receiver or signal generator 
available for operation across the desired band; in 
fact, except for some experimental types of radar re¬ 
ceivers which are single-frequency devices, there were 
no receivers at all available for frequencies above a few 
hundred megacycles. Another serious difficulty was 
that none of the conventional vacuum tubes would 
function either as oscillators, amplifiers, or detectors 
in this range. Tube development, however, was being 
pushed very hard at the time and some experimental 
tubes which would perform the required service be¬ 
came available. The development of silicon and iron- 
pyrite detectors assisted materially by supplementing 
tubes in some of the circuits. 

Under the project, two sets of equipment were de¬ 
signed, built and tested in the field with satisfactory 
results. A few of the 300- to 1,000-mc instruments 
were factory-built and proved valuable in testing 
radar receivers, in addition to their primary job of 

“Projects C-5 and C-5a, Contracts NDCrc-141 andOEMsr-289. 
General Radio Co. 


measuring field strength. Figure 1 shows a complete 
300- to 1,000-mc system including the signal gener¬ 
ator, receiver, and the associated power supplies. The 
1,000- to 3,000-mc units demonstrated in March 1942 
were adjudged too complicated for regular production 
at the time. 

192 DEVELOPMENT PROBLEMS, 

300 TO 1,000 MC 

The major problem in the design of the lower- 
frequency model was the development of an oscillator 
that would cover continuously the wide frequency 
range between 300 and 1,000 me in a single band. 
There were two aspects of this problem, choice of a 
suitable tube and design of a suitable continuously 
variable tuned circuit. 

19,21 Tuned Circuit 

An improved version of a tuned circuit previously 
described and used in a wavemeter covering the 
range of 50 to 400 me was employed in the signal 
generator. It consisted of an annular stator assembly 
carrying two sets of 90-degree stator plates mounted 
diametrically opposite each other within the ring and 
a rotor carrying fan-shaped plates that mesh with the 
stator plates. That portion of the ring not carrying 
stator plates served as a tuned-circuit inductance. Dis¬ 
placement currents passed from one set of stator plates 
to the other through the rotatable fan-shaped rotor 
plates and the complete plate assembly served as a 
continuously adjustable “series-gap” capacitor. When 
connections were made to two stator-plate assemblies, 
therefore, the developed impedance was that of an 
antiresonant circuit whose natural frequency could be 
varied without the use of sliding contacts by means 
of an angular shaft rotation. 

The mechanical design of a tuned circuit of this 
type, called a butterfly circuit from the appearance 
of the rotor (Figure 2), leads to wide tuning range, 
because the rotor plates act not only as capacitor plates 
but as short-circuited turns, or eddy-current shields, 
in the magnetic field. When the rotor plates are 
turned, they vary both the capacitance and the in- 


C ON FT D E NT T A L 


137 






138 


U-H-F FIELD-INTENSITY MEASURING EQUIPMENT 



Figure 1 . Complete setup of the signal generator and receiver with sources of power supply. 

ductance. The eddy-current shielding is most effective are, therefore, both minimum. Conversely they are 
when the rotor plates are completely out of mesh with both maximum when the rotor plates are fully meshed 
the stator and the effective inductance and capacitance with the stator plates. 


















RECEIVERS 


139 



Figure 2. Butterfly circuit used in 300- to 1,000-mc receiver 


higher frequencies some trouble was encountered in 
obtaining proper operation. 

19-3 1,000- TO 3,000-MC SIGNAL GENERATOR 

For this instrument the 368-A tube was used with 
a single-ended transmission line oscillator circuit in 
which the fundamental frequency range was from 
500 to 1,200 me, with production of strong second 
and third harmonics. Means were provided for cou¬ 
pling the transmission line to the attenuator so that 
the second or third harmonic of the oscillator could 
be selected at will, the second harmonic being em¬ 
ployed to cover the range 1,000 to 2,400 me and the 
third harmonic for the range 2,400 to 3,000 me. 


19-2-2 Tube Choice 

At the time, there were few tubes which would work 
at all much above 500 me. The particular tuned cir¬ 
cuit developed further restricted the tube types that 
could be employed, the choice rapidly narrowing down 
to the Western Electric 368-A which was then in 
commercial production. Western Electric Type D- 
160127 (1221-Y) coaxial-line tube and General Elec¬ 
tric Types ZP-423 and ZP-446 lighthouse tubes were 
not satisfactory for the butterfly-tuned circuit because 
of the distribution of interelectrode capacitances, al¬ 
though their ultimate frequency limits with other 
types of circuits are much higher than the 368-A tube. 

192 3 Output System 

Because a balanced output was desirable and be¬ 
cause a very simple design could be evolved, the 
mutual-inductance type of attenuator was adopted. 
To give maximum accuracy in standardizing the 
attenuator output, balanced iron-pyrite crystal recti¬ 
fiers were mounted immediately at the terminals of 
a probe in which the output cable terminated. The 
coupling from the oscillator to the attenuator was 
obtained by an open-wire pickup lead of variable 
length that coupled capacitively to the oscillator and 
that served as the source of magnetic field within the 
attenuator. 

19.2.4 Performance of Butterfly Circuit 

The butterfly circuit was found to perform satis¬ 
factorily over the range of 300 to 1,000 me, but at 


Figure 3. Details of flexible antenna made of steel tape 
which is extended to correct length. 

19.4 RECEIVERS 

In the design of the lower-frequency receiver, the 
oscillator frequency chosen was half the signal-gener¬ 
ator frequency. This choice was made to save power 



CONFIDENTIAL 



















140 


U-H-F FIELD-INTENSITY MEASURING EQUIPMENT 


and expense, since the Type 955 acorn tube could be 
used instead of the 368-A. Sufficient power for a het¬ 
erodyne voltage at the second harmonic could be ob¬ 
tained without difficulty. Antenna tuning was effected 
by a butterfly circuit. An iron-pyrite crystal acted as 
a mixer and was connected directly across the antenna¬ 
tuning circuit. The five-stage i-f amplifier with cath¬ 
ode-resistor stabilization of input admittance pro¬ 
vided a 2-mc band width with center frequency of 
30 me. An input signal of 10 fi\ could be distin¬ 


guished through the background noise produced by the 
crystal mixer and the first i-f stage. 

Design of the 1,000- to 3,000-mc receiver (Figure 
4) followed the design of the 300- to 1,000-mc instru¬ 
ment. The oscillator covered the range 500 to 1,000 
me, the antenna-tuning butterfly was scaled down 
three to one, the second harmonic of the oscillator 
beating with the incoming signal from 1,000 to 2,000 
me and the third harmonic on the range of 1,500 to 
3,000 me. 



Figure 4. The high-frequency (1,000- to 3,000-mc) receiver. 


CONFIDENTIAL 







Chapter 20 

AIRCRAFT FACSIMILE SYSTEM 


Development of lightweight, compact, high-speed scanner 
and recorder to be used to test the efficiency of facsimile com¬ 
munication between a military airplane and ground. Syn¬ 
chronization was effected by means of transmitted signals 
instead of by the customary method of using tuning forks. 
Transmission rate of 48 sq in. per minute with copy 8 in. 
wide was effected. 

20.1 STATE OF THE ART 

A t the time this project 8 was started early in 1942, 
- there were no facsimile systems operating faster 
than about 20 sq in. per minute. All the scanners were 
of the drum type in which each successive message had 
to be fastened on the drum and the machine started. 
Most of the recorders were also of the drum type, 
using photographic recording or a stylus on paper of 
the Teledeltos type. Some recorders using continuous 
carbon paper supplied from a roll were available, but 
their speed was limited by the mechanical movement 
of the printers to not more than 12 sq in. per minute. 
Also available were recorders using a continuous roll 
feed supplying electrolytic paper, but again the speed 
was low. 

Synchronizing methods in use on low-speed systems 
depended on fork frequency standards at the two ends 
of the circuit, the forks driving synchronous motors 
which in turn rotated the drums. Some work had been 
done in transmitting a synchronizing signal along 
with the picture over a radio circuit but circuits using 
this principle were not in operation. Little or no ex¬ 
perience had been had with synchronization of ma¬ 
chines running at drum speeds greater than 200 to 
300 rpm. 

20.2 accomplishments of the project 

Under Project C-8 much progress was made on 
mechanical designs of scanner and recorder, in the 
chemistry of recording solutions for higher speeds, 
and in synchronizing methods. In the scanner com¬ 
pleted under the contract, the optical system rotated 
and traveled along inside a stationary semicylindrical 
piece of clear plastic. The message could be easily 
loaded by simply raising the cover. With scanner am¬ 


plifier incorporated in the same case the weight of 
the whole unit was 31 lb. 

At the proposed speed of 48 sq in. per minute only 
a continuous damp-paper electrolytic recorder was 
feasible. Furthermore the sensitizing solutions which 
were satisfactory at the lower speeds would not re¬ 
spond at the higher speeds and new solutions were 
developed. By using a pretreated roll of damp paper, 
the mechanical construction of the recorder was made 
very simple and compact, and an arrangement was 
found which would retain the damp roll in satisfactory 
condition for a week or more and would still permit al¬ 
most immediate starting of the machine when desired. 

Because the pretreated damp rolls had limited 
shelf life, a portable paper-treating machine was de¬ 
veloped which represented considerable mechanical 
progress over earlier, bulky laboratory equipment of 
this nature. 

Under the project, much time was spent on solving 
the synchronizing problem, which was enhanced by 
the high speed (an increase from 200 to 600 rpm) at 
which this equipment was to operate. Three distinct 
synchronizing systems were developed. 

The equipment was field-tested by the Signal Corps 
General Development Laboratory at Fort Monmouth 
where the scanner was placed in a moving vehicle and 
messages were sent back to a fixed receiving station. 
Except for some jitters caused by gearing irregular¬ 
ities the system worked satisfactorily at a speed sev¬ 
eral times faster than that of previously existing 
equipment. 

About the time the unit was field-tested, the think¬ 
ing of the military people changed and what was now 
desired was a combined scanner-recorder with tuning- 
fork synchronization instead of by transmission of a 
reference over the circuit. The mechanical design of 
the compact scanner developed under Project C-8 
did not lend itself to direct coupling with a recorder. 
Field tests, therefore, did not include plane-to-ground 
transmission. 

Thus the original intent of the project, to develop 
a compact high-speed facsimile system to be tested 
between airplane and ground as a means of indicating 
the military value of such communication, was only 
partially carried out due to change of military plans. 


Project C-8, Contract NDCrc-88, RCA Manufacturing Co. 


CONFIDENTIAL 


141 





142 


AIRCRAFT FACSIMILE SYSTEM 


20 3 SYNCHRONIZING METHODS 
DEVELOPED 

Since tuning-fork synchronizing apparatus devel¬ 
oped before World War I was heavy and required 
considerable power (25 lb, 24 watts) other means of 
synchronization were sought. Furthermore, if forks 
were used at scanner and recorder, automatic phasing 
would be necessary, since hand phasing would be too 
slow at the operating speed desired and would require 
constant attention. Thus special phasing signals would 
be necessary. Synchronizing and phasing become two 
problems, therefore, and merge into a single problem 
only in the limiting case where the frame or phase 
signal also controls recorder speed. 

Where synchronism is maintained by a transmitted 
signal it is still important that the scanner speed be 
essentially constant. Some slow variation can be al¬ 
lowed, but abrupt change will result in a jog in the 
copy. To be effective, phasing and “sync” signals 
must be positive in action; that is, no change in 
synchronization or phasing should occur except as 
such change is initiated by the transmitted signals. 
In this manner, fading or drop-out of signal will not 
cause lack of synchronization or phase. This requires 
that the two signals be separated far enough from the 
picture signals so they can be separated by filters. 

Assuming that the sync tone is received without dis¬ 
tortion, the problem remains of controlling the re¬ 
corder motor speed from this tone. At the speeds of 
transmission required, the sync tone should be 300 
cycles or higher and this would require a 300-cycle 
synchronous motor, a product not available at the 
time of the project. It would be more desirable to use 
a standard 60-cycle motor, provided means could be 
found to limit the phase shift with changes in voltage 
and load. 

The research on synchronizing methods, therefore, 
was devoted to means for controlling a 60-cycle motor 
from various types of transmitted signals. 

20 3 1 Synchronizing Method No. 1 

In the first method investigated, sync and phasing 
were transmitted as separate signals, the former being 
a steady tone of 960 cycles generated by a tone wheel 
on the scanner motor and the second a 4-kc signal at 
the start of each scanning line. At the recorder, the 
two signals were separated by filters. The incoming 
960-cycle signal was used to lock in a 960-cycle phase- 
shift oscillator, which then provided a reference tone 


representing the scanner speed. It was stable enough to 
hold approximate control even when no 960-eycle tone 
was being received from the scanner. A second 960- 
cycle oscillator was locked in on a tone wheel on the 
recorder shaft and provided the reference tone repre¬ 
senting the recorder speed. These two tones were 
combined and rectified, the resulting current being 
zero when the phase difference was 180 degrees. 
Maximum current occurred when the two systems 
were in phase. This direct current could be employed 
to control the recorder motor speed in the following 
manner. 

The 60-cycle vibrator driving the recorder motor 
was self-excited at approximately the correct frequen¬ 
cy, but by varying the coil excitation voltage the fre¬ 
quency could be changed over a range of ±2 cycles. 
The rectified beat between the reference tones con¬ 
trolled a load tube across the vibrator excitation and 
therefore controlled the vibrator frequency. Normal 
setting was made with the two tones exactly 90 de¬ 
grees apart (center of the range) and the vibrator 
at exactly 60 cycles. If the recorder speeded up slight¬ 
ly, the phase shift increased and excitation of the 
vibrator was reduced by the increased loading so that 
the vibrator frequency was reduced. 

This method was capable of holding the phase dis¬ 
placement at ±6.53 degrees of the 60-cycle supply. 

Line phasing was accomplished by pulsing the load 
tube to the full slow position and beyond control of 
the phase corrector each time a 4-kc signal was re¬ 
ceived. When the recorder was jogged out of sync a 
sufficient number of times to come into correct phase 
position, a commutator on the recorder shaft shorted 
out any further action of the phasing pulses. 

This system had an important ability to work with 
poor received signals. The filtering action of the 960- 
cycle RC oscillators was so good that no other filter 
was required to separate the 960-cycle tone from the 
other signals. Phasing was fast and accurate, only a 
few scanning lines being lost at the start of a message. 

The method, however, proved to be too precise; 
hunting easily developed. To remedy this trouble, 
tightness of control was halved by dropping the sync 
tone to 480 cycles. This improved the action but the 
system was still too sensitive. 

20 . 3.2 Synchronizing Method No. 2 

In this method the sync reference was taken from 
the phasing signal once per scanned line. This meth- 


CONFTDENTIAL 





SYNCHRONIZING METHODS DEVELOPED 


143 


od would require no special sync frequency in the 
transmitted signal, filtering in the recorder would be 
simplified, and the full modulation capacity of the 
radio transmitter could be used for the picture signal. 

In this method the vibrator was tube-driven instead 
of being self-excited, the driver tube receiving its ex¬ 
citation from a 60-cycle oscillator of the EC phase- 
shift type and adjustable over a small frequency 
range. The phase signal of one pulse per scanning 
line was separated from the picture signals by a lim¬ 
iter-filter arrangement, rectified and shaped to ap¬ 
proximate a half cycle of a 60-cycle wave. This shaped 
pulse was fed to the 60-cycle oscillator and locked it 
into synchronism on the nearest half cycle. Accurate 
synchronism was held by this single locking pulse 
once per scanning line (every 6 cycles). 

Phasing was accomplished by having the phase pulse 
change the oscillator frequency to 58 cycles instead of 
60 whenever the recorder was out of phase. This was 
done by changing the network timing by commutator, 
so that the frequency change was effected instead of 
pulsing the EC oscillator. This caused a rapid drift 
of the recorder to a new phase position at which point 
the oscillator was immediately restored to 60 cycles 
and pulsed into synchronism. 

In normal operation this system was very accurate 
and phasing was even more rapid than in the first 
system. However, two commutators were required on 
the recorder and because of gradual changes due to 
brush wear, etc., they did not maintain their proper 


relation to each other over a long period. This caused 
jitter. Furthermore, the 60-cycle EC oscillator could 
shift in frequency more rapidly than the vibrator and 
motor could follow. The result was that heavy surge 
currents would pit and burn the contacts so that the 
vibrator was practically worn out after a few hundred 
hours of operation. 

20 . 3.3 Synchronizing Method No. 3 

A modification of the second system was finally 
adopted. In this method, the EC oscillator driving the 
vibrator was replaced with an LC oscillator which 
could not change its frequency so rapidly as the vi¬ 
brator and much better control was experienced. The 
vibrator had a normal life in this system. The control 
system for correcting the oscillator frequency was 
changed to a self-balancing bridge instead of depend¬ 
ing on pulsing to a new frequency with each successive 
scanning line. With this bridge no action took place 
if no phasing signal was received, the oscillator there¬ 
by holding the frequency to the value to which it was 
last set. 

Eapid drift on fade-out of the signal was thereby 
prevented. Only one commutator was required and 
this did not have to be phased in with the 60-cycle 
motor. Brush wear, therefore, did not disturb syn¬ 
chronism. A complete description of this system and 
its wiring diagram are to be found in the final 
report 1 of the project. 


I CONFIDENTIAL 








Chapter 21 


ULTRA-HIGH-SPEED FLASH TELEGRAPHY 


Magnetic tape driven at slow speed has impressed upon it 
a 20-word code message recorded at a rate of 30 words per 
minute; this 40-second message is then transmitted by run¬ 
ning the tape at 100 times the recording rate. At the receiving 
end of the circuit the message is recorded at high speed and 
transcribed at low speed. Transmission time is 0.4 second or 
less. Transmission speeds as high as 3,000 to 9,000 words per 
minute are possible. The essentials of the completed units and 
the requirements for the radio circuits are included in this 
summary. The final report, 1 from which this summary is 
condensed, gives more details of solutions to difficulties that 
occurred in development, also numerous oscillographs of test 
messages sent over the system. 

six INTRODUCTION 

T he objectives of this project 3 were to simplify 
code sending and receiving equipment, which usu¬ 
ally consists of perforated tape transmitters and os¬ 
cillographic or other recorders, and to achieve tele¬ 
graphic speeds of the order of 3,000 words per minute. 
Flash telegraphy as a means of radio communication 
appears to have very definite advantages when secur¬ 
ity from direction-finding determination and inter¬ 
ception is desirable. By use of special limiting-type 
amplifiers and modulators, discrimination against 
static and multiple-path transmission was obtained. 

Any method of flash telegraphy requires wider r-f 
bands and imposes stricter requirements on signal- 
to-noise ratios and selective fading than does trans¬ 
mission and reception at manual speeds. Although the 
Services did not adopt the flash system, largely be¬ 
cause of limitations in the then existing radio equip¬ 
ment, the limitations could have been reduced sub¬ 
stantially by engineering a complete flasher system, 
including not only the terminal units but radio trans¬ 
mitter and receiver equipment designed to meet the 
special requirements of flash telegraphy. 

The achievement of a speed-up factor of some 100 
to 1 and telegraphic speeds of 3,000 words per minute 
seems to be unique. The need for maintaining radio 
silence during naval operations and the great desira¬ 
bility of security in the field of combat indicate that 
the terminal equipment and methods described here 
offer distinct answers to these security problems. 


“Project C-28, Contract OEMsr-50, Bell Telephone Labora¬ 
tories, Inc., Western Electric Co., Inc. 


21.2 FUNCTIONAL OPERATIONS OF 
THE FLASHER 

The method of magnetic-tape recording employed 
involves essentially the transverse magnetization of a 
thin ribbon of magnetic tape. Ordinarily, one of two 
recording procedures may be used. 

D-C Erase. In this method the tape is erased by a 
d-c flux sufficient to saturate the tape. Hence, as the 
tape leaves the erasing pole piece it is magnetized in 
one direction. In recording, the signal or a-c flux is 
superimposed on a d-c bias flux having a sign opposite 
that of the d-c erasing flux. This is done in such a way 
that the signal flux increases or decreases the bias 
flux over a linear range of magnetization. As the tape 
leaves the recording pole piece, it is magnetized in 
one direction in varying amounts, in accordance with 
the signal flux variations. 

A-C Erase. In this method, the tape is erased by an 
alternating flux having a frequency that is high com¬ 
pared with any signal frequency and a wavelength on 
the tape that is small compared with the pole piece 
dimensions. As the tape leaves the pole piece, it is 
completely demagnetized. In recording, the signal 
flux is superimposed on a similar high-frequency bias 
flux. This is done in such a way that the magnetiza¬ 
tion passes linearly from positive to negative values 
in accordance with the signal flux. This second method 
gives a signal-to-noise ratio some 10 db better than 
that obtained with d-c erase and bias, but requires the 
use of a high-frequency oscillator. Direct-current bias 
may also be employed in the case where the erasing 
is done with high frequency. This gives essentially the 
same magnetization pattern for a given signal, as is 
obtained with d-c erase and d-c bias. 

213 RECORDING CONSIDERATIONS 

In the preliminary studies, two methods of record¬ 
ing the telegraph signals were considered. One method 
involved the recording of the d-c telegraph signals 
directly, which consist of current intervals correspond¬ 
ing to the dot and dash marks and no-current inter¬ 
vals corresponding to the spaces. The other involved 
recording an a-c carrier wave as modulated by the 


144 


\ CONFIDENTIAL 





RECORDING CONSIDERATIONS 


145 


d-c telegraph wave, the modulator being of the bal¬ 
anced type so that the modulating or d-c telegraph 
wave is balanced out. In this case, the signals to be 
recorded consist of intervals of alternating current 
corresponding to dot and dash marks and intervals of 
no current corresponding to the spaces. For low-speed 
recording, such a signal is obtained by simply keying 
an alternating rather than a direct current. 

D-C Recording 

At a telegraph word rate of 30 words per minute, 
the dot space interval corresponds to the period of a 
12.5-cycle wave. A sequence of d-c dots and spaces may 
be represented as a wave composed of a d-c component, 
a component of 12.5 cycles and odd harmonic compo¬ 
nents of this frequency having amplitudes diminish¬ 
ing inversely as the number of the component. Ordi¬ 
narily, three of the components are sufficient to define 
the waveform with enough accuracy for telegraph 
purposes. Hence a frequency range from 0 to some 
37.5 cycles is needed for low-speed recording and re¬ 
producing. 

Recording the d-c telegraph pulses presents no 
problem and may be done by employing d-c erase and 
d-c bias so arranged that the d-c telegraph pulses op¬ 
pose the bias. Thus, the magnetized elements of the 
tape will correspond to current intervals of the signal. 
The voltage developed by the reproducer, however, de¬ 
pends on the derivative of the tape magnetization, so 
that significant voltage is obtained only at the begin¬ 
ning and ending of the original current intervals. Re¬ 
produced d-c intervals may be obtained by employing 
a pair of thyratrons or trigger tubes so arranged that 
voltage of one sign strikes an arc in one tube and ex¬ 
tinguishes an arc in the second and voltage of opposite 
sign strikes the second tube and extinguishes the first. 

This method of recording and reproducing was tried 
in the preliminary studies and found to be feasible. It 
was felt, however, that the method would be particu¬ 
larly vulnerable to extraneous transient voltages strik¬ 
ing arcs in the tubes in random fashion, and the meth¬ 
od was abandoned in favor of the one involving the 
recording of a-c pulses. As the work has progressed, 
this decision has been questioned, and it is believed 
now that the first method might well be investigated 
further, particularly if higher speeds are of interest. 
The d-c method has the advantage of requiring about 
half the frequency range required for the a-c method, 
as will be seen in the following paragraphs. 


21-31 A-C Recording 

When a-c telegraph pulses are to be recorded, the 
low tape speed is set by the frequency of the a-c wave 
needed to define a dot interval. At a 30-words-per- 
minute rate, a frequency of 60 cycles was considered 
the lowest that could be used. This corresponds to 
about half of the lowest frequency used in commercial 
carrier telegraphy. The dot interval of 0.04 second is 
represented by about 2 V 2 cycles of the carrier wave. 

A sequence of 60-cycle dots and spaces may be rep¬ 
resented as a wave composed of a 60-cycle component 
and odd order summation and difference components 
having frequencies of 72.5, 47.5, 97.5, 22.5, etc., cycles, 
of amplitudes diminishing inversely as the orders of 
the components. Ordinarily, three components are suf¬ 
ficient to define the envelope, so that a frequency 
range from 22.5 to 97.5 cycles or a band width of 
some 75 cycles is needed. It was found that a tape 
speed of V 2 in. per second afforded, with a little equal¬ 
ization, a good response over the frequency range from 
25 to some 200 cycles, and this speed was adopted 
for low-speed recording. 

As the tape speed is increased, the frequency range 
increases. A high tape speed of 50 in. per second was 
chosen, which affords a good response up to frequen¬ 
cies of about 10,000 cycles and results in a speed-up 
factor of 100 to 1. At this speed, the signal frequencies 
cover the range from 2,250 to 9,750 cycles, or a band 
width of 7,500 cycles. The 60-cycle carrier frequency 
appears as 6,000 cycles. 

21.3.2 Teletypewriter and Perforated 
Tape Methods 

During the course of the preliminary studies, con¬ 
sideration was given to several operational procedures 
for the flasher equipment. One of the early procedures 
considered was that of using teletypewriters for the 
low-speed keying and transcribing operations, and 
experimental work was carried out with such equip¬ 
ment. This procedure was abandoned because its use 
appeared to require a higher degree of synchroniza¬ 
tion between sending and receiving flasher units than 
was desirable from a practical standpoint. The use of 
perforated paper tape for keying the sender and a 
Boehme ink recorder for recording the signals repro¬ 
duced by the receiver was considered also. Finally, the 
low-speed word rate of 30 words per minute, making 


CONFIDENTIAL 







146 


ULTRA-HIGH-SPEED FLASH TELEGRAPHY 


manual keying and aural reception possible, was 
adopted. This procedure obviated the need of any close 
synchronization between sender and receiver, although 
at that time a starting pulse was considered necessary 
to receive high-speed Hash messages in a fraction of 
a second. 

21,3 3 Starting-Pulse Problems 

Several methods employing a starting pulse for re¬ 
ceiving high-speed messages were considered. In prin¬ 
ciple, they involved a connection to the recording coil 
through a pair of commutator rings. A brush connect¬ 
ing the rings would he released by the starting pulse 
and would make one revolution only over the commu¬ 
tator, thus permitting a recording to he made during 


one revolution. The time of one revolution would cor¬ 
respond to the high-speed transmission time. Con¬ 
sideration was given also to the use of a coded type 
of starting pulse so that the receiver might not he 
triggered off by a static crash or other random dis¬ 
turbance. 

The reason for proposing the use of a starting pulse 
was to avoid the erasing of the recorded message by 
the recording bias current. Initially, bias current was 
considered to be essential, and to avoid erasing it was 
necessary to disconnect the bias at the end of tiie 
high-speed message. The purpose of a bias current is 
to confine the magnetization to a linear range and 
thus avoid nonlinear distortion of the signal. Subse¬ 
quently, it was realized that many of the distortion 
products of the high-speed signals would fall outside 



Figure 1 . Functional operations of flasher system of high-speed telegraphy. 


1. Recording message for transmitting (low-speed operation). Switch Si is thrown to position 2 and tape erased for at least one revolution. 
Switches Si and S 2 are then thrown to position 1. Cam-operated device dims record lamp momentarily, thus indicating beginning of time interval 
of 46 seconds during which messages may be recorded. 

2. Confirming recorded message (low-speed operation). Switches S 2 and S 3 are thrown to positions 2 and 1, respectively. Recorded message is 
picked up, amplified, and used to modulate or key a 1,000-cycle tone to produce audible signals in telephone receivers. 

3. Transmitting message (high-speed operation). Switches S 2 and S 3 are thrown to position 2. Recorded message is picked up, amplified, and 
sent on line to radio transmitter. Line should be capable of transmitting a band of from 2,250 to 9,750 cycles per second. When transmit key Is 
depressed, cam-operated device is energized and acts to close transmit relay for one revolution only of tape disk. Closing of transmit relay is 
indicated by transmit relay lamp. After lamp flashes, operator may release transmit key. 

4. Receiving message (high-speed operation). With tape erased, switches Si and S 2 are thrown to positions 3 and 1, respectively. Operator 
monitors incoming signals and throws switch S 2 to position 2 after message is received. The 6,000 cycle tone is modulated or keyed by recti¬ 
fied output of radio receiver detector circuit. Amplifier connecting receiver and modulator is provided as separate unit, so that it may be located 
near radio receiver. Line from amplifier to modulator should pass band from 0 to 3,750 cycles. 

5. Transcribing received message. Operation same as 2 above. 


CONFIDE 














































DESIGN OF FLASHER 


147 


the frequency range of the tape, and that a bias cur¬ 
rent would probably not be necessary. Recording with¬ 
out bias was tried and appeared to give satisfactory 
reproduction, so the procedure of employing a start¬ 
ing pulse was abandoned. 

Recording without bias simplified the high-speed 
receiving operation considerably. It is only necessary 
to erase the tape clean and wait for the flash message. 
When it arrives, it is recorded and remains on the 
tape. A manual operation is required to erase it. The 
recording process is made highly discriminatory 
against static, multiple-path transmission, and atten¬ 
uation or fading by employing a sharply limiting 
amplifier. It is only necessary to set the gain ahead 
of the recorder so that static and signals from un¬ 
wanted paths are just below the recording level. The 
wanted signal, if it is only a few db or 40 db above 
this point, will be recorded successfully. On this basis 
the functional operations indicated in Figure 1 were 
adopted. The tape disk consists of a loop of magnetic 
tape mounted on the periphery of a light wheel, and 
having a length sufficient for recording 22 words at 
the rate of 30 words per minute. 


given in Figure 2. The function of these circuits as a 
variable attentuator is as follows: Each series arm 
( R v ) is a stack-up of copper oxide disks having uni¬ 
lateral conductivity. A bias voltage is applied at points 
a and b of such polarity as to make the transmission 
path between 7\ and T 2 a high resistance. The d-c 
signal voltages are applied at a and b with the oppo¬ 
site polarity, making the resistance low. Thus the 
device acts as a variable attenuator whose loss is gov¬ 
erned by the signal voltage opposing the fixed bias. 
The change of level obtained is in excess of 40 db. 
Performance curves are shown in Figures 3 and 4. 
Further tests were also undertaken with regard to the 
response characteristics of the tape at the low and 
high speeds, as existing quantitative data were meager. 

Several features of the circuit design were to fol¬ 
low rather conventional lines, but some presented novel 
problems. Some of these novel features were recording 
on the magnetic tape without bias; a circuit for trans¬ 
mitting the message once only (or during one revolu¬ 
tion of the tape disk ; an amplifier to take the detector 
output of the radio receiver and operate a copper 
oxide modulator at its output; means for keying a 


SIGNAL 

INPUT 




Figure 2. Copper oxide modulator circuit 


21.4 DESIGN OF FLASHER 

Proposed design elements having been decided on, 
the physical design of the equipment as a whole was 
started. At the same time, detailed studies of the per¬ 
formance of certain of the elements were begun. One 
such element given detailed study was the copper 
oxide modulator circuits, the schematic of which is 


1,000-cycle tone to the headphones, corresponding to 
the message recorded on the tape at low speed; means 
for driving the tape wheel at two uniform rates of 
speed in a ratio of about 1/100 and for shifting rap¬ 
idly between these two speeds. One additional feature 
provided was the incorporation of circuit switching 
and gear shifting from high to low speed in one con¬ 
trol. This control has five positions providing two 


CONFIDENTIAL 


I 






























148 


ULTRA-HIGH-SPEED FLASH TELEGRAPHY 



Figure 3. Suppression characteristics of flasher modulator. 


circuit conditions at high speed, neutral, and two 
circuit conditions at low speed. 

21,41 Tape Drive Mechanism 

The high-speed drive consists of a worm on the 
motor shaft and a worm wheel concentric with tape- 
wheel shaft. The low-speed drive consists of a train 
of gears comprising three steps, steps (1) and (2) 
being spiral gears, and step (3) a worm and worm 
wheel concentric with the tape-wheel shaft. All gear¬ 
ing is in constant mesh, and the tape-wheel speed 
change is accomplished by means of a sawtooth clutch 
which couples the tape wheel to either the high- or 
low-speed worm wheels. Some care had to be taken 
to reduce flutter from the gear drive and also, because 
of the low voltage developed by the reproducing pole 
piece at low tape speeds, attention had to be given to 
reducing vibration. 


21,4,2 Amplifier for Use with Radio 
Receiver 

This booster amplifier, Figure 5, is a two-stage, 
high-gain unit of the limiting type. A minimum in¬ 
put voltage of about 0.3 volt (or —10 db per volt) is 
required for full output, but as the input voltage is 
increased beyond this value, little change in the out¬ 
put results. Thus a signal of substantially constant 
amplitude will be furnished to the input of the flasher 
regardless of variation of the level at the radio re¬ 
ceiver output, provided it always exceeds 0.3 volt. In 
this way, the effects of fading in the radio transmission 
circuit are minimized. 

A gain control is provided at the input of the booster 
amplifier to reduce its sensiitvity in order to discrim¬ 
inate against interference. The output tube is biased 
to plate-current cutoff. Under these conditions the 
output current versus input voltage is nonlinear. 


CONFIDENTIAL 




































DESIGN OF FLASHER 


149 



LEVEL ACROSS INPUT OF MODULATOR IN DB (ODB* 1 VOLT) 


Figure 4. Load characteristics of modulator. 


Small input voltages produce only very small plate 
currents, but input voltages of the order of the bias 
voltage produce plate currents which are proportion¬ 
ally much greater. In other words, a 6-db increase in 

VT-I C2 VT-2 



Figure 5. Booster-amplifier limiter circuit. 


input voltage, above some small value, will increase 
the plate current by 15 db or better. Thus, by properly 
adjusting the gain, a signal (5 db above the noise level 
prevailing over the recording period will, by virtue of 
this expansion, give satisfactory reception. 

21.4.3 pi rs t Trials as a Complete System 

After making volume runs and other tests on the 
completed machines, they w r ere interconnected by a 
wire line through a vacuum-tube rectifier of the biased 
type. This was used to simulate the entire radio link, 
including keyer, transmitter, and receiver. The out¬ 
put of this rectifier was connected to the input of the 
booster amplifier. A series of dots was recorded on the 
tape of the sending flasher by means of a commutator 
and test transmissions were made. The signal was ac¬ 
curately received, however, over only a very small 
volume range, about 8 db, rather than over a range of 
30 to 40 db as had been anticipated. Particularly un- 





























































150 


ULTRA-HIGH-SPEED FLASH TELEGRAPHA 7 


accountable was a filling-in of the received signal at 
high input levels. 

2144 Distortion from Tape Recording 

Oscillograms of the reproduction from the tape at 
low speed revealed that if pulses of uniform length of 
time were recorded, nonuniform pulses were repro¬ 
duced. Increase of the pulse length occurred due to 
transients of considerable amplitude, while shortening 
was also observed. This was due to recording with d-c 
erase and no bias when receiving signals at high speed. 

C L 

—c^c—-II- 

-m— 

R 


CONDITION A 


RECORDING 

COIL 


R 2 C L 



RECORDING 

COIL 


Figure 6. Pre-equalizers for magnetic tape recording. 


Under these conditions, the tape is magnetically satu¬ 
rated by the erase current. The received signal then 
can change the magnetization in one direction only. 
In other words, the tape acts as a rectifier in the same 
manner as a vacuum tube biased to cutoff. Under these 
conditions it is possible to incur a considerable short¬ 
ening of the pulse length, depending on the phase of 



10 2 0 4 0 6 0 80 100 2 00 300 

FREQUENCY IN C 


the received signal. For example, if the received signal 
corresponding to a dot happened to consist of 2 V 2 
cycles of the 6,000-cycle carrier, the wave will have 
2 half-cycles in one direction and 3 half-cycles in the 
other. Thus the recorded wave might be shortened 40 
per cent if the initial and final half-cycles had the 
same polarity as the erase current. The remedy for 
this distortion was to employ a-c erase so as to leave 
the tape nonmagnetized. When this was done using 
a 30-kc erase frequency, pulse shortening was elimi¬ 
nated. 

Pulse lengthening due to transients was reduced by 
the following prodecures. Care was taken to insure 
that the locally generated 60- and 6,000-cycle carrier 
waves and the 30-kc erase wave were free from dis¬ 
tortion. Suitable frequency equalization and control 
of the input levels to the recorder effected further 
improvements. 

I 11 particular one change helped reduce the tran¬ 
sients materially. Low-speed keying had been done 
simply by opening and closing the series circuit at the 
input to the recording pre-equalizer. It was found that 
shunting the equalizer input with a resistance and 
breaking ahead of this point made considerable im¬ 
provement (see Figure 6). Condition A shows the 
original circuit and condition B the improved one. 
The frequency response obtained off the tape using 



1000 2000 5000 10000 20000 

FREQUENCY IN CYCLES PER SECOND 


Figure 7. Relative response of magnetic tape at speed Figure 8. Relative response of tape at speed of 50 in. 

of one-half in. per second. per second. 







































































DESIGN OF FLASHER 


151 



Figure 9. Flasher unit complete. 


this pre-equalizer is shown by the lower curve of 
Figure 7. The upper curve gives the response of the 
reproducing amplifier, which was unequalized. 

The frequency response at high speed was not so 
good, relatively, as at the low speed. Post-equalization, 
or, in other words, equalization of the signal repro¬ 
duced from the tape was necessary in addition to the 


pre-equalization used. The response obtained under 
these conditions is shown by the lower curve of Figure 
8, while the upper curve shows the characteristics of 
the equilized amplifier plus a high-pass filter which 
was inserted in the output. This filter was found help- 
ful in discriminating against low-frequency noise of 
a microphonic nature originating in the amplifier and 
caused primarily by the impact of the timing cams 
on the timing cam contact springs, which occurs once 
per revolution of the tape disk. 

With these improvements and modifications, the 
total maximum distortion in the complete process of 
transmission from one terminal to the other, includ¬ 
ing final low-speed reproduction, was reduced from a 
possible 80 per cent to a possible 25 per cent. Satis¬ 
factory transmission was now obtained over a range of 
about 35 db at the input to the booster amplifier. It 
was found that the output-level adjustment of the 
sending machine was somewhat critical, as had been 























152 


ULTRA-HIGH-SPEED FLASH TELEGRAPHY 



Figure 11. Close-up of magnetic tape recording and 

reproducing unit. 

expected. The correct adjustment is one at which the 
transients, which are still present to some extent, 
will not be high enough in level to pass through the 
rectifier used to simulate the radio link. Thus the rec¬ 
tified output will equal the true pulse length. If the 
transient is added to the true rectified pulse, consid¬ 
erable time distortion results and may cause oblitera¬ 
tion of a space. The same considerations apply to the 
adjustment of input level to the radio transmitter 
keying equipment. Since the level off the tape is sub¬ 
stantially constant, a correct gain adjustment, once 
made, may be expected to remain correct. 

21-5 APPARATUS DETAILS 

A photograph of the flasher unit complete is shown 
in Figure 9. A view with the cover removed showing 
the arrangement of parts is shown in Figure 10. 

2151 Magnetic Tape U nit 

One unit performs the function of recording, re¬ 
producing and erasing. This unit is shown in Figure 
11. It consists of two pole pieces which bear on oppo¬ 
site sides of the magnetic tape. The coil is mounted 
over one pole piece. To obtain uniform magnetization 
of the tape at various frequencies, constant current 
should be supplied to the coil. When used as a repro¬ 
ducer, the coil must be connected to an impedance 
which will remain higher than its own impedance 
over the frequency range of interest in order to ob¬ 
tain substantially open-circuit voltage. 

2152 Tape-Recording System 

The tape-recording system is actually composed of 
two independent recording circuits, (1) a low-speed 
recording circuit used to record the outgoing message 
on the tape, and (2) a high-speed recording circuit 


used to record the incoming message on the tape. 
These circuits are best discussed separately with ref¬ 
erence to the schematic, Figure 12. 

The frequency recorded at low speed is 60 cycles, 
supplied by one-half the 6.3-volt winding of the power 
transformer. This voltage is only applied when the 
flasher is in the low-speed dial positions LS and LB. 
It is also necessary that the low-speed timing cam 
contact r l\ be in a closed condition. The 60-cycle tone 
is keyed by the telegraph operator by means of the 
telegraph key. A pre-equalizing network C 3 , L x , R 2 
is used to improve the frequency response character¬ 
istic of the tape unit. From the equalizer the keyed 
voltage is applied to the tape unit through switch S 51 
which must be on position 1 (dial position LS). It is 
well to note at this point that the 60-cycle tone is 
applied to the magnetic-tape unit only on dial posi¬ 
tion LS. On position LB the 60-cycle tone is applied 
only for the purpose of operating the low-speed tim¬ 
ing light. 

A d-c bias is provided for low-speed recording by 
the relay and bias-voltage supply PT 2 and VB X . When 
the flasher is on dial position LS the rectified d-c volt¬ 
age is applied through switch S 33 to the recording bias- 
supply circuit, which adjusts the bias current to 1 ma 
and filters the ripple due to rectification. 

Monitoring during the process of low-speed record¬ 
ing is provided for. Part of the recording voltage is 
applied through the bleeder 7? 46 , T 47 to the ampli¬ 
fier input. 

At high speed the signal to be recorded consists of 
the pulses appearing on the incoming line. When the 
flasher is on dial position IIB, these pulses are applied 
through switch S 53 to the control circuit of the 
modulating or keying varistor VR 3 , thereby keying 
the 6,000-cycle output of the signal oscillator. This 
is of the electron-coupled type using a 6V6 tube 
(FT-4). The keyed 6,000-cycle pulses pass through 
the pre-equalizing network /? 36 C 2S to switch S 51 , 
which applies the signal to the tape unit when the 
dial is on position JIB. No recording bias is applied 
to the tape unit during high-speed recording. The 
receiving operator can monitor the incoming pulses 
by listening on the phones at the output of the 
modulator. 

21.5.3 Tape Reproducing System 

In picking up the signal which has been magneti¬ 
cally recorded on the tape, the voltage level generated 
is directly proportional to the tape speed. Thus in 


I 


CONFIDENTIAL 










(10 AC _PY 
CC 1 “Lh 


filament and plate voltage Supply 


HIGH FREQUENCY 
PRE-EQUALIZER 


REC.VOL. 
CONT. 


PHONE JACKS 



RECORDING AND 
REPRODUCING COIL 



PHONES 


LINE OUT 


CC 2 


R 27 



LINE IN 


CC 3 


S 51-54-FUNCTION CONTROL SWITCH 

POSITION 


t. 

2 . 

3. 

4. 

5 . 


dial 

LS 

LR 

N 

HS 

HR 


FUNCTION 
LOW SPEED SEND 
LOW SPEED RECEIVE 
NEUTRAL 

high SPEED SEND 
high speed RECEIVf 


Figure 12. Flasher unit schematic. 





























































































































































































































































































































































































SUMMARY 


153 


changing the speed by a factor of 100/1 the level is 
increased by 40 db. Therefore, an appropriate ampli¬ 
fier-gain change has been incorporated in switching 
between low and high speeds. This operation is done 
by switch S 51 . An input transformer (T 3 ) is used 
in reproducing at low speed when the highest level 
obtained from the tape unit is approximately TO db 
below 1 volt. At high speed the unit is connected to a 
pad, which also serves to further equalize the high- 
frequency response, and thence directly to the grid 
of VT- 1. The amplifier consists of three stages having 
a voltage gain of approximately 100 db in the high- 
gain condition and 55 db in the low-gain. 

In the low-speed condition used in receiving a mes¬ 
sage on the headphones, the output of the amplifier 
is rectified by the bridge-type rectifier VR 2 . This 
rectified voltage controls the modulator VR Z passing 
the tone from the signal oscillator to the headphones. 
The operation of the modulator is briefly as follows. 
Each arm of VR 3 is a stack-up of copper oxide disks 
having unilateral conductivity. A bias voltage devel¬ 
oped across R 25 is supplied in the nonconducting di¬ 
rection, making the transmission path between T 1 and 
T 2 a high resistance. The rectified signal voltages 
are applied in the conducting direction, making the 
resistance low. Thus VR 3 acts as a variable attenu¬ 
ator whose loss is governed by the signal voltage 
opposing the fixed bias. The change of level obtained 
is in excess of 40 db. The frequency of the tone gen¬ 
erated by the signal oscillator is approximately 1,200 
cycles. A simple high-pass filter is included in the 
signal circuit to discriminate against 60-cycle ripple 
introduced in the modulator circuit. Volume control 
is provided by P x . 

At high speed the output of the amplifier is switched 
through a 2,000-cycle high-pass filter to discriminate 
against low-frequency noise. The signal frequencies of 
interest are 2,250 to 9,750 cycles. The line then goes 
through the timing relay to the line volume-control 
attenuator VC X and thence through the output line 
to the radio-transmitter keying circuit. 

21.6 REQUIREMENTS ON RADIO CIRCUITS 

Certain requirements are imposed on the radio cir¬ 
cuits to employ successfully flash telegraphy. Most 
of these have been discussed previously, but because 
of their importance they are summarized here. 

1. The radio transmitter must be equipped with 
electronic or other keying means capable of operation 
at the high keying speeds required by the flasher. 


2. The tuning of the transmitter and antenna must 
be sufficiently broad to radiate side-band energy over 
a 7,500-cycle band, 3,750 cycles each side of the 
carrier. 

3. The receiver must have broad enough tuning to 
receive a band width of from 5,000 to 7,500 cycles. 
Excellent low-frequency response is essential. For this 
reason it is desirable to take the output from the 
detector rather than after one or more stages of audio¬ 
frequency amplification. 

4. Signal-to-noise ratio requirements are such that 
a signal having an average value of 6 db greater than 
the average noise level over a receiving time interval 
(0.4 second) will give satisfactory results. The in¬ 
stantaneous noise level during this period should not 
be greatly in excess of the average value. 

5. Amplitude-fading and path-difference require¬ 
ments are such that fading of less than 30 to 35 db 
and path differences of less than 50 to 100 n sec will 
result in satisfactory reception. 

217 SUMMARY 

1. Except in minor respects, the magnetic-tape 
flasher units discussed herein afford practical termi¬ 
nal equipment for achieving flash telegraphy at the 
high rate of 3,000 words per minute. If still higher 
speeds are of interest, it is believed that speeds up to 
9,000 words per minute may be achieved by recording 
and reproducing d-c rather than a-c pulses, as is now 
done. 

2. Recording at high speed without the use of a 
bias current permits a received flash message to be 
retained until erased by a manual operation and obvi¬ 
ates the need, from a recording standpoint, of a 
starting or timing pulse. 

3. The employment of an expanding limiting type 
of booster amplifier and associated modulator for re¬ 
ceiving flash messages, provides considerable discrim¬ 
ination against static, multiple-path transmission, 
and fading. 

4. To make flash telegraphy a dependable means 
of radio communication, affording good security 
against direction-finding determinations and intercep¬ 
tion, it is desirable to engineer a complete terminal 
unit, a radio transmitter-receiver system employing 
suitable radio frequencies, types of directive antennas, 
types of limiting amplification, and operating tech¬ 
niques. The dependability may be further increased 
by designing radio transmitters to flash large powers 
for short time intervals. 


CONFIDENT!. 





Chapter 22 

FREQUENCY-STABILIZED MASTER OSCILLATOR 


A study of instability as caused by the oscillator tube and 
the effect of circuit components; an investigation of the possi¬ 
bilities of multitube oscillators; development of the triode 
single-tube oscillator and buffer. 

22.1 INTRODUCTION 

at the time this project" was started, a tank 
-/i- transmitter required 125 crystals to cover all the 
possible frequencies to be used and, at the time, there 
was considerable doubt whether enough crystals could 
be produced in time for use in the war. It was thought 
that a small master oscillator could be designed to 
cover the frequencies from 2 to 20 me with sufficient 
stability to meet the requirements of the Services. 

Under the project, an oscillator followed by a buffer 
amplifier was developed which was small and had 
good frequency stability. This oscillator consisted of 
a Hartley circuit using a triode (6C4) followed by a 
6AG7 buffer tuned to twice the oscillator frequency 
which delivered to a load consisting of 8,200 ohms 
and 20 nfii capacitance from 65 to 130 volts or 0.5 
to 2.0 watts when powered from a 6- to 8-volt source. 
The stability is indicated by Table 1. 

22.2 PRELIMINARY RESEARCH 

Before development of an actual oscillator was 
undertaken, research was instituted into the share con¬ 
tributed to frequency instability by the tube itself as 
distinct from circuit components. This was followed 

“Project C-59, Contract OEMsr-690, RCA Laboratories. 


by an investigation of the comparative advantages of 
one-tube oscillators and multitube circuits. 

The tube investigation showed that frequency drift 
introduced by the tubes during the warm-up period 
was produced largely by inadequate and changing 
cathode emission. Other factors which contribute to 
frequency drift are mechanical expansion, distortion 
or buckling of the tube elements, and secondary emis¬ 
sion from the bulb and mica parts. 

22 21 Tube Studies 

Effect of the tube on oscillator drift was studied in 
the following steps: 

1. Self-ballasting filament. A tube with a single 
0.008-in. carbon filament was constructed to investi¬ 
gate the usefulness of the negative temperature of 
carbon in combination with external resistance to 
provide ballast action. The effect was too small to 
be useful. 

2. Anode expansion tubes with anodes of molyb¬ 
denum, nickel, and copper to explore the effect of ex¬ 
pansion on grid-plate capacitance. The low plate dis¬ 
sipation caused so little heating that the differences 
among the three sets of tubes was inappreciable. 

3. Secondary emission. Secondary emission, prob¬ 
ably from bulb and mica insulators, caused a warbling 
frequency shift. Carbonizing the bulb helped to reduce 
the frequency change, but lowering the anode voltage 
to 100 or 125 or lower was recommended for master 
oscillator tubes where frequency drift was to be kept 
to a minimum. 


Table 1 . Summary of stability tests. 


Test 

2 megacycles 

Per cent stability 

10 megacycles 

20 megacycles 


Avg 

Max 

Avg 

Max 

Avg 

Max 

Reset. 

0.0004 

0.0011 

0.00037 

0.00065 

0.00026 

0.00045 

Backlash. 

0.00019 

0.00035 

0.00017 

0.00055 

0.00039 

0.00007 

Tube change—oscillator 
Tube change—buffer. .. 
Humidity. 

0.006 


0.0175 


0.02 


0.00068 


0.0042 


0.0069 


0.0019 

0.0025 

0.0069 

Temperature. 


0.00078 

0.0012 


±5% line voltage 

1 minute. 


0.0065 

0.0026 


0.0037 

5 minutes. 


0.0058 


0.0044 


0.0043 

Drift 




5 minutes. 


0.0024 


0.007 


0.006 

0.006 

2 hours. 


0.0017 


0.017 







154 
























































PRELIMINARY RESEARCH 


155 


4. Oscillator-coupling tube. A combined oscillator- 
buffer tube was studied but the decision to investigate 
possible improvement of the 6C4 was made rather 
than to pursue the combination tube further. 

5. Mica-spacer slippage. Sudden discontinuous 
changes in frequency were traced to slippage of mica 
insulators. Polishing or reaming the holes in the mica 
helped the situation. 

6. Advantage of generous emission. The warm-up 
time of a tube depends upon the cathode temperature. 
A heavier heater decreased the warm-up period but 
had no effect on the long-time frequency variations. 

7. Thermal expansion of metal tube parts. Expan¬ 
sion of the cathode due to the high temperature at 
which it is operated changes the grid-cathode spacing 
and the capacitance between the two elements. Com¬ 
pensating capacitor disks attached directly to the 
cathode made it possible to produce a tube with prac¬ 
tically zero temperature coefficient of capacitance with 
respect to heater voltage. Since the grid cathode is 
across few turns of the circuit inductance, a large im¬ 
provement in stability is needed before the overall 
frequency drift can be improved much by this expe¬ 
dient. 

No evidence was obtained for frequency change 
due to anode expansion. Grid expansion is more serious 
and some work was carried out to decrease frequency 
shift from this cause. Invar could not be employed 
because of the high temperature of the grid wires; 
molybdenum was finally recommended. 

22 2 2 Conclusions of the Tube 

Research Group 

Frequency drift in oscillators is partly chargeable 
to the tube and partly to the circuit in which the tube 
is used. The changes in frequency produced by elec¬ 
tronic effects in the tube may nearly always be com¬ 
pensated by proper circuit design. These tube effects 
are well understood. Defects in the tubes which are 
subject to correction are mostly of a mechanical 
nature. Actual thermal expansion of the tube ele¬ 
ments is not sufficiently great to cause serious capaci¬ 
tance changes in well-designed circuits, but if this 
expansion is magnified by mechanical leverages very 
large changes in tube capacitances may occur. Close 
spacing of tube elements makes the tube capacitances 
critically dependent upon maintenance of those spac- 
ings. Complex geometrical shapes of the elements pro¬ 
duce unexpected and unpredictable distortions and 


should be avoided. For use in a stable oscillator, the 
tube should have ample emission so that the number 
of electrons in the interelectrode space will be depend¬ 
ent only on the electrode potentials, the effect of 
which on frequency is predictable and may be nullified 
by proper circuit design. Low plate voltages, 125 and 
below, are important to reduce secondary emission 
which has considerable effect upon frequency stability. 

Frequency drift due to tubes alone, computed from 
15 seconds after the filament voltage is applied to the 
oscillator, can be held well within 0.002 per cent. 

22 2 3 Studies of the Oscillator Circuit 

An experimental study was made to determine the 
limitations imposed on frequency stability by standard 
oscillators of the Colpitts and Hartley types, with 
particular interest in the warm-up period. Other in¬ 
vestigations related to frequency versus filament and 
plate voltages, use of powdered-iron cores in the tun¬ 
ing coils, stability with output load coupling, and 
capacitance irregularities in ceramic and mica fixed- 
tuning capacitors. 

To avoid frequency drift due to heating of com¬ 
ponents, the tube socket was mounted 3 in. above the 
circuit parts at the end of wire rods. In addition, 
ceramic sockets which did not cause appreciable drift 
in frequency with temperature change were used, but 
even with these precautions there was appreciable 
drift due to the heat generated in the circuits by oscil¬ 
lations. This was overcome by operating the circuit 
for an hour with a spare tube prior to taking data. No 
attempt was made to temperature-compensate the com¬ 
ponents. Ceramic capacitors with approximately zero 
temperature coefficient were used. Tuning was ac¬ 
complished by moving a powdered-iron core within 
the coil by a micrometer adjustment. The choke coil 
for plate feed consisted of three universal-wound sec¬ 
tions having low distributed capacitance and a natural 
period of approximately 1,400 kc. 

The data in Figure 1 are characteristic of a Colpitts 
tapped-capacitanee type circuit. The Q of the circuit 
was about 140 and the total capacitance of the reso¬ 
nant circuit was about 270 ^f. The impedance of the 
tuned circuit was about 40,000 ohms. Figure 2 gives 
the same data at 11 me. In this case the inductance 
was adjusted to resonate with the same 270 ju./xf of 
capacitance employed in the 2-mc tests. Circuit Q 
was about 200, impedance about 10,800 ohms. 

Similar studies using the Hartley circuit gave 
about the same results as with the Colpitts circuit. 


CONFIDENTIAL 






156 


FREQUENCY-STABILIZED MASTER OSCILLATOR 


In each case use of a 955 tube gave somewhat greater 
stability than the 6C4. 

22 24 Component Effects 

Some of the ceramic capacitors were quite unstable 
even though of the same manufacture and same gen¬ 
eral type. Others were quite stable. Cores of copper, 
magnetite, powdered iron, and a combination of cop¬ 
per and magnetite were interchanged and the effects 
upon frequency change noted. It was discovered that 
a proper combination of copper and magnetite could 
be found which would give practically any degree of 
compensation with regard to changing plate voltage 
that might be desired. 

Over a test period of 120 hours, no effects due to 
permeability-aging of either powdered-iron or pow¬ 
dered-magnetite cores were noted. 

22-2-5 Oscillator plus Buffer Stage Stability 

The curves in Figure 3 give a general idea of the 
frequency drift of oscillator and output buffer stage. 


In this case the oscillator was loosely coupled to the 
6AG7 buffer acting as a class A amplifier. With 125 
volts on the oscillator plate, 1.25 volts were fed to the 
buffer grid circuit and 0.6 watt was developed in 
the buffer output load of 7,100 ohms. With 44 volts 
to the oscillator plate, the oscillator delivered 0.28 
volt to the buffer which developed 0.035 watt in 
the load. 

22-2-6 Conclusions of the Circuit Studies 

Frequency instability contributed by the tube in an 
oscillating circuit is caused mainly by changes in tube 
input and output resistances produced by changes in 
plate voltage or cathode emission. In this manner a 
phase shift of the regenerated oscillations occurs. This 
calls for an opposite phase shift in the resonant cir¬ 
cuit which is produced by operating at a different 
frequency. 

Proper choice of circuit elements can compensate 
frequency changes due to tube changes. Circuit Q 
should be as high as practical. Small tubes like the 



Figure 1. Experimental data showing effect at 2 me of plate and filament voltages on frequency and magnitude of 
frequency drift. 


CONFIDENTIAL 





































































































ADDITIONAL PRELIMINARY STUDIES 


157 



Figure 2.‘ Frequency stability with respect to time and plate and filament voltages at a frequency of 11 me. 


955 or the 6C4 should be used. The tube should be 
tapped across portions of the tuned circuit which are 
as small as is consistent with stable operation. A grid- 
circuit load of 500 ohms and a plate-circuit load of 
2,000 ohms seem to be a satisfactory arrangement. 
Low plate voltage is desirable; grid-leak resistance 
value is not critical. Coupling to the output stage 
should be low. Operating the buffer on the same fre¬ 
quency as the oscillator, which had a plate voltage 
near the upper limits used, made it possible to deliver 
a maximum of about 1 volt to the buffer. 

The change in permeability of a powdered-iron 
core in the tuning coil with change in r-f magnetic 
field can be utilized to improve the frequency stability 
with changes in plate voltage. 

22.3 additional preliminary studies 

Several multitube oscillator circuits were examined, 
but the conclusion was reached that better frequency 
stability could be achieved with well-designed single¬ 
tube oscillators. 


22 31 Oscillator Plus Amplifier 

For example, looser coupling to the resonant cir¬ 
cuit can be employed if the transconductance of the 
oscillator is higher. Therefore, if the oscillator could 
have its gain supplemented by an amplifier, very 
loose coupling could be used. But if there are other 
circuit phase shifts, the virtues of this system would 
not be realized. 

2232 Low-Frequency Amplifier Plus 

Heterodyne Detectors 

Amplification with minimum phase shift can be ac¬ 
complished by using a low-frequency amplifier. Two 
heterodyne detectors are required to accomplish the 
two necessary frequency conversions. In a case exam¬ 
ined, two i-f stages of high L/C ratio were employed, 
each stage having a single tuned circuit. One stage 
was tuned below the nominal intermediate frequency 
(0.29 me) and the other was tuned above the i-f fre¬ 
quency (0.39 me). The local oscillator frequency was 



































































































158 


FREQUENCY-STABILIZED MASTER OSCILLATOR 


9.32 me and the stabilized output was 9 me. At the 
mean frequency of the amplifier the coupling circuits 
acted like nearly pure reactances so that a change in 
transconductance of the tube or in tube capacitance or 
a change in intermediate frequency should produce 
very little change in phase shift. 

In this case, a change in local-oscillator frequency 
of 28 me produced a change in output frequency of 
only 2 me. 

22.3.3 Double-Heterodyne Oscillator with 
AFC and AVC 

Adding automatic frequency control to the circuit 
described above decreased appreciably the frequency 
shift. A maximum change of approximately 400 cycles 
from 10 me was produced by plate voltage changes up 
to 30 per cent. 

22,34 Oscillator with AYC 

Application of automatic volume control to a Col- 


pitts oscillator to prevent grid-current flow did not 
improve the frequency stability, indicating that, in 
the usual oscillator circuit, a better balance of input 
capacitance changes and better frequency stability are 
obtained when the grid is permitted to go slightly 
positive every cycle. 

22.3.5 Bridge-Stabilized Oscillator 

Another method to improve stability was by means 
of a thermally controlled resistance element which 
affected the balance of a bridged-T network in accord¬ 
ance with the amplitude of oscillation. Instead of the 
quartz crystal element originally employed by Meach- 
am, 1 customary circuit elements were used. This cir¬ 
cuit was more nearly independent of plate-voltage 
changes, although frequency change occurred as with 
the other multitube circuits if the filament voltage 
changed. Long time variations were much less with 
the bridge-stabilized oscillator, 10 to 20 cycles’ change 
being noted after a period of about 1 hour. 


1200 


w iooo 



OPERATING DATA : 


6C4: Ej * 6.3V E b * 127V 
6AG7:E f «6.3V E b *250V E S6 *I45V 
E b , as »3.IV 
INPUT * 1.25V RMS 
OUTPUT » G.5 V ACROSS 7100 OHMS 
• 0.6 WATT 







1 

_ D 











































0 20 40 60 80 100 

PER CENT POWER OUTPUT OF BUFFER STAGE 


















( 

BY D 

'ETUI* 

IING) 















































































l 

\ 





IFT 0 

IF OS 

iCILL 

ATOF 

* TUI 

SE A 

LONE 
















4 

















OF 

tIFT 

A 1 

OF 0 

ID B 

SCIL 

UFFE 

LATO 

R SI 

R TU 

rAGE 

BE 














































































0 


10 


5 6 7 

TIME t IN MINUTES 

Figure 3. Characteristics of an oscillator buffer stage operating at 10 me. 


| CONFIDENTIAL 






































































































DEVELOPMENT OF THE C-59 OSCILLATOR 


159 


224 DEVELOPMENT OF THE 

C-59 OSCILLATOR 

Since changes in the load circuit of any oscillator 
must have minimum effect on the frequency generated 
if high-frequency stability is desired, the first decision 
in developing an actual working model of a highly 
stable oscillator was to use a buffer amplifier tuned 
to twice the frequency of the oscillator. In this man¬ 
ner, the effect of the load upon the oscillator can be 
reduced by a factor of 100 to 1 for the reason that 
the output circuit of the buffer looks to the oscillator 
like a very low r value of inductive reactance and varia¬ 
tions in this reactance, which are low, have very little 
effect upon the oscillator frequency. 

22 4 1 Temperature Effects 

Temperature changes play an important part in 
instability of oscillator frequency. There are two sep¬ 
arate causes, change in temperature of a tube or com¬ 
ponent due to the self-heat generated by a current 
flowing through it and change in ambient tempera¬ 
ture. Both conductors and insulators are subject to 
this effect, the first because they have resistance and 
the second because of dielectric hysteresis or power 
factor. Power factor is practically independent of 
capacitance, voltage, or frequency but is influenced 
by temperature, nearly always becoming higher as the 
temperature is raised. 

The temperature coefficient of an oscillator tank 
circuit built up of commonly used parts will fall in 
the range of 10 to 100 parts per million per degree 
centigrade (ppm/°C). Variations in resistors, other 
dielectrics in the circuit and the tube will bring this 
value to the range of 10 to 500 ppm/°C. Some of the 
elements will have positive and others negative co¬ 
efficients. While there is no simple way of reducing 
the temperature coefficient of inductors, capacitors are 
available having a wide range of temperature coeffi¬ 
cients ranging from —750 X 10 _fi to -f- 120 X 10 -1 ' 
Hit/nd/oC. 

Most of the effect of temperature tends to lower 
the frequency of an oscillator so that capacitors hav¬ 
ing negative temperature coefficients can be used to 
correct this effect to some extent, particularly when 
ambient temperature changes are to be compensated. 
The use of such capacitors produces warm-up drift, 
however. 

Resistors of the metallized-filament type have low 
temperature coefficients and are useful in the present 
problem. 


22 4 2 Effect of Humidity 

Changes in humidity produce frequency changes be¬ 
cause of (1) change in the dielectric constant of air, 

(2) change in the surface resistivity of solids, and 

(3) change in the volume resistivity of solids. 

The first factor varies from approximately 1.0005 
to 1.0016 over a humidity range from 0 to 100 per cent 
and a temperature range of from 0 to 60 C. 8 

Since both L and C are affected by humidity, only 
two solutions are possible. Either make L and C so 
that humidity does not affect their constants or isolate 
humidity so that it does not reach L and C. 

In a typical 10-mc oscillator the frequency varied 
over a range of 0.002 and 0.05 per cent when the 
humidity w r as varied from 30 to 100 per cent. This 
oscillator was substantially airtight. Gaskets were 
used on all covers and packing was used around the 
control shafts. Entering leads were sealed with glyptol 
cement. All coil forms and solid dielectrics w r ere of 
ceramic or mycalex. A dessicant of silica gel w r as 
placed in the interior of the oscillator. 

22 4 3 Importance of High Q 

Although the variation of voltage on filament and 
plate elements of the tube and the magnitude and 
constancy of the applied load and of other circuit 
elements have important bearings on the frequency 
constancy, the effective Q of the tank circuit is per¬ 
haps the greatest factor having to do with the genera¬ 
tion of constant frequency. Frequency stability is di¬ 
rectly proportional to the Q of the tank circuit, since 
the frequency change necessary to compensate the 
phase shift resulting from changes in load, tube re¬ 
sistance, and other circuit resistance is proportional 
to 1 /Q. Thus Q should be as high as possible. 

2244 Tuning Methods and Mechanisms 

Since high values of Q are so important and since 
a miniature oscillator presents new problems of ob¬ 
taining good Q with small coils, a study was made 
of several methods of tuning the tank circuit. 

Rotary-coil tuning gives a large inductance ratio, 
uniform Q over the tuning range, but poor reset be¬ 
cause of sliding contact between winding and slider. 
Variometer tuning gives a large inductance ratio but 
very poor Q. Flat-spiral coil with copper-disk tuning 
gives a low inductance ratio, poor Q and a nonuniform 
tuning curve. Copper-plug tuning gives a low in¬ 
ductance ratio, somewhat higher Q than with the 
methods mentioned just above, and a Q ratio which 






160 


FREQUENCY-STABILIZED MASTER OSCILLATOR 


is approximately of the same magnitude as the induc¬ 
tance ratio. Its great advantage lies in the fact that 
the fineness of tuning and reset value are limited 
only by the mechanical perfection of the threaded 
actuating mechanism. 

Since the oscillator was to cover a wide range (2 
to 20 me) and because of mechanical limitations in 
making one tuning mechanism to cover the entire 
range continuously, it was decided to break up the 
range into bands, assuming a frequency change of 
0.02 per cent per dial division, dial divisions being 
limited to 1,000. Thirteen sets of coils were required 
for the oscillator and five for the buffer. 

22.4.5 jrj na j Oscillator Tuning Arrangement 

The final design of the tank circuit for band 13 
(17.50 to 20 me) is shown in Figure 4 to consist of 
the inductance properly tapped and the tank tuning 
capacitance of the ceramic tubular type having, theo¬ 
retically, zero temperature coefficient. Actually two 
such capacitors were connected in parallel and so 
chosen that the effective temperature coefficient of the 
combination was close to zero. The capacitors were 
rigidly mounted to the coil form platform by a metal 
end-on capacitor fastened with a machine screw to 
the platform. 



1’iuuKE 4. Oscillator coil witn transparent cover. 


The coil form was molded from Styramic, a chlori¬ 
nated diphenyl-polystyrene compound having a higher 
heat-distortion temperature and better machinability 
than polystyrene. The conductor was cemented to the 
form with polystyrene cement. The coil was held in 
the oscillator by means of three ball feet with threaded 
adjustment for length which allowed the tank assem¬ 
bly to be moved ±Vs inch from the nominal position 
with respect to the tuning plug. This adjustment pro¬ 
vides a ready means of producing oscillators which will 
have physically identical coils and which will produce 
the same frequency at a given dial setting for each 
oscillator without the necessity of altering the coils. 

The exact position of the taps for grid, cathode, 
and plate was determined for each band to give the 
smallest frequency change with line-voltage varia¬ 
tions. The temperature coefficient of the combination 
of coil, mounting, and tuning plug mechanism was 
within ±6 ppm/°C in frequency. The coil alone had 
a coefficient of ±1.5 ppm/°C. This was attained by 
proportioning the length of the coil to its diameter 
so that the inductance was decreased due to length¬ 
wise expansion at the same rate as the inductance was 
increased by radial expansion. For the combination 
of copper wire, close wound on a Styramic form this 
ratio was 1.75. 

The ±6 ppm/°C spread was due to the fact that 
the overall expansion of the tuning plug mechanism 
was different when the plug was withdrawn than when 
it was inserted in the coil. This effect was minimized 
by supporting the copper tuning plug on a pillar 
of Styramic, which did not touch the lead screw ex¬ 
cept at a point where the length of the pillar was cor- 



Figure 5. Tuning mechanism and dial, assembled. 


\ CONFIDENTIAL 












DEVELOPMENT OF THE C-59 OSCILLATOR 


161 


rect to counteract the expansion of the coil from its 
ball feet to its electrical center, lengthwise. 

( oil and capacitance were sealed in airtight con¬ 
tainers. 

To reduce further the effect of humidity on frequen¬ 
cy, the oscillator-buffer unit was made as tight as 
possible without resorting to sealed joints. A container 
of silica gel was attached to one end of the chassis 


with provision for removing it so that it could be 
baked for .3 hours at 300 F to prepare the crystals 
for another cycle. 

A great many measurements were made on this 
oscillator-buffer unit to determine its performance. 
The results of these measurements employing test 
specifications for checking Navy transmitters are sum¬ 
marized in Table 1. 



Figure 6. Oscillator, bottom side plate and shield, cans removed. Entire oscillator and buffer assembly requires 
7 H x 2 H x 3% in. of space and weighs 3 lb with 1 set of coils and tubes. Coils to cover entire range of 2 to 20 me 
weigh 2.1 lb, and coil-carrying case without coils weighs 4 lb. 


CONFIDENT! 


ti 




























Chapter 23 

PICKUP TUBE FOR RECONNAISSANCE TELEVISION 


A line-mosaic iconoscope or television pickup tube for use 
in reconnaissance. Several tubes were constructed but the 
project ended with the feeling that to eliminate leakage be¬ 
tween the elements would require considerable research and 
that an equal amount of effort would develop a two-dimension 
tube with greater sensitivity. 

23i INTRODUCTION 

roject C-6‘2 a called for the investigation of a 
line-mosaic pickup tube to determine whether or 
not it would be suitable for application in a high- 
definition (1,000-line) reconnaissance television sys¬ 
tem. 

Such a system requires a higher definition than can 
be obtained from the normal iconoscope, the latter 
being primarily limited by the size of the scanning 
spot produced by the electron gun. Since it was felt 
that a rather extensive research would be required to 
increase the definition of this type of iconoscope or a 
low-velocity iconoscope to a point where it could be 
used for this purpose, an investigation of the possi¬ 
bilities of the line-mosaic tube might obviate this ex¬ 
penditure of time and facilities if it were found that 
a tube of this type could be used. 

A line-mosaic pickup tube consists of an electron 
gun capable of producing a very narrow scanning 
spot and a mosaic made up of a row of fine vertical 
line elements. A signal plate on the back of the mosaic 
is connected to the external signal lead and serves as 
capacitance coupling to the elements. The scanning 
spot is deflected across the mosaic by means of a 
magnetic deflecting yoke. 

The optical system used with this pickup tube is 
different from that of a conventional iconoscope, in¬ 
asmuch as it must provide vertical deflection. This 
can be done as follows: An objective lens images the 
scene to be transmitted onto a slit image plane which 
permits the light from a single horizontal line of the 
image to reach the mosaic. The image as a whole is 
moved vertically across the slit at frame frequency by 
a rocking mirror located between the objective and 
the image plane. To simplify the geometric arrange¬ 
ment of the optical system, a weak cylindrical lens 
may be used in combination with the objective so that 
picture elements are imaged as horizontal lines on 

“Project C-62, Contract OEMsr-706, RCA Laboratories. 


the surface containing the slit and as vertical lines on 
the mosaic. 

The frame frequency used is much lower than is 
used in commercial systems. Its maximum is such 
that the picture can be transmitted over a channel 
having normal band width and the frame frequency 
may be much lower, depending upon the sensitivity 
needed, the means of reconstructing the picture and 
other factors. In view of the slow repetition rate, the 
picture cannot be reproduced on an ordinary kino- 
scope, but instead it is formed on a tube having a long 
decay period, or photographically, or by mechanical 
recording. 

A pickup tube based on a line mosaic obviously can¬ 
not employ surface storage of the photoelectric charge. 
Instead the charge is stored for the duration of a 
single line. This reduces the intrinsic sensitivity of 
the pickup device. However, since the design of the 
tube is such as to permit saturated photoemission and 
because the frame frequency is low, the overall sensi¬ 
tivity can he made comparable with that of the nor¬ 
mal iconoscope. 

Prior to the start of the investigation, slit-aperture 
guns (electron guns having a fine rectangular slit at 
the crossover) had been tested and found to have a 
horizontal resolution of well over 1,000 lines. There¬ 
fore, this type of gun could be used without change 
in the line-mosaic pickup tube. Furthermore, a wash- 
off relief photographic process had been developed 
which, with slight modification and the working out 
of a specific technique, seemed to offer a method by 
which the fine line structure of the mosaic elements 
could be readily made. 

23 2 LINE-MOSAIC PICKUP TUBE 

The arrangement of the various elements making up 
the line-mosaic pickup tube can be seen in Figure 1. 

As has been stated above, a slit-aperture gun has 
sufficient resolution in the horizontal direction to meet 
the requirements of the line-mosaic pickup tube. This 
gun employs a two-lens electron optical system, the 
cathode, grid, and first anode forming one lens, and 
the field between the first and second anode, the 
other. Electrons from the cathode are converged into 
a crossover near the first lens, this crossover being 
the exit pupil of the first lens system and the narrow- 


(ONFIDENTIAL 



162 






LINE-MOSAIC PICKUP TUBE 


163 



SIGNAL 

LEAD 


est portion of the beam on the cathode side of the 
second lens. A diaphragm having in it a rectangular 
aperture approximately 5 mils high and V 2 to 1 mil 
wide is placed at the crossover. The second lens is ad¬ 
justed to image this aperture on the mosaic, so that the 
resultant spot is a short narrow vertical line. It should 
be pointed out that the spot is larger than the true 
geometric image of the slit aperture, because of the 
spherical aberration of the second lens, and conse¬ 
quently, if the slit size is reduced, the current reach¬ 
ing the spot is reduced without appreciable reduction 
in size, and for this reason, the redesign of the elec¬ 
tron gun to give a thousand-line resolution in both 
directions represents a difficult research problem. 

The scanning spot is deflected horizontally by 
means of a conventional magnetic deflecting yoke and 
suitable driving circuits. No new techniques are in¬ 
volved in the horizontal scanning operation. 

23.2.1 Mosaic Types Investigated 

Two types of mosaic were tried in the investigation. 


Both used a glass plate 30 mils thick as dielectric 
but differed in the arrangement of elements. One of 
them consisted of a row of fine line elements insulated 
from each other. The elements were about V 4 in. in 
length with (500 to the linear inch. Platinum was 
used as the base of the line elements, which were pho¬ 
tosensitized with silver and cesium during the proc¬ 
essing of the tube. A horizontal strip of silver on the 
back of the glass dielectric serves as the signal plate. 
The arrangement of this mosaic is illustrated in Fig¬ 
ure 2 A. 

The second type of mosaic employed a barrier grid 
to prevent electrons scattered by one element from 
reaching the others. The nature of the barrier grid 
can be seen from Figure 2B. As before, the mosaic 
consists of line elements, hut between each pair of 
elements is a conducting strip of connected metalized 
bands at either end. These shielding strips not only 
serve as barrier grid but also as the signal plate. 

The elements of either type of mosaic are returned 
to their normal or reference potential each time the 



Figure 2. Details of signal plate for line-mosaic tube. 


CONFIDENT] 


























































































164 


PICKUP TUBE FOR RECONNAISSANCE TELEVISION 


beam passes over them by an equilibrium between 
the secondary emission electrons collected and the 
beam electrons. This means that the element poten¬ 
tials will be approximately that of the secondary elec¬ 
tron collectors and that these collectors cannot be 
used to obtain saturated photoelectric emission. There¬ 
fore, the tube is divided into two compartments by a 
partition at right angles to the lines forming the ele¬ 
ments. The beam strikes only the portion of the ele¬ 
ments extending into the lower compartment and this 
compartment contains the secondary emission col¬ 
lector (i.e., the second anode). The upper compart¬ 
ment contains the photoelectric collector, and the light 
image is projected on the portion of the lines extend¬ 
ing into this compartment. The photoelectric collector 
can be made positive with respect to the secondary- 
emission collector without disturbing the secondary- 
emission equilibrium of the mosaic elements. There¬ 
fore, the photoelectric emission from the illuminated 
elements can be saturated. Reference to Figure 1 will 
make clear the arrangement of partitions and com¬ 
partments. 

23.2.2 Construction of Pickup Tube 

Two line-mosaic tubes were built to test the feasi¬ 
bility of the pickup devices described above. Each tube 
contained sections of ordinary and barrier-grid line 
mosaics, thus making it possible to test the two types 
under the same conditions of activation and operation. 

The construction of the gun and glass envelope 
followed conventional lines and will not be further 
described. 

The preparation of the mosaic involved a special 
photographic washoif relief procedure which was in 
part developed for the purpose. A plate of 30-mil glass, 
4 1 / 4 in. long and 2% in. wide formed the base of the 
mosaic. After being thoroughly cleaned, the glass was 
coated with a sensitizing solution prepared as follows: 

Gum-Bichromate Process Solution 

Solution I 30 g gum arabic in 100 cc water 
Solution II 5 per cent potassium dichromate 

One part of Solution II is added to two parts of 
Solution I. The formula produces a gum-arabic film 
of low sensitivity but having a very high contrast and 
resolution, which is required for the line structure. 

The solution is flowed over the glass plate at room 
temperature and let to drain until dry. When dry, the 
plate is found to be covered with a thin transparent film 
of sensitive material which hardens when exposed to 
the light of the short-wave end of the visible spectrum. 


The negative used for preparing the line structure 
was a contact print transparency made from photo¬ 
graphically reproduced copies of a 600-line ruled grat¬ 
ing. The negative had 600 lines per inch, covering an 
area 4 in. long by 14 in- high. 

The sensitized plate was exposed to the light of a 
mercury arc through the 2,400-line negative and then 
was developed in water at 10 C until the unexposed 
portions of the sensitive film were completely washed 
away. Platinum was then sputtered on to the relief 
image thus obtained until the transmission was re¬ 
duced to 10 per cent. The plate was then immersed 
in boiling water. This dissolved the gum-arabic lines 
in the exposed portions and carried away the platinum 
covering them. An array of sharply resolved platinum 
lines suitable for the mosaic remained on the glass 
plate. 

On the half of the mosaic to be used as a barrier 
grid, a platinum strip, in the form of reduced Hano- 
via Platinum Bright painted on the glass, is made 
so that it is just in contact with the lines. Every other 
line is then cut mechanically with the aid of a fine 
needle so that it does not make contact with the plat¬ 
inum strip. These electrically insulated lines serve 
as the mosaic elements, while the lines remaining in 
contact with the strip form the barrier grid. A pho¬ 
tographic process could be used giving alternate long 
and short lines if it was desired to produce these 
barrier-grid mosaics in any quantity. 

On the back of the glass, opposite the portion of the 
mosaic which did not have the barrier grid, a signal 
plate was made in the form of a strip of reduced 
platinizing solution. 

The first mosaic to be formed in this way was then 
placed in an evaporating chamber and a very thin 
layer of silver was deposited on the surface. This sil¬ 
ver film was so thin that its electrical conductivity 
was not measurable by ordinary methods. The mosaic 
was then sealed in a glass envelope with a gun, col¬ 
lector electrodes, and partitions arranged as described 
above. During the preliminary exhaust the tube was 
baked at 450 C for more than an hour. This removes 
contamination and gas from the glass and metal parts 
and also will further reduce the conductivity of the 
silver film by breaking it up. The activation procedure 
which was followed was the usual oxidation of the 
silver, addition of cesium vapor, and heat treatment. 

Since the sensitivity of this tube was very low and 
there was considerable evidence of leakage between 
elements, a second tube was built. The platinum line 


CONFIDENTIAL 





CONCLUSIONS 


165 


structure was formed in the same manner but silver 
was not deposited on the platinum outside the tube. 
Instead, silver evaporators were mounted inside the 
glass blank in such a way that silver could be deposited 
after the initial exhaust was complete. The silver layer 
used was considerably thinner than that in the pre¬ 
vious tube. Activation was carried out in the same 
way as before. The sensitivity of this tube was con¬ 
siderably higher than that of the first tube. 

Figure 3 is a photograph of one of the line-mosaic 
pickup tubes. 



Figure 3. Photograph of one of reconnaissance tele¬ 
vision tubes. 


232-3 Experimental Results 

The equipment used to test the performance of the 
line-mosaic tubes operated with a horizontal deflec¬ 
tion frequency of 15 kc, but since the amplifier re¬ 
sponse extended to a high frequency and the ampli¬ 
tude could be decreased at will, the test equipment 
did not limit the ultimate resolution of the system. 

The first tests were to determine whether or not the 
gun was capable of resolving the line structure. It 
was found that the line structure could be resolved, 
but that the actual beam size was somewhat larger 
than the element width. Since an actual mosaic of this 
type would have two or three line elements per picture 
element, the spot size was considered adequate. 

Tests were made with a line of light of variable 
width to determine the definition in terms of the 
sharpness of the edge of the reproduced image of this 
line. Overall tests were also made with the projected 
image of a continuously run moving picture film 
whose frame speed was synchronized with the vertical 
frame frequency of the reproducing equipment. This 
second test closely simulates the actual working con¬ 
dition of the tube if allowance is made for the higher 
line and frame frequency. 


The first tube tested was found to be very insensi¬ 
tive, so much so that it was very difficult to distin¬ 
guish the light image from the spurious image due 
to the line structure. There was evidence that even 
it a stronger signal could be obtained, the resolution 
would not be adequate. The cause of this loss in reso¬ 
lution appeared to be due partly to leakage and partly 
to redistribution losses. No estimate could be made 
of the definition on the barrier-grid side. 

The second tube was prepared in such a way as to 
give a higher sensitivity and somewhat lower leakage. 
On test it was found that at high light levels, where the 
signal was well above noise and other spurious effects, 
the definition was very poor. As the light level was 
lowered the definition improved and at extremely low 
values of signal the definition approached that estab¬ 
lished by the line structure. Again leakage appeared 
to be the major cause of loss of definition with redis¬ 
tribution contributing an increasing amount as the 
light level was increased. The sensitivity of the barrier- 
grid side was very much lower than when the barrier 
grid was absent. Electric leakage between the elements 
and the barrier lines was responsible for the low re¬ 
sponse. In fact, this leakage was so high that the 
grid could not be made sufficiently negative to exclude 
all redistribution. 

23 3 CONCLUSIONS 

The tests outlined indicate that line-mosaic pickup 
tubes as constructed are not satisfactory from the 
standpoint of definition for the specific reconnais¬ 
sance television application. The primary reason for 
this is leakage between elements where there is no 
barrier grid, and between the elements and the barrier 
grid with the barrier-grid mosaic. Redistribution also 
decreased the definition when no barrier grid was used. 

This leakage probably is not fundamental. However, 
to overcome it a great deal of research on methods of 
making the line mosaic would have to be carried out, 

A similar amount of research on a tube using a two- 
dimensional mosaic would also lead to a tube which 
could be used for the required reconnaissance. Such a 
tube would inherently be much more sensitive than 
any line-mosaic device, and the possible greater size 
of the former would be compensated by the simpler 
optical system required. Therefore, if a more exten¬ 
sive research program is to be undertaken to develop 
a reconnaissance television system, it would be better 
to concentrate on the working out of a two-dimen¬ 
sional mosaic pickup tube. 


CONFIDENTIAL 








Chapter 24 

SOUND RECORDING ON MAGNETIC MATERIALS 


Development of a small spring-driven magnetic recorder- 
reproducer, in size about that of a 16-mm magazine-type mo¬ 
tion picture camera, weighing 6 lb, giving telephone quality, 
and capable of recording in any position for a total time of 
30 minutes. Further research should include reference to the 
final report on Project 13.3-87, 2 which gives a general sur¬ 
vey of recording equipment available at the time. 

24.1 INTRODUCTION 

at the initiation of Project C-69 a there was no 
lightweight portable pocket-sized recorder, in 
spite of the fact that all the different methods of re¬ 
cording— optical, mechanical, and magnetic — had 
been well developed. 

The completion of the project 1 was marked by the 
production of a model which weighed 6 lb, including 
amplifier, batteries, microphone, and earphone. The 
recording medium was a stainless-steel wire 0.006 in. 
in diameter from which the recording could be erased 
when desired whereupon another recording could be 
made. The instrument can be attached to an auxiliary 
drive unit which operates from any 12-volt supply, 
such as the battery of a jeep or tank. This auxiliary 
drive eliminated the burden of winding the spring and 
gave more audio power on reproduction so that a 
number of observers could listen to the message simul¬ 
taneously. 

The work was originally undertaken to provide re¬ 
connaissance men with a means of recording what they 
saw and as an aid to memory and to preserve the in¬ 
formation in case the scout was a casualty. 

The development was used by government services 
as a basis for specialized designs. Much of the expe¬ 
rience gained was successfully incorporated in mag¬ 
netic recording equipment of different construction. 

24.2 advantages of wire recording 

In preference to other means of recording, mag¬ 
netic wire recording was chosen for the following 
reasons: 

1. Mechanical vibrations do not seriously interfere 
with the recording or reproducing processes. 

“Project C-69, Contract OEMsr-833, The Brush Development 
Company. 


2. In any recording process in which space is a factor, 
the amount of recording medium used must be kept 
to a minimum. The space occupied by wire is a mini¬ 
mum for any required time of recording, as compared 
to that required by other media. 

3. Little power is needed to make a recording. 

4. The signal can be reproduced many times with¬ 
out deleterious effects on the recording. 

5. The signal can be erased and the medium re¬ 
used. 



Figure 1. Portable recorder in use. 


24.3 THE recorder 

To eliminate the weight of batteries to supply run¬ 
ning power for the reels, a spring-driven mechanism 
of the motion-picture camera type was selected. 

A test setup was built consisting of two reels (take- 


(/ONFIDE35TTIAL 


166 







THE RECORDER 


167 


up reel and supply reel), an electric-motor drive, and 
a magnetic ring head attached to a level-winding 
mechanism. This was used to determine what mechan¬ 
ical features were to be desired. From these investiga¬ 
tions it became clear that the best method of keeping 
the wire under continuous tension was to control the 
starting and stopping by means of a brake applied 
to the supply reel. It also was found to be desirable 
to provide a winding-shaft spring in parallel with the 
mainspring which would be wound by the mainspring 
when the drive mechanism is triggered to stop. 

The level-wind device (similar to that used in a 
fish-line reel) was employed to obtain reasonably level 
winding of the wire. Level winding is necessary to 
prevent tangling of the wire on the reel and also to 
give good feed through the head with a minimum of 
vibration. This is advantageous in as far as back¬ 
ground noise is thereby reduced. 

Several 16-mm cameras were secured and studied 
to derive the drive mechanism. A spring was finally 
selected from an Eastman Kodak Company Model Iv 
camera and Avas incorporated in the experimental 
model, designated as D-103. 

The mechanism contains a crank-wound spring 
which drives a gear train to the take-up reel. The reel 
rotates at about 350 rpm. This spring cannot be Avound 
while driving the unit since it is of the type which 
has one end of the spring held stationary. The free 
end of the spring rotates in one direction Avhen being 
Avound and in the opposite direction when driving 
the gear train. This method of winding permits auto¬ 
matic stopping of the drive mechanism before the 
spring is completely exhausted and assures relatively 
constant speed of the take-up reel during its operating 
cycle. An alternative spring system Avas proposed 
which would enable continuous Avinding, but could 
not be Avorked out in a small space to include the 
advantageous feature of automatic shutoff Avhen the 
spring is nearly exhausted. 

Associated with the reel gear train is the governor 
gear train which operates the governor at a higher 
speed. The power is applied to the take-up reel through 
a friction clutch. This maintains synchronism of the 
level Avind in both the rewind operation (when the 
supply reel is driven) and also for recording or play¬ 
back. In this arrangement this level wind shaft winds 
wire on the take-up reel Avith a constant pitch of about 
140 to the inch. Due to the variable speed of the sup¬ 
ply reel with reference to the take-up reel the pitch 
with which the Avire is rewound on the supply reel 
varies betAveen 90 and 140 to the inch. Such variations 


in pitch have not been found to interfere Avith the 
proper operation of the equipment. 

The recorder is controlled by a single lever which 
releases the brake acting on the supply reel and the 
winding shaft spring, putting the unit into operation. 
Operating time, starting Avith the main spring fully 
Avound is approximately 30 seconds. The reel contains 
sufficient Avire to record for a total period of 30 min¬ 
utes. The reel diameter is 3 in. and its width 3 A in. 

A feature of the portable recorder is the incorpora¬ 
tion of three automatic stops. There is a worm-driven 
lead screAV and nut which is timed so that when all 
the wire is wound on the take-up reel it automatically 
trips the control lever into the off position. When the 
portable recorder is used in conjunction with an ex¬ 
ternal drive, the same nut operates a miniature switch 
to open the power supply for the external drive thus 
insuring that the Avire will not be pulled off the reels 
on either playback record or rewind. 

Provision Avas made to use the crank for winding 
the spring and rewinding the Avire. The reel shafts and 
the spring shaft Avere fitted with splines to facilitate 
operation by the hand crank as Avell as by an external 
drive. 

243 1 Wire Details 

Wire 0.006 in. in diameter Avas decided on as a com¬ 
promise betAveen mechanical strength and a reasonably 
small storage factor. Magnetic qualities Avere deter¬ 
mined by running a loop of test wire continuously 
over recording, reproducing, and erasing heads. It 
Avas determined that a Avire speed of approximately 
3 ft per second gave a frequency response of satisfac¬ 
tory intelligibility. Several wire samples were tried 
out and it Avas found that heat-treatment of these 
wires is needed for satisfactory signal-to-noise ratio. 
Carbon-steel wire properly heat-treated has a signal- 
to-noise ratio of 28 db or better. Stainless-steel wire 
later became available, with the advantage of corro¬ 
sion resistance. 

24.3.2 The Erasing and Recording Process 

In this particular recorder the method of d-c eras¬ 
ing and d-c biasing is employed. This means that in 
the erasing process the wire is magnetically saturated. 
In the recording process a d-c field considerably 
smaller than the erasing field is superimposed upon 
the signal current. The biasing field is of opposite 
direction to the erasing field and chosen so as to select 
a zone in the hysteresis loop of the Avire which for the 
signal amplitude is essentially linear. 


CONFIDENTIAL 








168 


SOUND RECORDING ON MAGNETIC MATERIALS 



(0y* SPRING WIND 
SPLINE 


REMOVABLE COVER 


SPRING MOTOR 
CONTROL LEVER 


MICROPHONE 

JACK 


REWIND a RECORD 
STOP INDICATOR 


111 




AMPLIFIER SWITCH AND 
VOLUME CONTROL KNOB 


AUX. 12-VOLT 
pf' MOTOR DRIVE 
SPLINE 


AUX 12-VOLT 
CONNECTIONS 


REWIND SPLINE 


EAR PHONE 
CONNECTOR 



AUX 12-VOLT 
AUTOMATIC STOPS 


MAIN BATTERY 
CARRIER 


AMPLIFIER SWITCH AND 
VOLUME CONTROL KNOB 


ERASE AND 
NORMAL SWITCH 


PLASTIC TUBE 
BATTERY CARRIER 


SPRING MOTOR 
CONTROL LEVER 


I 




Figure 2. Construction of recorder. 














THE RECORDER 


169 


24 3 3 Recording Head 

The recording head is constructed of two L-shaped 
Mu-metal pole pieces. The air gap over which the 
wire passes produces the leakage field necessary to 
generate the longitudinal magnetic recording pattern. 
A shoe is provided to guide the wire as it passes 
through the head, thus securing good magnetic con¬ 
tact. 

The recording head is wound with 1,000 turns of 
No. 40 wire having a d-c resistance of about 70 ohms. 
The head has an inductance of 16 mh and a Q of 1.2 
measured at 1,000 cycles per second. The erase current 
is 20 ma, and the biasing current is 7 ma. The maxi¬ 
mum signal current can be 1 ma, with resultant undis¬ 
torted magnetic impression. The unequalized power 
requirement for a signal of 1,000 cycles is of the order 
of V 2 mw. To obtain better h-f response, equalization 
must be provided in the recording channel, which in¬ 
creases the power requirements. 

The recording circuit consists of a carbon micro¬ 
phone (Type T-45), a 4^-volt “recording” battery 
to supply the current for the microphone, a trans¬ 
former, and a resistor. The battery also supplies cur¬ 
rent for polarizing and for erasing the wire. A switch 
is provided which in its “normal” position (not 
“erase”), prepares the circuit for recording or play¬ 
back. The microphone plug has to be inserted into its 
jack to close a miniature switch which controls the 
signal and polarizing currents. 

The three cells, used for pen-sized flashlights, sup¬ 


plying the 4 V 2 volts for the recording circuit were 
good for two hours of use. 

The playback level obtained from the head is about 
1 mv at 1,000 cycles per second. 

24 3 1 Self-Contained Playback Amplifier 

Because of the low output level, some means of 
amplification is necessary during reproduction. A 
hearing-aid amplifier was found to be suitable and 
is employed for this purpose. The three-stage ampli¬ 
fier is connected to the head by a small transformer 
designed to match 200 ohms to 1 megohm. Special 
care had to be taken to prevent microphonics, since 
these amplifiers are not normally intended for use in 
connection with rotating equipment. Rubber isolation 
proved successful. 

24 3 5 1 2-Volt Auxiliary Drive 

A separate semiportable motor-driven playback- 
rewind-erase unit was designed to operate as follows: 
The portable recorder can be clamped into position, 
where it automatically makes the proper mechanical 
and electrical connections. The auxiliary drive con¬ 
sists of a 12-volt d-c motor, three-tube amplifier using 
a vibrator as B supply, crystal microphone, earphones, 
and controls. It is contained in a plywood cabinet 
10x16x17 in. and weighs about 30 lb. Power consump¬ 
tion is approximately 4 amp at 12 volts. 

In using the two units together, the spring mecha¬ 
nism is by-passed. This assembly of two units can be 
used for all functional operations. 


CONFIDENTIAL 














































PART VIII 

MISCELLANEOUS STUDIES 


|£ CONFIDENTIAL 










Chapter 25 

SUBSTITUTES FOR NATURAL QUARTZ FOR FREQUENCY CONTROL 


STATE OF THE ART 

rior to this project, the art of high-frequency 
control by means of quartz-crystal sections had 
been highly developed, but no synthetic (man-made) 
crystals for frequency control purposes had been 
found with the exception of Rochelle salt, which is 
not suitable on account of its chemical instability 
and extreme frequency variations with temperature. 
Since the sources for high-grade quartz crystal suit¬ 
able for oscillator use are almost entirely outside 
the United States, it appeared most desirable that a 
way be found to manufacture frequency controlling 
elements from domestically available raw materials. 

Project C-29 a had the purpose of surveying possible 
lines of attack for this problem. 

25.2 WORK ACCOMPLISHED 

By a survey of available information, classification 
of such information, and some original theoretical 
work, the conclusions were reached that the only suit¬ 
able substitute for natural quartz for frequency con¬ 
trol in the conventional r-f range would be synthetic 
piezoelectric crystals and that there was a chance of 
finding such crystals, both among materials requiring 
high-temperature methods for crystal growth, and 
water-soluble materials. A number of crystal sub¬ 
stances were proposed whose properties appeared 
promising. Some preliminary measurements were 
made. 

The final report 1 was of a survey nature and no 
data on temperature dependence of resonant frequen¬ 
cy, which is the most critical characteristic for the 
purpose considered, could be obtained within the 
scope and the time limit for the project. 

Easing of the quartz supply situation made continu¬ 
ation of work on synthetic crystals for frequency con¬ 
trol unnecessary at the particular time. As a long- 
range project, however, the Signal Corps has con¬ 
tinued its interest and the contractor plans to carry 
on the work begun under Project C-29. 


“Project C-29, Contract No. OEMsr-120; Brush Development 
Co. This summary was written from the final report 1 on 
the project and from a letter from A. L. Williams, September 12, 
1945, citing opinions of Jaffe and Baerwald, who did the work 
under the project. 


The final report contains an extensive bibliog¬ 
raphy (180 items) on frequency instability in oscil¬ 
lators, methods of increasing stability, quartz reso¬ 
nators, etc. In addition, the tables in the final report 
give much data on substances considered as possible 
quartz substitutes. 

253 CONCLUSIONS 

The only really promising way to replace natural 
quartz for frequency control in the range of 2 to 50 
me is to provide a suitable artificial piezoelectric mate¬ 
rial. Only thickness-controlled plate resonators vibrat¬ 
ing in a shear mode and preferably suspended in cor¬ 
ner- or edge-clamped holders come into question. In 
order to avoid length- or width-controlled parasitic 
vibrations, one of the outstanding difficulties at ele¬ 
vated frequencies, only crystals belonging to certain 
classes and only a few specified cuts are recommend- 
able. The simultaneous achievement of freedom from 
parasitics and of highly reduced temperature depend¬ 
ence is still problematical. Substances in which crystal 
lattices are spatial frameworks of strong chemical 
bonds are required. Associated general properties, 
mostly necessary, partly highly desirable, are high 
melting point, chemical stability, good mechanical 
properties, low mechanical and electrical losses, and 
high mechanical wave velocities; good growing habits, 
i.e., possibility of obtaining faultless single crystals of 
about centimeter size, and fair piezoelectric coupling 
coefficient of the selected cut are also necessary. 

Only high-melting crystals can be fully equivalent 
to quartz for application to high-frequency control. 
From the known crystal-chemical properties of beryl¬ 
lium oxide, it is concluded that this substance might 
yield useful high-frequency shear plates, possibly even 
superior to quartz plates. Several leads for an attack 
on the problem of obtaining beryllium oxide in large 
single crystals are available. Artificial growing of 
large quartz crystals is a problem of a similar nature. 
The advantage in case of quartz would be that all 
physical properties and parameters are well known 
and that it would represent the most direct replace¬ 
ment of the natural product. Aluminum orthophos¬ 
phate and aluminum metaphosphate, which are related 
to quartz in their crystal structures, may have sim- 


173 





174 


SUBSTITUTES FOR NATURAL QUARTZ FOR FREQUENCY CONTROL 


ilar physical properties but seem to be easier to grow. 

Certain water-soluble crystals offer some promise as 
substitutes for quartz, with some sacrifice in the pre¬ 
cision, constancy, and highest frequency attainable. 
The most hopeful crystals are KLiS0 4 , NaC10 3 , 
NaLiS0 4 , BeS0 4 • 4H 2 0. 


Production of large crystals of high-melting sub¬ 
stances appears as an arduous and time-consuming 
but hopeful task of pioneering nature. Development 
work and establishment of a manufacturing process 
for the water-soluble crystals could proceed along 
established lines of research and engineering. 





Chapter 26 

SHIELDING FOR DIATHERMY 


A study of methods of reducing radio interference from r-f 
power equipment used for noncommunication purposes. 8 Tests 
proved that, if adequate filtering were provided to prevent 
coupling of the r-f energy to the 60-cycle power lines a single 
shielded room of 16-mesh bronze screening will provide an 
attenuation of approximately 73 db and that adding another 
shield (thus making a double-shielded room) will add another 
73 db. In the final report 1 several types of line filters and 
several methods of constructing shielded rooms are described. 

26.1 the problem 

T he magnitude of the interference caused by dia¬ 
thermy or other noncommunication r-f power 
equipment at any receiver depends on several factors, 
notably the band width covered by the radiation and 
the power radiated, the geographical separation and 
the frequency separation between the source and the 
receiver being jammed, and the susceptibility of the 
receiver. 

The band width of most diathermy equipment is 
quite large, because raw alternating current is applied 
to a self-rectifying oscillator. The use of filtered di¬ 
rect current on the plate of the oscillator appreciably 
reduces the influence of the radiating source by reduc¬ 
ing the band width of the power radiated. However, 
such a scheme adds to the complexity and cost of the 
diathermy apparatus. 

The load circuits into which diathermy apparatus 
works are of two general types, one being made up 
of two large insulated flat metallic plates separated 
by the body of the patient under treatment and the 
other consisting of a coil of a few turns of insulated 
flexible conductor placed near the portion of the 
patient’s anatomy which is to be treated. In either 
case the load system represents a transmission line 
system with leads rather widely separated and with 
rather high voltages across them. Furthermore, the 
leads acting as a transmission line may be an appre¬ 
ciable fraction of a wavelength long and, not being 
terminated in their characteristic impedance, may 
have standing waves upon them. Thus a consid- 

a Project C-31, Contract OEMsr-225, Rensselaer Polytechnic 
Institute. 


erable radiation of energy may take place, since the 
load and the leads connecting the machine to it con¬ 
stitute a fairly good antenna system. 

Eliminating the radiation from the leads or the 
load circuit is not of much use unless the radiation 
from the oscillator itself is eliminated; this implies 
adequate shielding of the whole system, which can 
best be accomplished by working the whole system in 
a shielded room. 

26.2 TESTS CONDUCTED 

In the study of means of preventing troublesome 
interference the several components of a typical oscil¬ 
lator-amplifier equipment were separated and indi¬ 
vidually shielded. 

It was found that the use of balanced circuits such 
as push-pull oscillators or balanced load lines will re¬ 
duce the interference materially. 

The effect of grounding a shielded room was also 
investigated. It was ascertained that if all joints in 
the shield, such as doors and windows, were tight and 
if the filter used in the power line were adequate 
it made no difference in the attenuation obtained 
whether the system were connected to an external 
ground or not. 

It was found that an LC filter in which special 
pains are taken to keep the condenser leads short con¬ 
stitutes an excellent filter, provided it is enclosed in 
an electrically tight box and bonded directly to the 
shielding where power enters the room. Special pains 
must be taken to build and install a filter which will 
allow the full attenuation available to be obtained 
from a double-shielded room. Shielded transformers 
of conventional design were found to be inferior to 
LC filters. 

The final report gives the results of the individual 
efforts made and describes the shielded rooms and 
filters constructed and gives cost figures. The cost of 
reducing interference by shielding was approximately 
$,3.00 per db. 


CONFIDENTIAL 


17S 







Chapter 27 

LOCATING FAULTS IN WIRE LINES 


By a special bridge located at a terminal point, capacitance 
changes between conductors provide means for determining 
the location of an open fault in W110B twisted pair and 
WC548T1 “spiral four” lines. A keyed oscillator located at 
the terminals and a portable amplifier carried along the wires 
provide additional means of locating the open point. 

27.1 THE problem 

T he object of Project C-37 a was to find a method 
of determining the point of occurrence of open 
faults in wire lines usually of the W110B twisted pair, 
and WC548T1 four-wire spiral armored-cable types. 
It was desired that the fault be located first by meas¬ 
urements from a switchboard and then that some 
method be provided so that a lineman might proceed 
along the line with an instrument which would give 
an indication of the exact location of the break. 

Several suggested methods of attack were explored 
but all basically were methods of measuring the capac¬ 
itance between wires or from the open wire to ground. 
The best solution, therefore, appeared to be the accu¬ 
rate measurement of the capacitance of the wire to 
some nearby object. 

27 2 CAPACITANCE MEASUREMENTS 

Crude methods of open-fault locations by means of 
voltmeter readings and a-c readings have existed for 
many years, but their accuracy depends upon weather 
and line conditions. Bridge methods are much more 
accurate, yet the accuracy with which the length of 
a wire to an open fault may be measured depends 
greatly on which of several capacitances is measured. 
Where a wire or pair of wires is suspended above the 
earth but the position with respect to the earth is 
alterable, either by a change in distance or by a change 
in the apparent ground level due to rain or other 
cause, the capacitance value also varies. For this rea¬ 
son, the least variable of the capacitances associated 
with a wire is that between the wire and its mate in a 
pair or the other conductors of a cable. 

To measure this capacitance, however, it is neces¬ 
sary to remove the effects of other capacitances which 


“Project C-37, Contract OEMsr-316,Western Union Telegraph 
Co. Detailed drawings contained in the original project report 1 
showing mechanical layout of these instruments have been 
omitted from this summary. 


would otherwise vitiate the results. This can be done. 

Referring to Figure 1A, a pair of wires, 1 and 2, 
are shown together with their capacitances to each 
other and to the ground. If C\ g is the capacitance of 
1 to ground, C 2g that of 2, and C w the capaci¬ 
tance between wires, and if the wires are connected 
to a capacitance bridge as shown, with wire 2 
grounded, the measurement of capacitance (known 
as C\), will be C = C ig C w ; the grounding of 
wire 2 has removed C 2g from the measurement. If a 
second measurement ( C 2 ), as in Figure IB, is made 
with wire 1 grounded, 




Figure 1 . Several capacitances existing in pair of wires. 


A third measurement C 3 , is now taken, as in Figure 
1 C, with wires 1 and 2 connected together, and 
C 3 = C ig +C 2 g. 

We now have three simultaneous equations in three 
unknowns, so that bv combining 

„ _C 1 + C 2 -C 3 

c w — 2 ’ 

/V 

C w is the actual capacitance between the two wires. 


176 


CONFIDENTIAL 






















PORTABLE LOCATOR 


177 


This value should be recorded in the switchboard rec¬ 
ords as soon as the lines are installed. This value 
should be checked from time to time. 

27 2 1 Errors Due to Resistance 
and Leakage 

An error is introduced in the measurement by the 
effect of the resistance of the wires in series with this 
capacitance but this is eliminated by a modification 
of the bridge whereby S in the bridge arm (Figure 2) 



Figure 2. Capacitance bridge in which S balances out 
series resistance of line. 

balances out the series resistance of the wires. The 
line makes up one arm of the bridge and, since the 
two resistive arms are equal, the balancing arm S 
and C is direct reading. Leakage between wires or be¬ 
tween either wire and ground would also cause a meas¬ 
urement error and for this reason the bridge has been 
further modified as in Figure 3. The method used 



Figure 3. Further modification of elementary capaci¬ 
tance bridge to compensate both wire resistance and 
leakage. 

is first to apply direct current to the bridge, adjusting 
rheostat P for a null on the galvanometer G. This is 
a d-c balance and the result is that the sum of the 
resistance “in circuit” in P plus the total resistance 
of the potentiometer is equal to the sum of the series 
line resistance P s in line 1 plus the leakage resistance 
Pi P of line 1. 

The position of the arm of potentiometer S will not 
affect the d-c balance. Eheostat adjustment is left 


untouched from this point on during the measure¬ 
ment of C 1 ; S and C are alternately adjusted for the 
most complete null. C 2 and C 3 are measured by the 
same procedure, the a-c balance in each case being 
preceded by the d-c balance. The final adjustment of 
the bridge will be such that the resistance across C 
will equal the line leakage resistance P p ; the remain¬ 
der of the resistance in S will equal the line resistance 
The capacitance setting of C is still the line 
capacitance. 

This procedure makes possible the measurement 
of capacitance with any combination of line leakage 
and line resistance within reason, for all of the line 
connections shown in Figure 1. 

27.3 PORTABLE LOCATOR 

The second part of this project was to find means 
by which a lineman might proceed parallel to the line 
containing the open fault and obtain an indication 
of the location of the fault without placing too much 
dependence upon the switchboard measurements and 
calculations made therefrom. This would be particu¬ 
larly applicable in the case of W110B twisted pair, 
where the capacitance change is as great as 5 per cent. 

Bv energizing the line under test with alternating 
current at around 1,000 cycles, an alternating electro¬ 
static field is set up around the wire or pair. This may 
be detected on using a small antenna and a multi¬ 
stage amplifier, battery operated. Under favorable 
conditions, depending upon the shielding effect of 
other wires, the voltage of the test frequency, the 
height of conductor above ground, and noise from 
sources of interference and on open wire, the signal 
has been detected at points 2,000 ft from the line 
using a 50-volt signal to line and an amplifier with a 
sensitivity such that a 160-//,v input gives full output. 

If there were only one wire in the line such an 
arrangement would suffice and it would be necessary 
only to follow the line in a car on which an antenna 
had been attached, going out to the point where the 
signal disappeared. With other wires in close proxim- 
it} r , however, sufficient energy would probably leak 
past the break to give a poor indication of the fault. 

To provide better discrimination, the arrangement 
shown in Figure 4A may be used. The faulty wire and 
a good wire (preferably the second wire of the pair) 
are connected to the secondary terminals of a trans¬ 
former which has its mid-point (secondary) grounded. 
The two wires are thus carrying voltages 180 degrees 
out of phase with each other and, if the electrostatic 


A 


CONFIDENTIAL 



























178 


LOCATING FAULTS IN WIRE LINES 


A 



Figure 4. Switching cycles to make easier location of 
an open fault at position A. 


field at point B is investigated with the portable re¬ 
ceiver, there will he very little signal, since the two 
will cancel each other almost completely. As we go 
through point C to point D, the signal increases, due 
to the fact that only one wire is involved at this point, 
therefore no cancellation results. The transition from 
point B to point D is very distinct. The change can 
first be noticed when the distance from the antenna 
to the break is about twice that from the antenna to 
the wire. While the above method is an improvement 
on the use of a single wire and permits the open fault 
to be located in the presence of other wires, the degree 
of cancellation of the two out-of-phase voltages may 
not be complete enough to avoid confusion. If, how¬ 
ever, the two wires are connected as in Figure 4B, for, 
say, Va second and then connected in phase opposition 
for a like period, we will have a condition where on 
the near side of the fault a signal will he received 
which is strong for one period and weak, if audible 
at all, for the second period. On the far side of the 
break, the signals will he approximately equal in 
strength. 

This is shown in Figure 5. On the near side a 
signal similar to Figure 5A is received in which al¬ 
ternate signals are weak or inaudible, becoming equal 
after the open fault is passed. Thus, a lineman may, 
if necessary, only approach the line at intervals, yet 
he will immediately know from the character of the 
signal he receives whether he has passed the break. 
If it so happens that both wires of a pair are open, 
it is only necessary to use a third wire paralleling the 


two faulty ones as a good wire, the two faulty wires 
being strapped together to form one conductor. The 
switching operation and the source of alternating 
current have been combined in one instrument which 
is equipped with a power supply operating from 110 
volts of alternating current. This instrument, known 
as an oscillator-keyer, contains a small motor-driven 


A 

E AT POINT § 


SMALL SIGNAL 



E AT POINT 0 


FIRST SIGNAL APPROX 
^EQUAL TO SECOND 



2 NO 


1 ST 


t NO 



T 1 


T I 

COMPLETE CYCLE I 

... *•) 


Figure 5. Effect of switching on electrostatic field near 
wire. In A, listener is at point B; in B, listener has 
passed fault and gets equal signals on both of switching 
cycles. 

switch, a voltmeter indicates the power level, and two 
signal lights indicate whether power and motor are 
connected to the power source. 

The portable locator is a multistage battery-oper¬ 
ated amplifier. A small metal wand or an antenna, 
permanently attached to a car, may he plugged into 
a socket on one end of the instrument. The signals are 
noted by watching a rectifier-type meter in the output 
stage of the amplifier or by listening to the output 
of a miniature speaker. 

In operation, the level of gain is raised to a point 
where full deflection of the meter occurs upon the 
reception of the signal. No harm results from excessive 
gain since the meter is protected by the limited power 
output of the driving tube. 

Schematics of the capacitance bridge, the oscillator- 
keyer, and the portable fault locator are shown in 
Figures 6, 7, and 8. 


CONFIDENTIAL 































PORTABLE LOCATOR 


179 


8SJ7 




CONFIDEN 






















































































































































































































180 


LOCATING FAULTS IN WIRE LINES 


V, V, V, 



Figure 8. Schematic of portable fault locater. 


CONFIDENTIAL 





























































































STORAGE BATTERIES FOR COLD CLIMATES 


Research leading to the development of small storage 
batteries giving 40 to 46 per cent of normal, or 80-F, capac¬ 
ities at —40 F. Previous to this development, capacities of 
19 to 25 per cent had been obtained. 

28.1 STATE OF THE ART 

T he research covered by Project C-40 a was con¬ 
sidered for an appreciable period of time before 
the entry of the United States into active warfare. 
Studies of the conditions under which combatant 
forces were engaged, along with the rapid develop¬ 
ment of various types of highly specialized apparatus 
requiring self-contained power supplies, brought forth 
several important facts. 

1. The performance of the primary cell (dry cell), 
although adequate for operations at normal tempera¬ 
tures, was entirely inadequate at low temperatures. 

2. The performance of the lead-acid secondary cell, 
although operative at temperatures much lower than 
those at which dry cells failed to operate, was not 
generally satisfactory. 

3. The secondary or storage-battery cell with ade¬ 
quate capacity to meet the operating conditions re¬ 
quired at low temperatures and the fixed dimensions 
of the apparatus with which it would have to be used 
was not suitable because of its physical size. 

4. Studies of low operating temperatures as re¬ 
corded by the U. S. Weather Bureau at Fairbanks, 
Alaska, indicated that although extreme temperatures 
of —60 F might be met infrequently and for short 
periods of time, the temperatures during the winter 
months of November to March ranged generally be¬ 
tween —30 and —40 F. Adequate capacity for opera¬ 
tion at —40 F would not only be desirable but entirely 
necessary. 

28.2 THE research problem 

The research problem resolved itself into two dis¬ 
tinct operations or projects: 

1. The investigation of the possibilities of produc¬ 
ing small nonspill or leakproof storage batteries, sim¬ 
ilar in shape and dimension to the primary or dry 

“Project C-40, Contract OEMsr-420, Willard Storage Battery 
Co. 


cell, which would have satisfactory performance at a 
temperature of —40 F. 

2. The refinement of design and the investigation 
of manufacturing methods leading to the production 
of satisfactory storage-battery cells which would meet 
the prevailing operation conditions. 

28,2,1 General Plan of Attack 

As the storage battery is a chemical device, the gen¬ 
eral plan of attack had to be one wherein the func¬ 
tions of each component part were considered and 
studied. The investigation of the individual compo¬ 
nent parts Avas carried out in this order: 

Insulation. To obtain maximum capacity with a 
minimum of free electrolyte it was necessary to in¬ 
vestigate new types of insulation. To accomplish this 
it was necessary to evaluate the electrolyte-retention 
values of various types of packed insulation as well as 
performance at normal 80 F and at —40 F. Materials 
such as glass wool, cellulose paper, Latex sponge, 
pressed wood pulp, and Fibrite were examined. 

Grid Thickness. By the use of plates pasted Avith 
the same positive and negative mixes, the determina¬ 
tion of plate combinations to obtain the best ratios be¬ 
tween warm and cold capacities and the best plate 
thickness, surface, and contour was determined. Plate 
thicknesses of 0.04, 0.07 and 0.093 in. were tested. 
It was determined that the use of multiple thin plates 
was superior to the use of fewer thick plates. 

Mixes. Having established the best experimental 
plate designs, various positive and negative mixes 
were investigated. This was carried out in cells haA’- 
ing no insulation. Mixes superior to standard practice 
were developed. 

Insulation. Tests carried out by the addition of 
various types of insulation to the cells built with the 
best mixes showed that five of the insulating mate¬ 
rials used would not seriously affect the capacity at 
high or low temperatures. 

Following these tests, work was done to determine 
the best grid design on the basis of conductivity and 
manufacturing facility, and on the physical and chem¬ 
ical properties of various container materials as well 
as their ease of manufacture and availability. 


CONFIDENTIAL 


181 







182 


STORAGE BATTERIES FOR COLD CLIMATES 


283 RESULTS ATTAINED 

Final testing resulted in combinations where capac¬ 
ities of 40 to 46 per cent of the normal or 80-F 
capacities were obtained at —40 F. Previous to this 
development capacities of 19 to 25 per cent had been 
obtained. Considering the fact that primary cells are 
practically inoperative at —40 F or adjacent tem¬ 
peratures, and that the available capacity has been 
increased as indicated, the conclusion was reached 
that it was possible to develop the chemical and physi¬ 
cal properties of small storage batteries to the point 
where satisfactory capabilities may be obtained at 
cold temperatures. 

The design and development of production models 
was taken over by the Signal Corps Development 
Laboratories. Production models of the D and No. 6 
sizes of primary cells and the addition of a new type 
DD or double-D cell were made under this program. 

28.4 DEVELOPMENT OF A 

ONE-CYCLE CELL 

After the initiation of the research described above, 
urgent demands were made by the several agencies 
for the exploration of single-cycle (one-shot) storage 
cells or batteries to replace the basic cell used in B 
batteries. 

The exploration work accomplished under the early 
part of the project had led to the development of a 
satisfactory cell for use at —40 F. The element of this 


cell was made up of flat plates and took the conven¬ 
tional form of a rectangular prism. 

To be interchangeable with the dry cell, the flat- 
plate element would have to be installed in a cylin¬ 
drical container. Such installation would be expensive 
from the standpoint of the volume required, since a 
rectangular element in a cylinder would not use the 
volume efficiently. Capacity would be sacrificed. 

Much work was done on elements of various shapes. 
For example, certain rolled elements, especially of the 
“herringbone design,” indicated good possibilities, in 
fact better than any other investigated especially for 
installation in cylindrical containers. However, new 
and special operating machinery would be required to 
put this element into production. For this reason it 
was thought advisable not to recommend production 
of this type but to concentrate upon flat-plate cells in 
the rest of the work. 

Data were collected and procedures developed which 
led to the production of a line of miniature sizes of 
lead-acid flat-plate type cells for nonspill, nonslop 
construction for either one-cycle or for repeated cy¬ 
cling service. The assembly of such cells into simple 
packs could be accomplished by merely assembling a 
pack of the individual cells or by assembling the ele¬ 
ments into monobloc plastic containers, as was done 
in the Signal Corps 20-volt Type ER-B-20. 

The utility of the single-cell unit lay in quick as¬ 
sembly into simple packs for nonstandard usage or 
where out-of-the-ordinary applications made it prefer¬ 
able to other types of cells. 


* CONFIDENTIAL 






Chapter 29 

AN ELECTRICAL CANCELLATION AND INDICATING SYSTEM 


A system which will cancel out any recurrent complex wave 
and give an alarm indication automatically when new wave 
components appear. Useful in monitoring systems, since, by 
establishing an alarm threshold, an alarm can be given when 
new signal or noise elements occur. 


29.1 STATEMENT OF PROBLEM 


T he problem as presented in NDRC Project C-67 a 
was to design an electrical system which would 
provide: 

1. An adjustable complex voltage waveform which 
could be made very nearly identical to any repetitive 
wave that may be produced by an unknown source. 

2. Automatic comparison of the known and un¬ 
known waves in such a manner that their resultant 
will be zero until a change occurs in the unknown 
wave shape. 

3. An aural or visual alarm to indicate such change 
in the unknown wave regardless of the polarity of this 
change. 

29.2 SOLUTION TO THE PROBLEM 


After considering several possible methods described 
in the final report, 1 the following scheme was devel¬ 
oped. Beginning with a repeating square wave of 
duration $ and period r, of variable amplitude, any 
wave of the same fundamental period t can be ap¬ 
proximated by a series of pulses of equal duration 
and period, but so time shifted that the start of each 
coincides with the end of the preceding pulse (a time 
delay of 8 seconds). (See Figure 1.) 

Considering a composite wave made up of m such 
pulses with their durations adjusted to completely 
fill the period t, i.e., 8 = r/m, the amplitude coeffi¬ 
cient A na for any one pulse a becomes 



where a = 1, 2, 3, 4--- {m — 1), m, 

m = total number of pulses in the repeating 
wave, and 


“Project C-67, Contract OEMsr-748, RCA Laboratories. Only 
basic elements of the project are given in this summary. The 
analytical portions of the final report with necessary diagrams 
and circuits will be found in the report and on microfilm. 


n — order of the harmonic for a particular 
pulse. 

These pulses have fixed durations and fixed phase 
displacements. The only variable for a particular pulse 
is its amplitude E. Moreover, each adjustment is in¬ 
dependent and need be set only once. The cancella- 



._Jl 


INDIVIDUAL WAVES 



n 


Figure 1. Composite wave made up of individual ele¬ 
ments of varying heights, each occuring immediately 
after one preceding. 

tion procedure can thus be organized in any simple 
manner. For example, pulse No. 1 can be adjusted in 
amplitude to match the unknown wave for its portion 
of the total cycle, pulse No. 2 for its portion, etc. 
(See Figure 2.) 

Figure 2 also illustrates the method of obtaining 
function 1. The polarity of the local or known wave 
is inverted and mixed with the unknown wave in a 
comparator circuit the output of which is proportional 
to the difference between the two waves. Actually, this 
output is a measure of the degree of approximation 
achieved and will improve with the number of pulses 
composing the synthesized wave. Using a Fourier 
schedule as a guide, it was decided to utilize 40 rec¬ 
tangular pulses in the experimental work (m — 40). 
With this value of m, Figure 2 illustrates only a por¬ 
tion of the wave period. 


CONFIDENTIAL 


183 































184 


AN ELECTRICAL CANCELLATION AND INDICATING SYSTEM 


/ 


ORIGINAL WAVE 



RESULTANT WAVE 
AFTER BALANCING 




DURING THIS PORTION OF THE 



DURING THIS PORTION OF THE 

CYCLE THE ORIGINAL WAVE HAS 



CYCLE THE RECTANGULAR WAVES 

BEEN COMPARED TO ITS RECT- 



ARE STILL AT ZERO — SO NO 

ANGULAR APPROXIMATION AND 



CANCELLATION OCCURS 

THE RESULTANT SHOWN OBTAINED 





Figure 2. Partial cancellation of wave by opposing to it 
a series of rectangular waves. 


In general, the rectangular pulse method of approxi¬ 
mation will leave a triangular residual of noise as 
shown. This noise can be considerably reduced by 
shunting the synthesized electrical wave with an 
appropriate capacitance as illustrated in Figure 3. 
The degree of approximation is thus improved cor¬ 
respondingly. 






RESULTANT WAVE 
AFTER BALANCING 


H 


4 5 6 7 8 9 10 II 






DURING THIS PORTION OF THE CYCLE 
THE ORIGINAL WAVE HAS BEEN 
COMPARED TO ITS RECTANGULAR 
APPROXIMATION AND THE RESULT¬ 
ANT SHOWN OBTAINED 


Figure 3. Use of capacitance to further cancelling effect. 


Having determined rn, the number of pulses re¬ 
quired in the local wave, consideration was next given 
to the method of their generation. One proposal was 
to use 40 trigger circuits so arranged that each would 
trip immediately following the preceding one. Such 
an arrangement is straightforward from a design 
standpoint, but the number of tube circuits involved 
is considerable. Consequently, this method was dis¬ 
carded in favor of a commutator and brush assembly 
requiring very little additional equipment. 


Preliminary tests of the commutator brush-arm 
assembly as a complex-wave generator were carried out 
using a 42-segment face plate originally built for tele¬ 
graph work. In the initial setup, each of 40 of these 
segments was connected to the movable arm of one 
of 40 potentiometers. The potentiometers in turn had 
their respective high-impedance terminals connected 
in parallel. The remaining two segments were used 
to provide external synchronizing pulses to a standard 
oscilloscope. A schematic diagram of the experimen¬ 
tal test arrangement is shown in Figure 4. The an- 



Figure 4. Elements of cancelling system employing 
rotating commutator to produce cancelling waves. 


gular velocity of the rotating brush arm in this case 
was approximately 15 cycles per second, thus requir¬ 
ing very good low-frequency response in the succeed¬ 
ing amplifiers. This angular speed could not be in¬ 
creased materially without causing noticeable splits 
between the rectangular pulses unless great care was 
taken in the adjustment of the brush position. It was 
determined experimentally that a better method was 
to tie diagonally opposite segments in parallel and 
provide twice the number of segments. The fundamen¬ 
tal frequency would then be doubled for the same 
angular velocity since the local wave would be repeated 
twice for each rotation of the brush. In the same way, 
the frequency could be tripled with 3 m segments, etc. 
The results of these tests indicated conclusively that 
the commutator method was the simplest and most 
satisfactory type of complex-wave generator. 

On the basis of these experimental tests, it was de- 


C 0 NFI DENT 1A 
















































































































PROPOSED ELECTRICAL CANCELLATION AND INDICATING SYSTEM 


185 


cided that the complete cancellation and indicating 
system should include: 

1. A complex-wave generator of the commutator 
type. 

2. An electronic mixer with linear input-output 
characteristics. 

3. An oscilloscope to show the accuracy of approxi¬ 
mation to the unknown wave. 

4. An alarm system. 

A block diagram of this system and its interconnec¬ 
tions is shown in Figure 5. 



Figure 5. Block diagram of system. 


29.3 PROPOSED ELECTRICAL 

CANCELLATION AND INDICATING 
SYSTEM 

To realize fully the advantages of the system shown 
in Figure 5, it was deemed necessary to design and 
experimentally verify each of the major circuits. More¬ 
over, this would permit an accurate evaluation of the 
overall system from a practical standpoint. These cir¬ 
cuits, as finally developed, are described below. 


29-3 ' 1 Complex- Wave Generator 

The multisegment commutator was constructed in 
accordance with standard telegraph practice. Four 
concentric rings are provided in the arrangement, of 
which two are solid and two are segmented. The out¬ 
side ring (No. 1) is divided into 84 equilength seg¬ 
ments of which the first 80 are paired by interconnect¬ 
ing diametrically opposite segments. This was done 
to double the fundamental output frequency. Thus 
the complex-wave output, made up of 40 successive 
rectangular pulses, has a fundamental frequency of 
30 cycles per second even though the brush actually 
rotates at 15 cycles per second. The angular speed of 
the brush is derived from an 1,800 rpm synchronous 
motor operating through a two-to-one step-down gear 
train. The remaining 4 segments (41, 42, 43, and 44) 
are connected to segment 40. During the time the 
brush is passing over these segments, a second brush 
on the same arm makes contact with the isolated seg¬ 
ments shown on ring (No. 2) (see Figure (>). These 
segments form part of the sawtooth generator which, 
by periodically discharging capacitor C 17 produces 
the sawtooth voltage used for oscilloscope deflection 
and external synchronizing purposes. 

Mixer Stage. To utilize the cancellation device in 
various applications, it was necessary to insure linear 
operation throughout. The actual circuit is given in 
Figure 6. Since tubes Y 1 and V 2 have nonlinear 
input-output characteristics, one must resort to graph¬ 
ical analysis for an exact solution. 

Because of resistance elements common to both 
halves of V 1 (Figure 6), the mixer stage of the sys¬ 
tem, it was noted that the d-c operating conditions of 
the right-hand triode changed as the average grid po¬ 
tential applied to the left-hand triode varied. The ques¬ 
tion arose whether these d-c changes would adversely 
affect the mixer linearity over its operating range. 

To determine this effect, it was necessary to obtain 
the magnitude of the shift in the operating point of 
the right-hand triode, as the average potential applied 
to the grid of the left-hand triode varied over its max¬ 
imum range. Once this had been determined, assum¬ 
ing the input versus output voltage characteristics 
for the right-hand triode were drawn for the two ex¬ 
treme operating points, the effect of the shifting 
operating point would be known. 

This problem was approached graphically, Figure 
7 showing that linearity does exist in the mixer stage. b 

b The analysis of this circuit with the necessary voltage-cur¬ 
rent characteristics will be found in the final report. 1 


CONFIDENTIAL 
































'86 


AN ELECTRICAL CANCELLATION AND INDICATING SYSTEM 


A complex WAVE GENERATOR C AMPLIFIER SCOPE 



II 


CONFIDENTIAL 


>0- 





































































































































































































































PROPOSED ELECTRICAL CANCELLATION AND INDICATING SYSTEM 


187 


29 3 2 Monitor Oscilloscope 

Throughout the design of the monitor oscillo¬ 
scope, conventional television technique has been fol¬ 
lowed. All amplifiers have been carefully compensated 
for low frequencies to insure accurate reproduction of 
the complex wave. Linear tubes were selected and 
their characteristics further improved by adequate 
use of degeneration. 




Figure 7. Operation of 6SN7 as mixer showing linearity. 

The sawtooth voltage used for horizontal oscillo¬ 
scope deflection and for synchronizing external equip¬ 
ment is generated in a very simple manner. Capacitor 
C 17 (Figure 6) is charged from the 300-volt supply 
through resistance 7?, i9 and periodically discharged 
through commutator rings No. 2 and No. 4. The time 
during which the brush is passing over segments 41 
and 42 has been assigned for synchronizing purposes. 
Consideration of Figure 6 will show that appropriate 
segments are located on commutator ring No. 2 to 
achieve this discharge in the allotted time. It will also 
be noted that C 17 is prevented from recharging until 
the start of the first rectangular pulse. This is neces¬ 
sary since, if C 17 was permitted to recharge imme¬ 


diately, a part of the synchronizing time would appear 
on the oscilloscope screen. The waveform of the volt¬ 
age developed across C 17 is plotted in Figure 8. 



COMMUTATOR SEGMENT NUMBER 

Figure 8. Sawtooth voltage wave developed across Cn 
charging through R 69 from the 300-v supply and dis¬ 
charging through the commutator segments. 

29.3.3 Alarm and Indicating Arrangement 

In a push-pull oscilloscope deflection arrangement, 
one of the deflection plates must go positive with re¬ 
spect to the centering potential for either an upward 
or a downward motion of the oscilloscope trace. Con¬ 
sequently, by direct coupling the grids of a pair of 
thyratrons to the vertical plates of the oscilloscope, 
one thyratron can be biased to ignite on an upward 
deflection of the electron beam and the other on a 
downward deflection. 

The flow of plate current that is initiated when 
either or both of the thyratrons Th 1 and Tlt 2 are 
ignited energizes a relay as shown in Figure 6. This 
relay, in turn, can be used to operate any type of 
alarm desired. Since the grid of a thyratron loses con¬ 
trol over the plate current after the tube is conducting, 
the relay will remain in its alarm position until an 
attendant opens either switch *SW 3 or *STT 4 (see 
Figure 6). This action interrupts the flow of plate 
current through the thyratrons, thus de-energizing 
the relay. 

To provide an indication of the direction of an un¬ 
balance between the two complex waves, an inherent 
characteristic of the thyratrons has been employed. 
When the thyratron is ignited, its grid endeavors to 
assume a potential somewhere between the potentials 
of its cathode and plate. Consequently, since the im¬ 
pedance of the deflection circuit is high, the conduct¬ 
ing thyratron grid changes the d-c level of the par¬ 
ticular deflection plate to which it is connected. This 
action changes the centering of the oscilloscope in the 
direction of the original unbalance between the two 
complex waves. The shift in oscilloscope centering, 


CONFIDENTS 





























188 


AN ELECTRICAL CANCELLATION AND INDICATING SYSTEM 


like the relay closure, remains until an attendant opens 
either switch SW 3 or N1T 4 . 


29 4 APPLICATIONS OF THE SYSTEM 

As the development work advanced, it became evi¬ 
dent that modified forms of the basic cancellation sys¬ 
tem could be used in several applications other than 
those originally proposed. 

294 1 Electrical Cancellation 

This application utilizes the four functional units 
shown in Figure 9. It is assumed that an unknown 
wave of repetition frequency equal to the fundamental 
frequency of the complex-wave generator is available 
in current or voltage form and that it is desired to 
automatically excite an alarm system when changes 
occur in the unknown wave. 



Figure 9. Drawing of possible cathode-ray screen pat¬ 
tern showing how alarm might be given when wave not 
cancelled by apparatus described makes its appearance. 


The known wave from the commutator assembly and 
the unknown wave are combined in phase opposition 
at the mixer as previously described. A monitor oscil¬ 
loscope at the mixer output then provides a sensitive 
measurement of the degree of approximation obtained 
after the potentiometers and their combined shunt 
capacitance have been properly set. 


The alarm system in this case is essentially a tube- 
operated relay which, when energized, closes a power 
switch to an ordinary doorbell. By suitably biasing 
the tube below cutoff, the mixer-output noise residual 
can he rendered ineffective insofar as the tube plate 
current is concerned. An alarm threshold is thus es¬ 
tablished. If a change occurs in the unknown wave 
regardless of polarity, the mixer output will increase 
in direct correspondence. The manual setting of the 
alarm threshold determines the amount of change 
required to operate the alarm. 

A visual illustration of the operation is best ob¬ 
tained on the oscilloscope. For example, with no local 
wave, the unknown wave only will show on the face 
of the tube. As potentiometers 1 to 40 are progressive¬ 
ly adjusted, the unknown wave is sliced away until 
only a small noise residual remains. Suppose now we 
draw a hypothetical line just above the noise. This 
line corresponds to the alarm threshold. As an alter¬ 
native, we might consider the average value of the 
noise to be the alarm threshold and then displace the 
noise downward. In either case, the effect is the same. 

2942 Panoramic Reception 

The electrical cancellation and indicating system 
can be ideally applied to panoramic reception. The 
functional units of Figure 5 also provide a sawtooth 
wave of the same fundamental frequency as the local 
wave. If this wave is used to sweep the heterodyne 
oscillator frequency of the panoramic receiver, the 
output of this receiver can he cancelled out in iden¬ 
tically the same manner as described above. Actually, 
if the panoramic receiver sweeps over a relatively 
wide spectrum with respect to its effective i-f pass band, 
the receiver output will consist of short pulses, each 
corresponding to a received r-f carrier. These pulses 
may require only a few segments from the local 
generator to displace them below the alarm threshold. 

One advantage of this method is that on and off 
keying of known carriers does not interfere with the 
normal cancellation principle. This is evident, since 
proper adjustment of the alarm threshold when the 
unknown carrier is on will merely leave a downward 
displacement in the line when the unknown carrier 
goes off. In the same way, fading of the known signal 
does not actuate the alarm. A further advantage of 
this method is that no portion of the received fre¬ 
quency spectrum is blocked off. For example, if a 
known carrier is displaced downward and an unknown 
carrier of the same frequency starts up, the two will 


CONFIDENTIAL 






















































































































































































APPLICATIONS OF THE SYSTEM 


189 


beat together giving a received signal of maximum 
amplitude equal to the sum of the two signals. This 
peak amplitude will exceed the alarm threshold, in 
the practical case, so long as the amplitude of the sec¬ 
ond received carrier is greater than the upward dis¬ 
placement of the alarm threshold with respect to the 
normal horizontal axis. 

It is thus seen that the cancellation and indicating 
system when used in conjunction with a panoramic 
adapter or when constructed as an integral part of 
a panoramic receiver, provides completely automatic 
operation after the existing known signals have been 
cancelled out. One might therefore expect that several 
such receiver systems could he monitored by one ob¬ 
server without severe optical strain, since he would 
use the oscilloscope only during the initial setting-up 
procedure and during those times when the alarm 
indicated the appearance of a foreign signal. 

294 3 Coding Signal Generator 

The complex-wave generator portion of Figure 5 
can duplicate any wave shape of fixed repetition period. 
It is, therefore, an ideal coding signal generator. In 
a practical application, the 40 amplitude-control po¬ 
tentiometers might be ten-step voltage dividers num¬ 
bered one to ten. The code designation for a particular 
wave would then consist of 40 consecutive numbers 
indicating the individual settings of tlie 40 voltage 


dividers. The total number of possible combinations, 
that is, the variety of coding waves available, is given 
by the relation 

400! 

r — _ 

" 10! 390!’ 

or approximately 

C n = 10' 9 . 

These combinations can be further complicated by 
adding various shunting capacitances across the out¬ 
put of the complex-wave generator. 

A practical adaptation of this coding device for 
two-way operation is indicated in Figure 10. For ex¬ 
ample, let F c (t) be the coding wave and F s (t) be 
the signal intelligence to be transmitted. These waves 
may be scrambled in any desired manner in their 
common mixer circuit. If we assume the mixer func¬ 
tions as a mathematical multiplier, the coded wave 
will be F c (t ) X F a (t). This wave arrives at the re¬ 
ceiver where an identical coding generator has its 
output inverted and multiplied into the incoming wave 
giving F c {t) X F s (t) X 1 /F c (t) = F s (t). It will 
be understood that the complex-wave generators at 
each end must be running in time and phase synchro¬ 
nism. The reverse transmission operates in exactly 
the same manner. However, if the coding wave at the 
original transmission point is to be satisfactory for 
decoding as well, it cannot lead the incoming scram- 


XMTG FACSIMILE 
SCANNER 


UHF 


RECEIVING FACSIMILE 
SCANNER 



Figure 10. Use of equipment as coding signal generator for radio circuit. 
































































































190 


AN ELECTRICAL CANCELLATION AND INDICATING SYSTEM 


bled signal by more than a few degrees. This means 
the transmission time around the radio loop must be 
small compared to the period of the coding wave. The 
arrangement of Figure 10 would therefore be useful 
for short-distance circuits only. Long-distance two- 
way operation could utilize an additional segmented 
ring and brush arm on the transmitting commutator. 
Proper phase adjustment could then be obtained by 
angular displacement of the brush arm. 

2944 Electrical Network Analysis 

The introduction of radar equipment into the Mil¬ 
itary Services has necessitated rather extensive train¬ 
ing programs for the operating and maintenance 
staffs. In particular, it is necessary that the person¬ 
nel obtain a good knowledge of electrical network 
analysis, both steady-state and transient, in the short¬ 
est possible time. The instructor can either teach the 
required mathematical background including differ¬ 
ential equations, the Fourier series, and the Laplacian 
transform as applied to networks, or teach the funda¬ 
mentals of network theory and apply them heuristi- 
cally to deduce the output response for applied force 
functions. The former method is essentially equivalent 
to the requirements for an engineering degree and 
little can be done to short-circuit the present three- 
year minimum time required. In this connection, it is 
believed that a modification of the complex-wave gen¬ 
erator can serve as a very useful tool. 

Suppose we incorporate in one unit a complex-wave 
generator, two oscilloscopes, and an assortment of 
ladder networks. The wave generator can furnish a 
large variety of waveforms to the inputs of the several 
networks and be observed on one of the oscilloscopes. 


Similarly, the output response of the networks can be 
studied individually on the second oscilloscope by ap¬ 
propriate switching. For classroom instruction, the 
oscilloscope faces should have at least 9-in. diameters 
and might include transparent scales to form a rec¬ 
tangular coordinate system. 

A device of this type can be used to verify existing 
mathematical solutions of network responses. In ad¬ 
dition, it can derive approximate responses of very 
complex networks for which no exact solutions have 
as yet been obtained. Moreover, it would appear that 
such a device would be a very useful tool for class¬ 
room instruction, particularly when it is necessary to 
reduce the instruction time to an absolute minimum. 

29.4.5 Multiple Station Monitoring 

There are certain minor applications of the device 
that might have merit in specific cases. If it is de¬ 
sired to monitor the frequency of several transmitters 
simultaneously, one might use the arrangement de¬ 
scribed above under Panoramic Reception. If the 
frequency of one carrier varied, it would be displaced 
on the time axis of the panoramic scope and rise out 
of its declivity, thus ringing the alarm. If it is de¬ 
sired to monitor the transmissions of several trans¬ 
mitters simultaneously, one might have several re¬ 
ceivers with their outputs connected to the individual 
segments of a commutator. The rotating brush arm 
would then serve as a scanner with its output con¬ 
nected to a page-type facsimile recorder. By synchro¬ 
nizing the drum speed of the recorder with the rota¬ 
tion of the brush arm, each transmitter signal would 
be recorded as a horizontal track similar to variable- 
density movie records. 


d; CONFIDENTIAL 






Chapter 30 

LAYING FIELD WIRE BY AIRPLANE 


Laboratory investigation and field tests leading to methods 
of packing field wire in a suitable form for paying out from 
an airplane in flight, and to techniques for laying the wire at 
speeds up to 200 mph. 

301 HISTORY OF THE DEVELOPMENT 

he interest of NDRC in air-laid wire originated 
in a request for a study of listening devices to de¬ 
tect enemy movements in jungle areas. It was sug¬ 
gested to NDRC in July and August 1942 that it 
might be possible to carry listening devices over into 
enemy territory by planes. It was thought that planes 
could string numerous long camouflaged wire sections 
over treetops and ahead of advance positions. Small 
microphones at the wire ends would enable listeners 
to detect the position and extent of enemy movement. 
It was also believed that the ends could be laid with 
sufficient accuracy to permit sound ranging on gun 
positions. 

This early interest later extended to possibilities of 
laying wire from planes for special communication 
purposes, with emphasis largely on jungle terrain and 
with the hope of materially reducing the hazard of 
direction finding, jamming, and interception. 

Accordingly Project C-72 a was initiated to investi¬ 
gate the possibilities of laying wire by air. 

After a preliminary survey the problem divided it¬ 
self into several lines of investigation: selection of the 
wire, behavior of the wire during uncoiling and pas¬ 
sage to the ground, determination of the best form of 
package for the wire, technique of laying wire, and 
winding machines. The final report 1 on Project C-72 
gives the results of these studies. 

After considerable laboratory work, a few trial 
flights were made at Fort Knox in April 1943. The 
results were so encouraging that the Army gave wide 
circulation to the NDRC report. This report eventu¬ 
ally reached the China-Burma-India theater where 

“Project C-72, Contract OEMsr-879, Western Electric Co., Inc. 


conventional methods of laying wire were not only 
inadequate but fraught with danger. In January 1944, 
Army Air Forces authorities in the CBI theater cabled 
a request for a continuation of the development work 
which had stopped because of termination of the proj¬ 
ect. Shortly after this the Air Technical Service Com¬ 
mand [ATSC] requested that the work be resumed 
ou a rush basis, and a contract covering the comple¬ 
tion of the work was negotiated directly between the 
Western Electric Company, Inc., and ATSC. 

302 RESULTS OBTAINED 

The following summary includes the work accom¬ 
plished under the NDRC contracts as well as the di¬ 
rect contract mentioned above. 

An outdoor laboratory was set up at Murray Hill, 
New Jersey, including a machine capable of “pulling” 
wire from packs and coils at speeds up to 200 mph. 
Many fundamental tests were made at this location 
and these were followed by more than 200 flight tests 
at the Army Air Base, Fort Dix, New Jersey. 

Three methods of packing wire were investigated: 
the carpet pack, the solenoid coil, and the crisscross 
coil equipped with center guides. The last system 
proved most practicable and was eventually standard¬ 
ized. This system was found capable of laying WD-1 
wire in 16-mile lengths at speeds up to and exceeding 
150 mph. It was found possible to lay Canadian D-8 
wire and standard W-110B in 16-mile lengths at some¬ 
what slower speeds. Following the experimental flights 
at Fort Dix, New Jersey, a demonstration run was 
made for the Army with complete success over a 
rugged 16-mile course in the Great Smoky Mountains. 

Ten machines capable of winding field wire in criss¬ 
cross coils were assembled and turned over to the Air 
Forces. Also, five complete equipments for laying 
wire from airplanes were furnished. Included with 
the latter was a manual covering the entire technique 
of winding and laying wire at high speeds. 


CONFIDENTIAL 


191 





Chapter 31 

FLOATING INSULATED WIRE 


A lightweight wire of 700-yd maximum length was desired 
which would have sufficient buoyancy to float and be strong 
enough to pull a rubber raft. tt In preliminary experiments made 
in the theater, a wire taped to a 1^-in. rope was used. Coral 
cut the rope and there was a limit to the length that could 
be used because of the weight. 

3i i SUMMARY OF RESULTS 

ince the specific gravity of polyethylene (0.92) 
is sufficiently below that of sea water to float, it 
was thought that a floating wire of considerable 
strength could be obtained with reasonable dimensions. 
The tensile strength of polyethylene is relatively low, 
about 1,000 lb per sq in. for an elongation of about 
25 per cent. Accordingly, an efficient design of float¬ 
ing wire would be one in which the polyethylene serves 
to provide buoyancy and insulation and the metallic 
conductor in the core provides strength. 

The simplest design to produce is one consisting 
of a single conductor of stranded steel encased in the 
insulating material. This assumes a sea-water return 
for the electric circuit. 

A smaller wire of equal strength could be provided 
by using Fiberglas continuous-filament tying cords 
for strength and a small copper wire for conductivity. 
The tensile strength of Fiberglas cords is approxi¬ 
mately 75,000 to 100,000 lb per sq in. with an elonga¬ 
tion that is low. The specific gravity is 1.5. Table 1 
below gives a comparison of the two proposed methods 
of solving the problem. 

“Project 13-104; Contract OEMsr-1413, Western Electric Co., 
Inc. Several matters were considered under this project as 
noted in the bibliography. Aside from the subject of this sum¬ 
mary, the other matters were largely discussions of problems. 
Division 13 undertook no actual work on these subjects. 


Table 1. Comparison of steel core and Fiberglas- 
copper core floating wires. 


Breaking 

strength 

(lb) 

Outside diameter (in.) 

Weight per 1,000 ft (lb) 

Steel core 

Fiberglas- 
copper core 

Steel core 

Fiberglas- 
copper core 

300 

0.31 

0.23 

33 

18 

500 

0.40 

0.27 

56 

25 

1,000 

0.57 

0.36 

111 

44 

2,000 

0.81 

0.49 

222 

82 


Other materials, such as nylon and Fortisan, were 
considered and their possibilities for this application 
discussed. 

Provisions for protecting the wire against coral 
reefs, the effects of ocean currents, and handling 
properties are discussed in the final report. 1 

3i-2 CONCLUSIONS 

The most promising design would employ a core 
of Fiberglas or other lightweight high tensile strength 
material and one or more small copper conductors 
and polyethylene insulation of sufficient thickness to 
float the wire. A size providing a tensile strength of 
500 lb would have a diameter slightly over 14 in. and 
a weight of about 25 lb per 1,000 ft. Greater strength 
could be provided by using a twisted pair of such 
wires. 

The suggested design was speculative. It was ex¬ 
pected that a year would be required to obtain wire 
in quantity for testing under field conditions, readying 
for production, and so forth. The matter was referred 
to the Services for such direct action as they might 
wish to make. 


192 


CONFIDENTIAL 

















Chapter 32 

NOISE INVESTIGATIONS 


Several projects under Division 13 were concerned with the 
problem of noise from power-generating machinery. The ques¬ 
tion of audible noise from mobile field generators was covered 
by an investigation of methods of quieting such apparatus con¬ 
ducted as part of the broad work carried out in Project C-79. 1 
Two other projects, one dealing with noise measuring equip¬ 
ment, are being continued by the Services after the NDRC 
contracts were cancelled in October 1945. 


32.1 NOISE IN AIRCRAFT ELECTRICAL 
MACHINERY 

U nder this project 3, experimental studies were to 
be conducted to determine the factors affecting 
the generation of r-f voltage in 28-volt d-c aircraft 
rotating electric machinery and to obtain informa¬ 
tion leading to the design of such machinery so that 
the radio noise produced would be minimized and 
would lead to the simplification of filtering such 
machinery. 

During the period of the project, 2 30 pieces of 
equipment obtained from Wright Field and from the 
Bureau of Aeronautics were examined. They included 
motors, dynamotors, inverters, etc., and the measure¬ 
ments included conducted noise and noise field in¬ 
tensity. 3 

At the end of the project, the sample equipment 
had been tested and charts prepared (included in the 
terminal report) to show the noise levels measured. 
Analysis had been carried to the point where some 
of the features associated with radio noise were evalu¬ 
ated and procedures for investigating other features 
were indicated. 

In the tests, conducted noise on the 28-volt circuit 
ranged from 1 to 12,000 fix. Some of the samples had 
capacitors or filters. The “50 per cent curve,” plotted 
from the noise values that were equalled or exceeded 
in 50 per cent of the tests at the respective frequencies, 
ranged from 25 to 750 fix, a ratio of 1 to 30. 

Conducted noise on the output of dynamotors with 
resistance load varied directly as the machine voltage 
and inversely as the frequency at which the noise was 
measured. In different machines at the same voltage, 
the noise varied inversely as the number of commu¬ 
tator bars between brushes. 

“Project 13-117, Contract No. OEMsr-1475, General Electric 
Co. 


The conducted noise in fix on the dynamotor out¬ 
put circuit was approximately equal to 
1300 X d-c volts 

(Bars per brush) (Frequency in me) 

The results pointed to the fact that most of the 
radio noise produced by the samples was caused by 
the passage of the brushes from bar to bar of the com¬ 
mutator. 

32 2 NOISE MEASUREMENT 

The subject of this project 13 was “to conduct miscel¬ 
laneous studies and investigations involving radio 
interference phenomena including development of 
standards for measurements and measurement tech¬ 
niques, comparison of various types of noise meters 
and construction of a standard noise meter.” This 
work will be carried forward under contract with the 
Bureau of Ships. 

The need for investigation of radio noise measure¬ 
ments arises from the fact that such measurements 
using noise meters now available show anomalous dis¬ 
crepancies. It is reported that various instruments 
having been calibrated with sine wave signals, read 
differently to as great an extent as 1,000 to 1 when 
they are presumably the lield intensity from the same 
noise source. With the same meter very different re- 
sidts are frequently obtained when readings are taken 
on two frequency bands in the region of their overlap. 
The degree of discrepancy is reported to be different 
for different noise sources. The discrepancy is greater 
when radiated noise is being observed than when di¬ 
rect conducted noise is being observed. 

Under the project, representative noise meters were 
to be examined critically in a screened room. At the 
termination of the contract, the noise meters had not 
been received from the manufacturers, so that the 
experimental program was not begun. Certain topics 
were taken up, however, including a search of the 
existing literature, examination of reports concerning 
the performance of existing noise meters, and certain 
mathematical analyses which will be useful when the 
experimental phase of the investigation is possible. 

The final report 4 contains the analysis of present 

b Project 13-123, Contract OEMsr-1478, University of Penn¬ 
sylvania. 


CONFIDENTIAL 


193 








194 


NOISE INVESTIGATIONS 


noise meters, a definition of noise, present noise-meter 
specifications as set up by the Joint Coordination 
Committee on Radio Reception of the Edison Electric 
Institute, the National Electrical Manufacturers As¬ 
sociation, and the Eadio Manufacturers Association. 


Methods of coupling a noise meter to a source are con¬ 
sidered. The appendices contain discussions of anten¬ 
nas for noise meters, detectors of electrical noise, 
methods of calibrating a noise meter, the problem of 
frequency conversion, and a bibliography. 1 ' 4 





INTERFERENCE REDUCTION COMMITTEE 


Until late in World War II, the problems of elec¬ 
trical interference between pieces of electronic equip¬ 
ment produced by the several NDRC divisions bad 
not reached the point where the situation was critical, 
although much good could have been done had some 
organization existed from the start to minimize such 
interference. Typical examples of such difficulty were 
the interference in communication equipment created 
by certain types of radar and the spurious responses 
registered on countermeasure search receivers caused 
hv other electronic apparatus mounted in the airplane 
carrying the search receiver. 

The effectiveness of only a small amount of coordi¬ 
nating activity was fully appreciated in October 1944 
when an informal meeting was held at which repre¬ 
sentatives of the Army, the Navy, and NDRC dis¬ 
cussed the organization of such a group. 

In January 1945, the Interference Reduction Com¬ 
mittee [IRCOM] was established by Ward F. David¬ 
son as a special committee of the Chairman’s Office 
of NDRC, with Alan Hazeltine as chairman and with 
representatives from Divisions 5, 13, 14, 15, and 17 
of NDRC. Although the life of the committee was 
short, terminating in October 1945, certain accom¬ 
plishments were made, and enough experience gained to 
indicate the importance of such a coordinating agency 
and to set the pattern for future work in the subject. 

IRCOM Accomplishments 

To limit its activities and thus to work most effec¬ 
tively, the committee excluded from its interest all 
interference caused by “administrative decision,” for 
example, if in a given campaign a radar and a commu¬ 
nications system interfered with each other by being 
on the same frequency, administrative action would 
be required to shift one of the frequencies or to take 
one of the Services off the air. Enemy jamming was 
beyond the scope of the committee, as were precipita¬ 
tion static and meteorological phenomena causing in¬ 
terference. Interference problems which fell entirely 


within the sphere of a single NDRC division were 
similarly excluded. 

IRCOM, therefore, considered matters where in¬ 
terference was caused accidentally by electronic de¬ 
vices which, given a sufficient refinement in design, 
should be able to work together in harmony. By and 
large, its work was mostly concerned with interference 
to communication services by noncommunication 
systems. 

Standards of Measurement 

The principal difficulty with an intelligent study 
of the interference problem proved to be the lack of 
standardized methods of making interference meas¬ 
urements. The Army and Navy, independently and 
on their own initiative, were attempting to establish 
specifications to standardize such methods of meas¬ 
urement so that limits on interference could be writ¬ 
ten into new contracts for electronic equipment. 
IRCOM took on this problem of measurements and 
produced a report on noise generators and noise 
meters. 

One of the principal jobs of IRCOM was to con¬ 
sider potential NDRC contracts referred to it. Its 
recommendations on interference measurements were 
as follows: 

Using the 931 vacuum tube as a source of “white” 
noise, the development of a stable noise generator 
should be undertaken by one of the Services. 

A study of the stability of the Detroit Signal Labo¬ 
ratory noise generator should be carried out at the 
Psycho-Acoustic Laboratory, Harvard University. 

When the hydrogen thyratron tube reaches the 
proper point of development, a pulse generator should 
be constructed around it as the nucleus. 

Exhaustive tests should be made of the noise gen¬ 
erator being developed by Purdue University (Project 
13-113, AC-238.04, OEMsr-1431) to further develop¬ 
ment work along similar lines or to aid in production 
of models. 5 ' 6 


CONFIDENTIAL 


195 




Chapter 33 

TRANSIENT RESPONSE OF BAND-PASS AMPLIFIERS 


Analysis of band-pass network response, including i-f, r-f, 
and stagger-tuned amplifier response to delta functions, i.e., 
pulses of zero duration, infinite amplitude, and finite area. 

33.1 INTRODUCTION 

T he response of a band-pass amplifier 1 to signals 
with waveforms having severe discontinuities is of 
great interest for many applications. For example, it 
is common practice to put a limiting circuit in a radio 
receiver to reduce interference from static. Such a 
limiter allows a signal to pass through the audio sec¬ 
tion of the receiver only if its amplitude is less than 
some predetermined maximum. Noise of amplitude 
greater than this is clipped and its effect greatly re¬ 
duced. If the limiter is to remove extraneous noise 
effectively, that noise must be present in the form of 
pulses of large amplitude and short duration at the 
input to the limiter. Static coming into the receiver 
often approximates a delta function (a pulse of zero 
duration and infinite amplitude but having finite 
area) in waveform. The r-f and i-f amplifiers of the 
receiver constitute a band-pass network. 

Equations derived and given in the final report 1 
indicate that to obtain output pulses of short duration 
and large amplitude, the frequency-response curve of 
r-f and i-f amplifiers should have a large band width 
with sides which slope very gradually. Of these two 
requirements the latter is more important. If the band 
width is increased, the primary effect is to increase 
the amplitude of the pulse. This results because the 
increased band width allows the circuits to obtain 
more energy from the input delta function. The large 
amplitude pulse is easily removed by a limiter, but 
there is a greater amount of energy which must be 
removed. Since a limiter clips at a fixed rather than 
a relative voltage, the duration of the truncated pulse 
passing through the limiter will be increased if the 
geometrical shape of the pulse is maintained and its 
amplitude increased. If the slope of the side of the 
response curve is decreased, maintaining constant 
band width, the amplitude of the pulse remains almost 

a Project 13-110, Problem No. 4, Contract No. OEMsr-1441, 
Harvard University, in collaboration with Project NS-108 of 
Division 17.3 NDRC, Psycho-Acoustic Laboratory, Harvard 
University. 


constant. The shape of the pulse is changed so that 
the truncated pulse from the limiter is of shorter 
duration. 

The calculation of the delta-function response for 
a complex amplifier network is unfortunately a tedious 
problem. In a Telecommunications Research Estab¬ 
lishment report, Twiss has calculated the step-func¬ 
tion response of a number of networks of the type de¬ 
scribed by Wallman 2 as stagger-tuned circuits. In 
these calculations he has replaced the actual band¬ 
pass r-f network with its video analogue and so ob¬ 
tains directly the envelope of the response. The band¬ 
pass network having a steady-state response curve 
symmetrical about its mid-frequency is replaced by a 
hypothetical low-pass network having a response curve 
identical except that it is symmetrical about zero 
frequency. The transient response of this low-pass 
network is directly the envelope of the response of 
the band-pass network. This procedure is valid only 
provided the band width is small compared with the 
mid-frequency. The fact that it yields the envelope 
of the response is usually a distinct advantage, since 
the envelope is normally the feature of greatest 
interest. 

For such stagger-tuned networks Twiss 3 has shown 
that the steady-state response as a function of fre¬ 
quency may be expressed as 
1 

3 ~ (1 + .r 2r )" /2 ’ ^ 

where g is the relative response; 

r is the number of tuned circuits in one 
staggered unit (one r-uple) ; 
n is the number of such units; 
x is the normalized frequency variable 2f/B r ; 
/ is the frequency measured from the mid¬ 
band frequency; 

B 1 is the band width of one circuit, measured 
between frequencies where the response is 
1/V2 times the maximum. 

This equation assumes that the band width of the net¬ 
work is small compared with its mid-frequency. Twiss 
gives calculations for networks of this sort for a rea¬ 
sonably wide range of values of both r and n. He 
further points out just what physical circuits corre¬ 
spond to these mathematical values. His data are given 


c 


ONFIDENTIAL 


i 


196 






CONCLUSIONS 


197 


in the form of curves plotting the voltage output due 
to a unit step-function input as a function of the 
normalized time variable, ( t/t 0 ) — -n-BJ. 

In the present study the primary interest was in 
obtaining some sort of approximate relations from 
which important characteristics of the delta-function 
response could easily be calculated. If possible, these 
relations should apply to all band-pass systems in gen¬ 
eral and should be simple and accurate enough for 
engineering use. The procedure used was to find the 
delta-function response of the stagger-tuned networks 
of Twiss by finding the time derivative of his step- 
function response curves, making use of a curve-fitting 
process. These delta-function responses were then 
related to the shapes of the steady-state responses of 
the networks. 

33.2 RESULTS OF ANALYSIS 

The response of a stagger-tuned band-pass network 
to a delta function may be approximated by two em¬ 
pirical relations. The first of these is 
t' - 2.6 0 = 0.21 4 > 
t"= 1.8 0 = 0.15 4> 

where V is the duration of the delta-function response 
measured 60 db below its peak, and t" is the duration 
measured 40 db below the peak. The slope of the side 
of the steady-state frequency-response curve is 0, where 
g is the relative response and the slope is measured at 
g = l/y/2. If the curve is expressed in decibels and 
the slope is measured at G = —3 db, the slope is 4>. 
In either case the slope is measured in terms of a 
frequency scale in units of cycles per second. 

The second relation is 

0.6 7.5 

e * =B ’' + -T =Bn + -r> (3) 

0 0 

where e s is the peak of the delta-function response 
and B n is the overall band width of the network meas¬ 
ured at g — l/y/2, or at G = —3 db. This peak is 
for an amplifier network of unit voltage amplification 
and for a delta function of unit area. 

These relations were obtained empirically for cer¬ 
tain stagger-tuned networks, but it is contended that 
they are equally useful for band-pass networks in 
general, so long as the band width of the network is 
small compared with its mid-frequency. 


Typical Example 


As an example of the application of the approxi¬ 
mate relations, consider a typical i-f amplifier with 
the following characteristics: 

Mid-frequency, 465 kc. 

Band width, 3 db down, ±5 kc. 

Slope of response curve, 3 db down, 2.5 db/kc. 

Voltage amplification, 1,000. 

Input-pulse duration, 0.5 fisec. 

Input-pulse amplitude, 10 volts. 

Since the input-pulse duration is much smaller than 
the reciprocal of the mid-frequency, the equations de¬ 
rived may be used to find the duration of the pulse 
at the output of the amplifier and its amplitude. The 
duration is 

t f = 0.21 4> = 0.21 X 2.5 X 10~ 3 = 525 p sec, 
r = 0.15 4> = 0.15 X 2.5 X 10- 3 - 375 /isec. 
Assuming that the input pulse is rectangular, its area 
a is 


a = 10 X 0.5 X 10~ 6 = 5 X 10 -6 volt seconds. 


The amplitude of the output pulse is 

e=B n -f — = (5 + s) X 10 3 -f — 
5 ” ' $ v 1 ' 1 2.5 


7.5 

X io- 3 


= 13,000 volts. 

This amplitude is for an input pulse of unit area 
and does not consider the voltage amplification of the 
amplifier. Taking account of these factors, the actual 
amplitude of the output pulse is 

e d = area X voltage amplification X 13,000, 

= 5 X 10- 6 X 1,000 X 13,000, 

= 65 volts. 


333 CONCLUSIONS 

While the approximate relations for the calculation 
of the duration and peak of the output response of a 
network which has a delta function applied to its in¬ 
put are admittedly empirical, they give results which 
are of engineering value for the stagger-tuned net¬ 
works from which they were derived. Of course, this 
gives no assurance that they are equally valid in the 
case of any other arbitrary network. They are of the 
correct form to agree with experimental observations, 
however, and it is felt that they should be useful in 
the design of any band-pass system whose band width 
is small compared with its mid-frequency. Qualita¬ 
tively they may lead to useful generalizations. 


CONFIDENTIAL^ 








Chapter 34 

MEASUREMENTS OF MAGNETIC PROPERTIES OF 
FERRITE CORE MATERIALS 


Examination 11 of the characteristics of certain ferrite ma¬ 
terials brought to the United States from the Phillips Com¬ 
pany, Eindhoven, Holland, in May 1945. Two of the ma¬ 
terials, MnZn ferrite, appeared to be suitable for low fre¬ 
quencies of 10 to 60 kc, and the third, CuZn ferrite, was less 
suitable for low frequencies but showed possibilities at fre¬ 
quencies as high as several megacycles. All were unstable 
with regard to temperature and voltage. 


34i MEASUREMENTS 


T he core samples were in toroid form and it was 
only necessary to wind upon them an appropriate 
number of turns (5 to 215) for a given frequency 
range and to measure values of inductance and either 
Q or the apparent resistance. The relations of the 
several coil constants are as follows: 

Magnetic Permeability, p . For a closely wound 
toroid on a uniform core, the permeability and the 
inductance are related by 


1.25 6N 2 A X 10- 8 ’ 


where l is the effective length of the flux path in cm, 
L is the inductance of the coil, N is the number of 
turns and A is the cross section area of the core in 
sq cm. 

Specific Loss Factor, p. The power dissipated in the 
coil carrying an alternating current is 

W = I 2 R, (2) 

where R is the total equivalent resistance of the coil. 
This resistance can be divided into R c , the resistance 
of the coil without the core, and Rp the resistance 
produced by the core losses. If W* is the power dis¬ 
sipated because of the core then 

1U = PPn, (3) 

Now p is defined as 

1 Vi _ PRi watts/cm 3 ,. * 

9 B 2 V B 2 V gauss 2 ’ 
where V is the volume of the core material and B is 
the magnetic induction. 

For a toroid 


4cTtN Ip 

B = ■ 1Q l gausses, 


(5) 


“Project 13-110, Problem No. 8; Contract OEMsr-1441; 
Harvard University. 


where l is the length of the flux path. 
Then 


100 PRil 2 
9 ~ 16tt 2 N 2 I 2 p 2 T 
The inductance L is 

4:7rN 2 IpA 

T — _ r 

10Z7 X 10 s ’ 


(6) 

(?) 


and, since V = IA, 

_ Rj watts/cm 3 
9 4:irpL X 10 7 gauss 2 

Values of Ri were obtained by measuring the total 
resistance and subtracting from it the resistance of a 
similar coil without the core. At frequencies below 100 
kc the d-c resistance was practically the same as R c . 

Q Measurements. R was obtained by means of a Q 
meter which measured the ratio of the inductive re¬ 
actance Lo) to the total equivalent resistance R of the 
coil and core. The Q meter gave values of Q and of G, 
the capacitance required to tune the coil to a given 
frequency. Plotting 1/w 2 as a function of C gave a 
graph whose slope is L, the true inductance. From this 
value and that of Q, the 'resistance can be obtained. 
Thus 


Lm 



By subtracting R c , the resistance of an identical coil 
with an air core, Ri is determined and can be calcu¬ 
lated by equation (8). 


3411 Equipment Employed 

In the range between 60 and 400 kc, a Boonton 
Type 100-A Q meter was used with a precision capac¬ 
itor (General Radio) in parallel with the Q meter 
capacitor for greater precision. 

In the range of 400 to 5,000 kc, a General Radio 
Type 916-A r-f bridge was used. Trouble was en¬ 
countered in the range 1 to 140 kc because of the 
necessity of applying too much voltage to the coil. 
By using a General Radio Type P-508 bridge as 
Owen bridge for measuring inductance and as a reso¬ 
nant bridge for measuring Q, P could be calculated. 

With the Owen bridge and the several resonance 
bridges employed, balance was indicated by an oscil- 


198 


CONFIDENTIAL 











MAGNETIC PERMEABILITY 


199 


loscope preceded by a tuned circuit and 200 db of 
available amplification. Visual discrimination between 
the fundamental and harmonic aided in detecting cor¬ 
rect balance. 


34.2 


MAGNETIC PERMEABILITY 


The magnetic permeability p. was virtually constant 
over the frequency range used. Some variations were 
apparently due to changes in temperature, to changes 
in applied magnetic intensities, and to flux leakage. 
Measured values are as follows: 


Material 
MnZn ferrite A 
MnZn ferrite B 
CuZn ferrite 


/x (avg 
757 
846 
392 


Range of /x 
734- 770 
834-860 
382 -396 


The final report 1 gives all the data taken and 
shows that the materials were superior to molybdenum 
permalloy in that they have higher permeabilities and 
lower losses but are inferior in stability. A marked 
change in properties occurred when the material was 
subjected to high magnetization — 1,000 gausses. A 
permanent change was caused, although for several 
days the permeability slowly changed in the direction 
of its former value. 

Although the materials showed promise, it was con¬ 
cluded that they would be useless unless means were 
found to stabilize their characteristics to reduce both 
temperature coefficient for /x and p and to reduce the 
change in values which occurs when magnetic induc¬ 
tion values greater than 10 gausses were applied. 


CONFIDENTIAL 







Chapter 35 


AIRCRAFT ANTENNA POWER, IMPEDANCE, AND 
TUNING-NETWORK SURVEY 


Study to determine tuning and loading range of certain 
transmitters, methods of measuring antenna impedance, and 
methods for measuring antenna power. 

35.1 PROJECT SCOPE 

he objectives of this project 3 may be summarized 
as follows: 

1. The development of a method and necessary 
equipment for determining the range of complex an¬ 
tenna impedances into which a given transmitter will 
tune the load properly, proper loading being defined 
as that which will draw normal plate current when 
the tuning control is set for minimum plate current, 
measurements to be made at the antenna terminals 
and/or at the power-amplifier plate terminals. 

2. A survey of the ranges of complex antenna imped¬ 
ance into which specific aircraft transmitters will tune 
properly as defined above. The antennas into which 
these transmitters operate may have resistive compo¬ 
nents ranging from V 2 to 15,000 ohms, and reactive 
components ranging from —15,000 to -|-15,000 ohms. 

3. The development of a method and equipment for 
measuring the resistance and reactance of an aircraft 
antenna within an accuracy of 5 per cent and over a 
frequency range of 1 to 20 me. The method is to be 
simple in application and is to require no extensive 
calculations. 

4. The development of a method and test equip¬ 
ment for measuring the power output of a transmitter 
into its antenna over a range of 5 to 125 watts within 
an accuracy of 5 per cent over a substantial portion 
of the range. 

At the end of the war and termination of the con¬ 
tract, no work had been accomplished on 4, which is 
being carried on under contract with the Office of 
Naval Research. 

35.2 TRANSMITTER ANTENNA 

IMPEDANCE RANGE 

This problem is not simple, largely because of the 
wide range of frequencies and of antenna impedances 
which must be considered, and also because of the 
variety of circuits in the power-amplifier stage of 
military transmitters. 

Two methods have been tried for determining the 

“Project 13-110, Problem No. 6, Contract No. OEMsr-1441, 
Harvard University. 


loading range. In one method a dummy antenna or 
adjustable load is used for actually loading the trans¬ 
mitter. The other method obtains the loading range 
from impedance measurements made at the power 
amplifier and antenna terminals. 

352 1 Method Using Adjustable 
Loading Unit 

The adjustable load provides the combinations of 
resistance and reactance normally encountered in the 
operation of aircraft transmitters in the 2- to 18-mc 
range. In addition, it facilitates measurement of the 
transmitter power output and hence provides data giv¬ 
ing the useful loading range of the transmitter. The 
variations of resistance and reactance are obtained 
through the use of several circuits composed of vari¬ 
able inductance, variable capacitance, and resistance. 
Physically large circuit elements are used to dissipate 
125 watts or more and to withstand the large currents 
and voltages which may exist. 

It was found impractical to provide an accurate 
calibration of the adjustable loading unit. The stray 
capacitances due to wiring, the many possible com¬ 
binations of components, and effects due to heating, 
all preclude reliable calibration of the unit. While 
approximate calibration curves have been prepared 
to assist in the adjustment of the loading unit, a more 
accurate value of the loading impedance was obtained 
by measurement with impedance bridges. 

The application of the adjustable loading unit for 
determining the loading range of a power transmitter 
is described in the final report. 1 Time permitted 
only a partial survey of the BC-375-E transmitter, 
but the loading range provided by each of its four 
antenna circuits was determined at one frequency, 4 me. 

35 - 2 - 2 T-Network Method 

Section IV of the final report describes a method 
for determining the loading range of a transmitter 
by means of a series of impedance measurements. The 
method utilizes the facts that (1) a tuning and loading 
network can be represented by a T-network and (2) a 
properly tuned and loaded power-amplifier tube op¬ 
erates into a plate load which is a pure resistance R L . 
By measuring the resistance R L suitable for the power- 
amplifier tube, by making open-circuit impedance 



200 


CONFIDENTIAL 





COMPLEX IMPEDANCES OF AIRCRAFT ANTENNAS 


201 


measurements at the power-amplifier (Z 0 1 ) and an¬ 
tenna terminals (Z 02 ), and by making a measure¬ 
ment at the power-amplifier terminals with the an¬ 
tenna terminals short-circuited (Z^), the unknown 
(antenna) impedance Z x can be computed from 


The plate load R L must be measured whenever the 
operation of the power-amplifier tube changes with 
frequency, and the other measurements must be made 
for a large number of adjustments of the tuning and 
loading controls. It should be observed that the ex¬ 
treme settings of the tuning and loading controls 
do not completely define the range of impedances 
which may be used with a given antenna circuit. 

The T-network method has been used at a single 
frequency in determining the tuning and loading 
range provided by four of the thirteen circuits pres¬ 
ent in the AN/ART-13 transmitter. 

35 - 2 - 3 Conclusions 

In the somewhat limited application of the two 
methods, certain opinions were formed which may 
assist in their evaluation. These may be summarized 
as follows: 

1. Methods using an adjustable loading unit and 
T-network measurements will each yield the results 
desired. 

2. Both methods require the availability of im¬ 
pedance-measuring equipment, and one requires an 
adjustable loading unit in addition. In any event, 
facility in transmitter operation and in the use of 
r-f impedance-measuring equipment is required. 

3. Both methods are time consuming because of 
the large amount of data required to specify complete¬ 
ly a transmitter loading range. However, the method 
which utilizes an adjustable load is more direct and 
appears to have the advantage in this respect. 

4. The relative accuracies of the methods depends 
upon the circuit being analyzed. The accuracy of the 
T-network method is reduced when its application re¬ 
quires measurement of the impedance of parallel reso¬ 
nant circuits. The results obtained with the adjustable 
load are more indefinite in those cases in which trans¬ 
mitter tuning and loading adjustments are broad. 

5. The adjustable load facilitates measurements of 
power output; the T-network does not. This is an 
important consideration, for the impedance loading 
range as determined by rated plate current may not 
be the useful loading range as determined by r-f 
power output. 


353 COMPLEX IMPEDANCES OF 
AIRCRAFT ANTENNAS 

A survey has been made of the literature and of 
present-day impedance-measuring equipment appli¬ 
cable to this assignment. The bibliography included 
in the final report contains the most pertinent ref¬ 
erences. For such an important subject, surprisingly 
little work has been reported. Of the commercial in¬ 
struments available, the Boonton Radio Corporation 
$-Meter line and the General Radio Company com¬ 
panion pieces, the Type 916-A R-F Bridge and the 
Type 821-A Twin-T Impedance-Measuring Circuit, 
seem to be most applicable at the present time. 

None of the available impedance-measuring equip¬ 
ment has been found to combine the requirement of 
simplicity of operation (without extensive calcula¬ 
tions) with that of 5 per cent accuracy over the wide 
ranges of impedance and frequency. Furthermore, 
the impedance bridges with their required accessories 
are considered to be too bulky and heavy to be used 
conveniently in airplanes in flight. Accordingly, plans 
have been directed toward the modification of existing 
methods and equipment and toward the development 
of new methods. 

Methods under investigation, which include certain 
original features, are the following: a substitution 
which utilizes a decade r-f resistor unit, a method 
which uses substitution (dummy) antenna units to 
provide duplicate values of antenna impedances so 
that measurements may be performed in the labora¬ 
tory, and a low-power transmitter which utilizes 
power-amplifier plate current to measure both resist¬ 
ance and reactance. Time permitted only partial in¬ 
vestigations of these methods. Pending the outcome 
of this work, no effort has been made to adapt com¬ 
mercial equipment for this purpose. 

The following additional conclusions are indicated 
in connection with the impedance-measuring methods 
considered: 

The possibility of using small dummy antenna 
units for transferring the antenna impedance char¬ 
acteristics to the laboratory seems entirely feasible. 
A more direct method may be preferred even though 
it be somewhat less accurate. 

The principle of the impedance-measuring trans¬ 
mitter seems to satisfy the requirements of size, 
weight, and simplicity of operation. Incomplete tests 
indicate that the equipment may be made substantially 
direct-reading over a considerable portion of the im¬ 
pedance range. 


CONFIDENTIAL 

















































i 

. 



























































' .. u 


.. . 






































































































BIBLIOGRAPHY 


Numbers such as Div. 13-200.1-MI indicate that the document listed has been microfilmed and that its title appears 
in the microfilm index printed in a separate volume. For access to the index volume and to the microfilm, consult the 
Army or Navy agency listed on the reverse of the half-title page. 


Chapter 1 

The final report on Project C-79 (Contract OEMsr-lOlS) 
was composed of three parts, each made up of numerous indi¬ 
vidual reports as indicated below: 

1. Systems Engineering for Army A ir Forces Communications, 

Part I, A. B. Clark, OSRD 1442, OEMsr-1018, Report 
2519, BTL, Apr. 27, 1943. Div. 13-200.1-MI 

Report of Progress, A. B. Clark, Apr. 27, 1943. 
Requirements for Radio Sets for Ten Air Warning Systems 
for Overseas Shipments Beginning August 1, 1943, A. 
Tradup, Apr. 26, 1943. 

Tentative Requirements for Radio Sets for Air Warning 
Systems for Overseas Shipments Beginning January 1 , 1944, 
A. Tradup, Apr. 26, 1943. 

Atmospheric Static at Radio Frequencies, R. S. Tucker, 
Apr. 21, 1943. 

Atmospheric Static at Radio Frequencies, R. S. Tucker, 
Apr. 23, 1943. 

Range of Radio Sets vs. Frequency — Summary, K. Bulling- 
ton, Mar. 24, 1943, revised Apr. 24, 1943. 

Range of Radio Sets vs. Frequency — Detail, K. Bullington, 
Mar. 24, 1943, revised Apr. 24, 1943. 

The Effect of Trees on Radio Propagation, K. Bullington, 
Apr. 28, 1943. 

Transmission in the 10-Meter Band, R. P. Booth, Apr. 26, 
1943. 

Antennas and Field Operation, R. V. Crawford, Apr. 27, 
1943. 

Radio Repeaters or Relays, M. L. Almquist, Apr. 27, 1943. 
Notes Concerning Maintenance, A. Tradup, Apr. 19, 1943. 

2. Systems Engineering for Army Air Forces Communications, 

Part II, A. Tradup, OSRD 1925, OEMsr-lOlS, BTL, 
Oct. 1, 1943. Div. 13-200.1-M2 

Foreword, A. B. Clark, Oct. 1, 1943. 

Summary of Progress Report Material, M. B. McDavitt, 
Sept. 7, 1943. 

Procurement Status of Items Recommended by Section 13.7, 

M. B. McDavitt, Sept. 8, 1943. 

Program for Radio Equipment to be Used in Airborne Air 
Warning Systems, M. B. McDavitt, Aug. 2, 1943. 
Participation of Section 13.7 in Mountain Maneuvers, H. II. 
Abbott, Sept. 8, 1943. 

Transmission in the 27-40 Megacycle Range, R. P. Booth, 
Sept. 13, 1943. 

Field Intensity Charts for Use in Circuit Layout, K. Bulling¬ 
ton, Sept. 6, 1943. 

Report of Field Intensity Tests Made During the Asheville, 

N. C. Maneuvers—Case 23263-34, R. P. Booth, July 14, 
1943. 

Report on 37.1 Me Radio Propagation Tests at Millburn, 
N. ./.—-Case 23263, E. W. Borden, Aug. 3, 1943. 
Comparison of Measured with Estimated Field Intensities— 
File 23263-34, R. P. Booth, Sept. 18, 1943. 

Modifications of Motorola FM Radio Equipment and Rec¬ 
ommendations for Use, W. R. Young, Aug. 30, 1943. 

Line Voltage Requirements for Motorola FM Radio Sets with 
Proposal for Modification to Avoid Overheating—Case 
23263-34, W. J. Kopp, July 15, 1943. 


Harmonic Suppression Requirements—Case 23263-34, R- P. 
Booth, July 14, 1943. 

Tentative Specification for Radio Equipment for Airborne 
Air Warning System, M. E. Maloney, Sept. 3, 1943. 
Maintenance Considerations, M. E. Maloney, Sept. 3, 1943. 
Outline of Instruction Manual for Airborne Air Warning 
System Communications Equipment, M. E. Malonev, 
Sept, 3, 1943. 

Laboratory Comparison of FM and AM Radio Systems, 
Particularly with Regard to the Effect of Jamming, K. G. 
Van Wynen, Sept. 14, 1943. 

Hand Powered Transmitter for Isolated Observers for Air 
Warning System, M. L. Almquist, Aug. 3, 1943. 

Report on Development of Remote Controls for Motorola 
Radio Sets, M. E. Maloney, Sept, 3, 1943. 

Engineering Description of Remote Control Circuit for Mo¬ 
torola FMT-50 BC Transmitter and FSKR-15 BC Receiver, 
M. E. Maloney, Sept. 3, 1943. 

Remote Operation of Radio Equipment at Radar Point, G. C. 
Reier, July 10, 1943. 

Optimum Talking Volume for Motorola FMT-50 Trans¬ 
mitters, K. G. Van Wynen, July 16, 1943. 

Volume Limiters for Air Warning System, G. C. Reier, 
July 6, 1943. 

Calibration of Link Adjustable Coaxial Antenna, R. P. 
Booth, Sept. 9, 1943. 

Flexible Dipole Antennas—Description and Performance, 
R. P. Booth, Sept, 22, 1943. 

Field Test of the Performance of Antenna Coupling Units, 
II. P. Booth, Sept. 10, 1943. 

Antenna Kits for Radio Communication in Airborne Air 
Warning Systems, H. H. Abbott, July 31, 1943. 

Masts for Radio Antennas in Airborne Air Warning Sys¬ 
tems—Design Considerations, H. H. Abbott, Sept. 4, 1943. 
50-Foot Plywood Mast for Radio Ardennas—Instructions 
for Use, H. H. Abbott, Sept. 15, 1943. 

Power Plants for Radio Sets Used in Airborne A ir Warning 
Service, A. Tradup, Sept. 6, 1943. 

Investigation of Engine Noise Problems, V. T. Callahan and 
D. F. Seacord, Aug. 14, 1943. 

3. Systems Engineering for Army Air Forces Communications, 
Part III, A. Tradup, OSRD 4293, OEMsr-1018, BTL, 
Aug. 30, 1944. Div. 13-200.1-M3 

Summary, P. B. Fairlamb, Aug. 30, 1944. 

IIF Sky-Wave Transmission over Short or Moderate Dis¬ 
tances Using Half-Wave Horizontal or Sloping Ardennas, 
A. J. Aikens, R. S. Tucker, and A. G. Chapman, July 15, 
1944. 

Field Intensity Charts for Radio Circuit Layout in the VHF 
Band, K. Bullington, June 12, 1944. 

Summary of Results of Field Trial of SCR-300 Radio Sets 
for Air Forces Use, R, V. Crawford, June 22, 1944. 
History of Development of Radio Set AN/TRC-19, P. B. 
Fairlamb, June 23, 1944. 

Laboratory Investigation of Radio Set AN/TRC-19, W. J. 
Kopp and W. R. Young, June 26, 1944. 

Circuit Description of Radio Set AN/TRC-19 and Outline 
of Alignment Procedure, W. R. Young, May 30, 1944. 
Telephone Sets and Remote Control Features Incorporated in 
Radio Set AN/TRC-19, K. G. Van Wynen, June 13, 1944. 


CONFIDENTIAL 


203 






204 


BIBLIOGRAPHY 


Volume Limiting Preamplifier for the AN/TRC-19 Radio 
Transmitter, J. A. Weller, June 13, 1944. 

Laboratory Investigation of the Use of AN/ TRC-1 Radio 
Sets in Army Air Forces Applications, K. G. Van Wynen 
and W. R. Young, Jan. 17, 1944. 

Remote Control of Radio Set AN /TRC-1; Development of 
Remote Control Equipment AN/TRA-2, P. B. Fairlamb 
and E. C. Borman, June 14, 1944. 

Frequency Assignment Studies for Various VHF Radio 
Sets, R. W. Grigg, June 21, 1944. 

Flexible Dipole Antennas for the 40-48 Megacycle and 70-100 
Megacycle Ranges, H. W. Nylund, Apr. 26, 1944. 

Vertical Coaxial Antenna with Reflector and Director Ele¬ 
ments for 70-100 Megacycle Range, H. W. Nvlund, May 1, 
1944. 

CW Operation with FM Radio Sets Applique Unit for 
AN/TRC-1 and AN/CRC-3 FM Receivers, W. R. Young 
and W. J. Kopp, Mar. 17, 1944. 

Manual Tone Modulated ( MCW) Telegraph on Radio 
Channels Using Telegraph Set TG-5-( ) Western Electric 
Co. Type 111 Mechanical Amplifier or Tone Telegraph Set 
Similar to TG-5-{ ), P. B. Fairlamb and E. C. Borman, 
June 22, 1944. 

Speed Comparison of Man ual Telegraph and Voice on Cipher 
Transmission, K. G. Van Wynen, Jan. 10, 1944. 
Description of Telegraph Terminal TH-3 ( )/TC, R. A. 
Vanderlippe and E. Von Nostitz, July 21, 1944. 
Application of Teletype to Radio Transmission Systems in 
the Army Air Forces, R. A. Vanderlippe and W. R. Young, 
Jan. 5, 1944. 

Laboratory Investigation of the Application of Single Chan¬ 
nel Voice-Frequency Carrier Telegraph to Modified Motorola 
Radio Sets, W. J. Kopp and A. Wilson, Jan. 10, 1944. 
Laboratory Investigation of the Application of 4 Channel 
Voice Frequency Carrier Telegraph to Motorola Radio Sets 
Operating in the 30-40 Megacycle Range, K. G. Van Wynen 
and A. Wilson, Dec. 4, 1943. 

Simultaneous Transmission of Speech and Teletypewriter 
Signals over Radio Telephone Channels, K. G. Van Wynen 
and R. A. Vanderlippe, Aug. 14, 1944. 

Air Laid Field Wire Using Carpet-Type Packs, L. R. 
Montfort, July 1, 1944. 

60-Foot Telescoping Pneumatic Tubular Steel Antenna Mast, 

G. H. Duhnkrack, June 21, 1944. 

30-Foot Tubular Plywood Antenna Mast, G. H. Duhnkrack, 
June 8, 1944. 

General Instructions for the 30-Foot Tubular Plywood An¬ 
tenna Mast, G. H. Duhnkrack, June 8, 1944. 

Test of Power Units PE-214-A, V. T. Callahan, Mar. 7, 
1944. 

Final Disposition of Recommended Changes of Motorola 
Radio Equipment Used in AN/CRC-3A, W. R. Young, 
May 9, 1944. 

Circuit Description of AN/CRC-3A Radio Transmitter and 
Receiver Units, W. R. Young, May 3, 1944. 

Handbook on Transmission in the 27-40 Megacycle Band, 
E. W. Borden, July 26, 1944. 

Calibration of Link 27-40 Megacycle Adjustable Coaxial 
Antennas, R. P. Booth, June 19, 1944. 

Completion of Design of 50-Foot Tubular Plyivood Antenna 
Mast, G. H. Duhnkrack, June 15, 1944. 

Chapter 2 

1. Effect of Trees and Hills as Obstructions to Radio Propaga¬ 
tion, Delmar C. Ports, OSRD 3070, OEMsr-1010, Report 
13.2-83, Jansky and Bailey, Nov. 1943. Div. 13-200.2-M4 


2. Radiotelephone Communication Between Mobile Units, 
Stuart L. Bailey, OSRD 1618, OEMsr-174, Project C-30, 
Jansky and Bailey, June 1943. Div. 13-200.2-M3 

Chapter 3 

1. Study of FM vs. AM for Use in Airborne VHF Communica¬ 
tion, Alexander H. Wing, Jr., OSRD 6280, OEMsr-1441, 
Project 13-110, Problem No. 2, Cruft Laboratory, Har¬ 
vard University, Dec. 1, 1945. Div. 13-100-M2 


Chapter 4 

1. The Use of Pulse Modulation for Communication Systems, 

H. O. Peterson, H. P. Thomas, and others, BTL, May 12, 
1943. 

2. “Pulse Time Modulation,” E. M. Deloraine and E. 
Labin, Electrical Communication, Vol. 22, 1944-1945, pp. 
3-10. 

3. Bridge Circuits for Microwaves, W. A. Tvrell, Report MM- 
42-160-10, BTL, Feb. 12, 1942. 

4. A Broad-Band Balanced X-Band Crystal Converter, W. M. 
Sharpless, Report MM-44-160-134, BTL, June 12, 1944. 

5. The Use of the Magic Tee Microwave Bridge in Measuring 
Impedance, R. L. Kyhl, OEMsr-262, Radiation Labora¬ 
tory Report 643, MIT, Dec. 12, 1944. Div. 14-252.1-M6 

6. A Microwave Frequency Discriminator, R. V. Pound, 

OEMsr-262, Radiation Laboratory Report 662, MIT, Aug. 
4, 1945. * Div. 14-212.8-M7 

7. A Survey of Microwave Communication Systems in the 
United States, OSRD 6278, Project 13-110, Problem No. 
9, Cruft Laboratory, Harvard University, Sept. 1, 1945. 

Chapter 5 

1. 3000 Megacycle Communication, H. H. Beverage, OSRD 
474, OEMsr-32, Project C-24, RCA, Mar. 10, 1942. 

Div. 13-201-MI 

2. Microwave Telephone, H. H. Beverage, OSRD 1512, 
OEMsr-442, Project C-42, RCA, Mar. 22, 1943. 

Div. 13-200.3-MI 

3. Klystron Transmitter for 3000 Megacycle Operation, OSRD 
210, NDCrc-191, Project C-10, Westinghouse Electric and 
Manufacturing Co., Nov. 11, 1941. Div. 13-201.1-MI 

4. R. F. Generator ( 2000-3000 Me), R. K. Potter, OSRD 205, 
NDCrc-177, Project C-7, BTL, Oct. 1, 1941. 

Div. 13-201.2-Ml 

5. 3000 Me Receivers, R. S. Holmes, OSRD 482, NDCrc-75, 

Project C-2, Feb. 19, 1942. Div. 13-201.1-M2 

6. A New Method for Measuring Dielectric Constant and Loss 
in the Range of Centimeter Waves, S. Roberts and Arthur 
R. von Hippel, OEMsr-262, MIT, March 1941. 

CP-521-MI 

Chapter 6 

1 . R. F. Generator (2000-3000 Me), R. K. Potter, OSRD 
205, NDCrc-177, Project C-7, BTL, Oct. 1, 1941. 

Div. 13-201.2-MI 

Chapter 7 

1. Precipitation Static Reduction Research, W. H. Huggins, 
OSRD 1907, OEMsr-92, Project C-21, Oregon State Col¬ 
lege, Mar. 31, 1943. Div. 13-202.1-MI 


5 


CONFIDENTIAL 





BIBLIOGRAPHY 


205 


Chapter 8 

1. Precipitation Static Research, Sheldon H. Dike, OSRD 
1411, OEMsr-678, Project 0-41, University of New 
Mexico, May 1, 1943. Div. 13-202.21-MI 

Chapter 9 

1. Study of Effect of Aircraft Surface Treatment on Electrical 

Charges Causing Precipitation Static, Roscoe H. George, 
OSRD 1446, OEMsr-679, Project C-64, Purdue Research 
Foundation, May 11, 1943. Div. 13-202.21-M2 

2. Static Electricity, Francis B. Silsbee, Bureau of Standards, 
Cir. C438, June 10, 1942, p. 10. 

Chapter 10 

1. Precipitation Static Research, E. P. Buckthal and K. M. 
Cummings, OSRD 1807, OEMsr-893, Project C-68, United 
Air Lines, Aug. 31, 1943. Div. 13-202.21-M3 


13. “Automatic Volume Control for Radio Receiving Sets," 
Proceedings of the Institute of Radio Engineers, Vol. 16, 
1928, p. 30. 

14. “Zur Theorie der Frequenzanalyse mittels Suchtons,” H. 
Salinger, Elektrische Nachrichten-Technik, Vol. 8, 1929, 
p. 293. 

15. “Panoramic Reception for Increased Receiving Effi¬ 
ciency,” Electronics, Vol. 14, December 1941, p. 36. 

16. Transmission Networks and Wave Filters, T. E. Shea, 
D. Van Nostrand Co. Inc., Sec. 80, 1929, p. 439. 

17. “Transmission Characteristics of a Short Wave Telephone 
Circuit,” R. K. Potter, Proceedings of the Institute of Radio 
Engineers, Vol. 18, 1930, p. 581. 

18. “Das ‘Tonfrequenz Specktrometer’,” E. Freystedt, Zeit- 
schrift fur Technische Physik, Vol. 16, 1935, p. 533. 

19. U. S. Patent 2,273,914, M. Wallace, Feb. 24, 1942. 

20. “Spectrum Analyzers,” Everard M. Williams, Proceedings 
of the Institute of Radio Engineers, Vol. 34, 1945, p. 18. 

21. Zur Theorie der Frequenzanalyse mittels Suchtons, Elek¬ 
trische Nachrichten-Technik, Vol. 6, August 1929, p. 293. 


Chapter 11 

1. Precipitation Static Reduction Research, Homer J. Dana, 
OSRD 3595, OEMsr-848, Project C-70, Washington State 
College, Mar. 20, 1944. Div. 13-202.21-M4 

Chapter 12 

1. The final report on Project C-36 was presented in three 
sections as follows: 

The Fundamentals of Panoramic Reception, Estill I. Green, 
OSRD 1224, OEMsr-357, BTL, Jan. 20, 1943. 

Div. 13-203.1-MI 

Improved Panoramic Receiver, Part I, E. R. Taylor, OSRD 
909, OEMsr-357, BTL, Sept, 1, 1942. Div. 13-203.3-MI 
Improved Panoramic Receiver, Part II, E. R. Taylor, OSRD 
909, OEMsr-357, BTL, Jan. 22, 1943. Div. 13-203.3-M2 

2. Panoramic Receiver with Moving Screen Indicator, R. A. 

Heising, OSRD 396, OEMsr-49, Project C-27, BTL, Feb. 
19, 1942. Div. 13-203.2-Ml 

3. Receiver for Pulse Signals, OSRD 1021, OEMsr-311, Proj¬ 
ect C-39, BTL, Nov. 2, 1942. Div. 13-203.4-Ml 

4. “An Analyzer for the Voice Frequency Range,” C. R. 
Moore and A. S. Curtis, Bell System Technical Journal, 
Vol. VI, 1927, p. 217. 

5. “Analyzer for Complex Electric Waves,” A. G. Landeen, 
Bell System Technical Journal, Vol. VI, 1927, p. 230. 

6. “Eine neue Methode der Klanganalyse,” M. Grutzmacher, 
Zeitschrift fiir Technische Physik, Leipzig, Vol. 8, 1927, 
p. 506. 

7. “Panoramic Reception,” Electronics, Vol. 11, July 1938, 
p. 36. 

8. “Panoramic Reception,” Electronics, Vol. 13, No. 6, June 

1940, p. 14. 

9. “Automatic Frequency Control,” C. Travis, Proceedings 
of the Institute of Radio Engineers, Vol. 23, 1935, p. 1125. 

10. “Intermediate Frequency Values for Frequency Modu¬ 
lated Wave Receivers,” D. E. Foster and J. A. Rankin, 
Proceedings of the Institute of Radio Engineers, Vol. 29, 

1941, p. 546. 

11. “The Double Superheterodyne Receiver,” R. I. Kinross, 
Wireless Engineer, Vol. 14, 1937, p. 351. 

12. “Devices for Controlling Amplitude Characteristics of Tel¬ 
ephonic Signals,” A. C. Norwine, Bell System Technical 
Journal, Vol. XVII, 1938, p. 539. 


Chapter 13 

1. Panoramic Receiver with Moving Screen Indicator, R. A. 
Heising, OSRD 396, OEMsr-49, Project C-27, BTL, Feb. 
19, 1942. Div. 13-203.2-MI 

Chapter 14 

1. The final report on Project C-36 was presented in three 
sections as follows: 

The Fundamentals of Panoramic Reception, Estill I. Green, 
OSRD 1224, OEMsr-357, BTL, Jan. 20, 1943. 

Div. 13-203.1-MI 

Improved Panoramic Receiver, Part I, E. R. Taylor, OSRD 
909, OEMsr-357, BTL, Sept, 1, 1942. Div. 13-203.3-MI 
Improved Panoramic Receiver, Part II, E. R, Taylor, OSRD 
909, OEMsr-357, BTL, Jan. 22, 1943. Div. 13-203.3-M2 

Chapter 15 

1. Receiver for Pulse Signals, OSRD 1021, OEMsr-311, 
Project C-39, BTL, Nov. 2, 1942. Div. 13-203.4-Ml 

Chapter 16 

1. Study of Interference Generation, D. Iv. Gannett, OEMsr- 
262, Project C-56, BTL, Sept. 22, 1942. Div. 13-204-M4 

2. Effectiveness of Various Audio-Frequency Noises in Mask¬ 

ing Speech, D. K. Gannett, OEMsr-626, BTL, Aug. 25, 

1942. Div. 13-204-M3 

3. Tests of Effect of Interference on Radio Telegraph Reception, 
D. Iv. Gannett, BTL, May 27, 1942. 

4. Effect of Resistance Noise on the Intelligibility of Telegraph 
Signals and Speech, D. Iv. Gannett, BTL, June 2, 1942. 

5. Energy-Frequency Distribution in a Noise-Modulated FM 
Wave, D. Iv. Gannett, BTL, July 28, 1942. 

6. Effect of Resistance Noise on Radio Reception, D. Iv. 
Gannett, BTL, June 8, 1942. 

Chapter 17 

1. Developmental Model of an Interference Generator for the 
2 to 20 Megacycle Spectrum, Madison Cawein, OEMsr-89, 
Project C-25, Farnsworth Television and Radio Corpo¬ 
ration, May 1, 1942. Div. 13-204. Ml 


CONFIDENTIAL 






206 


BIBLIOGRAPHY 


2. Developmental Model of an Interference Generator for the 

15 to 30 Megacycle Spectrum, Albert Preisman, OEMsr-285, 
Project C-26, International Telephone and Radio Labo¬ 
ratories, June 19, 1942. Div. 13-204-M2 

3. Speech arid Hearing, Harvey Fletcher, D. Van Nostrand 
Co., Inc., 1929. 

4. “Sound Measurement Objectives and Sound Level Meter 
Performance,” J. M. Barstow, The Journal of the Acoustical 
Society of America, Vol. 12, July 1940, pp. 150-166. 

5. “Measurement of Telephone Noise and Power Wave 
Shape,” J. M. Barstow, P. W. Blye, and H. E. Kent, 
Electrical Engineer, Vol. 54, December 1935, pp. 1307-1315. 

6. “Auditory Patterns,” Harvey Fletcher, Reviews of Modern 
Physics, Vol. 12, January 1940, pp. 47-65. 

7. Tests of Effect of Interference on Radio Telegraph Recep¬ 
tion, D. K. Gannett, 3430-DIvG-EEL, BTL, May 27, 1942. 

8. “Articulation Testing Methods,” Harvey Fletcher and 
J. C. Steinberg, Bell System, Technical Journal, Vol. VIII, 
1929, pp. 806-853. 

Chapter 18 

Note: Project number in parentheses indicates a continua¬ 
tion of a previous contract. Thus, Project C-44 is a continuation 
of Project C-9, C-49 of C-13, etc. 

1. Radio Transmission Handbook, Frequencies 1,000 to 30,000 

Kc, OSRD 385, Project C-9(44), National Bureau of 
Standards, Jan. 1, 1942. Div. 13-200.2-MI 

2. Coordinated Study of Ionospheric Transmission and Direc¬ 

tion Errors at High Radio Frequencies, T. R. Gilliland, 
Project C-13(49), National Bureau of Standards, Apr. 26, 
1943. Div. 13-101.3-MI 

3. Report on College ( Alaska) Observatory, March 1941 through 

June 1943, E. H. Bramhall and S. L. Seaton, OSRD 783, 
NDCrc-144, OEMsr-200, Project C-14(48), Department 
of Terrestrial Magnetism, Carnegie Institution of Wash¬ 
ington, Aug. 15, 1942. Div. 13-205-M3 

4. Ionospheric and Field-Intensity Measurements, R. A. Helli- 

well, OSRD 786, OEMsr-227, Project C-20(47), Stanford 
University, July 5, 1942. Div. 13-205-MI 

5. Ionospheric and Field Intensity Measurements, T. Parkin¬ 

son, OSRD 715, OEMsr-378, Project C-22(46), Louisiana 
State University, July 10, 1942. Div. 13-205-M2 

6. Supplement to Radio Transmission Handbook, J. H. Dell¬ 
inger, Newbern Smith, and C. O. Marsh, OSRD 666, 
Project C-44, National Bureau of Standards, June 1, 1942. 

Div. 13-200.2-M2 

7. Ionosphere Measurements Research, G. W. Kenrick, OSRD 

1883, OEMsr-632, Project C-45, University of Puerto 
Rico, June 30, 1943. Div. 13-205-M4 

8. Ionospheric Measurement Research, T. Parkinson, OSRD 

1662, OEMsr-573, Project C-46, Louisiana State Uni¬ 
versity, July 10, 1943. Div. 13-205-M5 

9. Measurement of Radio Field Intensities and Ionospheric 
Heights, R. A. Helliwell, OSRD 1718, OEMsr-590, Project 
C-47, Stanford University, July 26, 1943. Div. 13-205-M6 

10. Results of Ionospheric and Signal-Intensity Measurements 

Affecting Radio Wave Propagation, E. H. Bramhall and 
S. L. Seaton, OSRD 1831, OEMsr-558, Project C-48, 
Department of Terrestrial Magnetism, Carnegie Institu¬ 
tion of Washington, Sept. 1, 1943. Div. 13-205-M8 

11. Radio Transmission Measurements, T. R. Gilliland, New¬ 
bern Smith, and F. N. Gracely, OSRD 1736, Project C-49, 
National Bureau of Standards, Aug. 1, 1943. 

Div. 13-205-M7 

12. Correlation of Solar and Geomagnetic Observations with Con¬ 
ditions of the Ionosphere, A. H. Shapley and H. W. Wells, 


OSRD 1890, OEMsr-594, Project C-53, Department of 
Terrestrial Magnetism, Carnegie Institution of Wash¬ 
ington, Sept. 18, 1943. Div. 13-205-M9 

13. “2-Mc Sky-Wave Transmission,” J. A. Pierce, Electronics, 
Vol. 19, No. 5, May 1946, pp. 146-153. (See also J. A. Pierce, 
Proceedings of the Institute of Radio Engineers, July 1938, 
p. 892.) 

14. Field Equipment for Ionosphere Measurements, T. R. Gilli¬ 
land and A. Hoyt Taylor, Bureau of Standards research 
paper RP 1384. 

15. “Trends of Characteristics of the Ionosphere for Half a 
Sunspot Cycle,” Newbern Smith, T. R. Gilliland, and 
Samuel S. Kirby, Journal of Research, National Bureau of 
Standards, RP 1159, Vol. 21, 1938, p. 835. 

Chapter 20 

1. Aircraft Facsimile System, Charles J. Young, OSRD 481, 
NDCrc-88, Project C-8, RCA, Feb. 7, 1942. 

Div. 13-206.2-MI 

Chapter 21 

1. Ultra High Speed “Flash” Telegraphy, J. C. Steinberg, 
D. W. Farnsworth, and R. W. Buntenbach, OSRD 535, 
OEMsr-50, Project C-28, BTL, Mar. 31, 1942. 

Div. 13-206.3-Ml 

2. “Romac High Speed System,” Wireless World, London, 
July 1935, p. 197. 

3. Instructions for Operation of the Flasher, British Model, 
OSRD 488. 

4. General Research in the Field of Communications, M. L. 

Almquist, OSRD 5741, OEMsr-1413, Project 13-104, 
BTL, Aug. 31, 1945. Div. 13-200-MI 

Chapter 22 

1. “Bridge Stabilized Oscillator,” L. A. Meacham, Proceedings 
of the Institute of Radio Engineers, Vol. 26, 1938, p. 1278. 

2. “Constant Frequency Oscillator,” F. B. Llewellyn, Pro¬ 
ceedings of the Institute of Radio Engnieers, Vol. 19, De¬ 
cember 1931. 

3. “Limits of Inherent Frequency Stability,” Walter Van B. 
Roberts, RCA Review, Vol. 4, 1940, p. 478. 

4. Theory and Design of Valve Oscillators, H. A. Thomas, 
Chapman and Hall, London, 1939. 

5. “Frequency Variations of Valve Oscillators,” D. F. Mar- 
tyn, Wireless Engineer, Vol. 7, 1930, p. 3. 

6. “Power Losses in Insulating Materials,” E. T. Hoch, Bell 
System Technical Journal, Vol. I, November 1922, p. 100. 

7. Radio Engineering, F. E. Terman, Second Edition, 
McGraw-Hill Book Co., 1937. 

8. “Dielectric Constant of Air,” Isaac A. Koga, Electro¬ 
technical Journal, Vol. 4, No. 5. 

9. Frequency Stabilized Oscillator, Part I, W. R. Ferris, H. L. 

Donley, and others, OSRD 1412, OEMsr-690, Project 
C-59, RCA, Mar. 31, 1943. Div. 13-206.4-MI 

10. Master Oscillator, Part II, J. G. Beard, OSRD 1412, 
OEMsr-690, Project C-59, RCA, Mar. 31, 1943. 

Div. 13-206.4-M3 

11. Master Oscillator, J. G. Beard, OEMsr-690, Project C-59, 

RCA, Mar. 31, 1943. Div. 13-206.4-M2 

Chapter 23 

1. Pick-up Tube for Reconnaissance Television, G. A. Morton 
and G. L. Ivrieger, OSRD 1343, OEMsr-706, Project C-62, 
Mar. 3, 1943. Div. 13-206.5-MI 


1 


(’OX FI DKNTTAL 









BIBLIOGRAPHY 


207 


Chapter 24 

1. Sound Recording on Magnetic Materials, C. B. Jones and 
D. B. Davies, OSRD 3099, OEMsr-833, Project C6-69, 
Brush Development Company, Dec. 31, 1943. 

Div. 13-206.6-M2 

2. A Survey of Magnetic Recording, Roland A. Lynn, OSRD 
3027, OEMsr-1086, Project 13.3-87, NBC, Oct. 30, 1943. 

Div. 13-206.0-Ml 

Chapter 25 

1. Possibilities of Substitute for Natural Quartz for Frequency 
Control at High and Ultra-High Frequencies, OSRD 287, 
OEMsr-120, Project C-29, Brush Development Com¬ 
pany, Dec. 6, 1941. Div. 13-207.1-Ml 

Chapter 26 

1. Shielding for Diathermy, Warren C. Stoker and William 
W. Seifert, OSRD 3013, OEMsr-225, Project C-31, 
Rensselaer Polytechnic Institute, Oct. 29, 1943. 

Div. 13-207.2-MI 

Chapter 27 

1. Device for Locating Faults in Wire Lines. Capacitance 

Bridge and Portable Locator, Howard P. Corwith, OSRD 
634, OEMsr-316, Project C-37, Western Union Telegraph 
Company, June 15, 1942. Div. 13-207.31-M2 

2. Capacitance Bridge. Theory, Operation and Maintenance, 
Howard P. Corwith, OEMsr-316, Project C-37, Western 
Union Telegraph Company, June 15, 1942. 

Div. 13-207.31-M 3 

3. Oscillator-Keyer and Portable Fault Locator. Theory, Oper¬ 

ation and Maintenance, Howard P. Corwith, OEMsr-316, 
Project C-37, Western Union Telegraph Company, June 
15, 1942. Div. 13-207.31-MI 

Chapter 23 

1. Storage Batteries for Cold Climates, C. H. Endress and 
E. M. Sutherland, OSRD 948, OEMsr-420, Project C-40, 
Willard Storage Battery Company, Aug. 14, 1942. 

Div. 13-207.4-MI 

2. Storage Batteries for Cold Climates arid One-Cycle Cells, 

C. H. Endress and E. M. Sutherland, OSRD 1832, OEMsr- 
420, Project C-40, Willard Storage Battery Company, 
Aug. 16, 1943. Div. 13-207.4-M2 

Chapter 29 

1. An Electrical Cancellation and Indicating System, H. H. 
Beverage, OSRD 1408, OEMsr-748, Project C-67, RCA, 
Apr. 6, 1943. Div. 13-207.5-MI 


Chapter 30 

1. Laying Communication Wires from Planes, J. J. Gilbert, 

OSRD 1651, OEMsr-879, Project C-72, BTL, June 30, 
1943. Div. 13-207.32-M2 

2. Trial of Wire Laying by Aircraft, J. J. Gilbert, OSRD 1513, 
OEMsr-879, Project C-72, BTL, May 10, 1943. 

Div. 13-207.32-MI 


3. “Air Laid Field Wires Using Carpet-Type Packs,” L. R. 
Montfort, System Engineering for Army Air Forces Com¬ 
munication, Part III, Memorandum No. 7, OSRD 4295, 
OEMsr-1018, Project C-79, BTL, July 1, 1944. 

Div. 13-200.1-M3 

4. “Laying Field Telephone W ire by Airplane,” P. W. Blye, 
Bell Laboratories Record, May 1945, p. 145. 

Chapter 31 

1. General Research in Field of Communication, M. L. Alm- 
quist, OSRD 5741, OEMsr-1413, Project 13-104, BTL, 
Aug. 31, 1945. Div. 13-200-Ml 

Chapter 32 

1. “Investigation of Engine Noise Problems,” V. T. Callahan 

and D. F. Seacord, Systems Engineering for Army Air 
Forces Communication, Part II, Memorandum No. 6B, 
OSRD 1925, OEMsr-1018, Project C-79, BTL, Oct, 1, 
1943. Div. 13-200.1-M2 

2. Investigation of Radio Noise Generation in Aircraft Electrical 
Machinery, C. W. Frick, OSRD 6631, OEMsr-1475, 
Project 13-117, GE, Jan. 2, 1945. Div. 13-202.22-M3 

3. Interference Measurements, Measurement of, 150 Kc to 20 
Me, for Components and Complete Assemblies, JAN-I-225, 
June 14, 1945. 

4. Investigations of the Measurement of Noise, W. J. Bartik, 
T. H. Bonn, and others, OSRD 6358, OEMsr-1478, 
Project 13-123, University of Pennsylvania, Moore School 
of Electrical Engineering, Oct. 31, 1945. 

Div. 13-202.22-M2 

5. Final Report of Interference Reduction Committee, K. C. 
Black, Nov. 27, 1945. 

6. Note on Interference Reduction Committee for Keith Ilenney, 

Editor, Division 13 Summary Report, L. W r . Baldwin, 
Apr. 15, 1946. Div. 13-202.1-M2 

Chapter 33 

1. Investigation of Principles Underlying the Maximizing of 
Communication Intelligibility, W. J. Cunningham, OSRD 
6280, OEMsr-144, Project 13-110, Problem No. 4, Cruft 
Laboratory, Harvard University, Dec. 1, 1945. 

Div. 13-100-M2 

2. Stagger-Tuned IF Amplifiers, H. Wallman, OEMsr-262, 
RL Report 524, MIT, Feb. 23, 1944. Div. 14-241.32-M4. 

3. Transient Response of Multistage IF Amplifiers, Twiss, 
Report T-1834, Telecommunications Research Establish¬ 
ment, Great Britain. 

Chapter 34 

1. Measurements of Magnetic Properties of Ferrite Core Ma¬ 
terials, Orin Cornett and S. W r oodsum, OSRD 6280, 
OEMsr-1441, Project 13-110, Problem No. 8, Cruft Labo- 
tory, Harvard University, Dec. 1, 1945. Div. 13-100-M2 

Chapter 35 

1. Aircraft Transmitter Antenna Power, Impedance, and Tun¬ 
ing Network Survey, E. C. Easton, R. E. Kirkland, and 

S. E. Parker, OSRD 6280, OEMsr-1441, Project 13-110, 
Problem No. 6, Cruft Laboratory, Harvard University, 
Dec. 1, 1945. ' Div. 13-100-M2 




CONFIDENTIAL 









OSRD APPOINTEES 


Division 13 

Chief 

C. B. Jolliffe (December 1942 to May 1945) 
Haraden Pratt (May 1945 to May 1946) 

Deputy Chief 
Iv. C. Black 

Technical Aides 

,1. L. Allison J. F. McClean 

C. F. Dalziel A. F. Murray 


208 


K. C. Black 
0. E. Buckley 
J. H. Dellinger 
W. L. Everitt 
G. 0. Fick 


Section 13.1 
Section 13.2 
Section 13.3 
Section 13.4 
Section 13.5 
Section 13.6 


Members 

R. H. George 
C. H. G. Gray 
A. Hazeltine 
if. A. Hutchinson 
C. M. .Jansky 
L. F. -Tones 


D. G. Little 
R. Iv. Potter 
H. Pratt 
C. A. Priest 
F. M. Ryan 


L. F. Jones 
J. H. Dellinger 
R. Iv. Potter 

C. A. Priest 
H. Pratt 

D. G. Little 


Section Heads 

Direction Finding 

Radio Propagation Problems 

Speech Secrecy 

Special Communications Problems 
Precipitation Static 
Miscellaneous Projects 


Consultants 

L. Y. Berkner E. D. Blodgett 

H. H. Beverage D. G. Little 

R. K. Potter 


Interference Reduction Committee 


K. C. Black 
II. D. Doolittle 
R. G. Fluharty 


A. Hazeltine 
-T. C. R. Licklider 
C. T. Morgan 


A. F. Murray 
S. S. Stevens 
0. W. Towner 


CONFIDENTIAL 






CONTRACT NUMBERS, CONTRACTORS, AND SUBJECTS OF CONTRACTS 


Contract Name and Address Refer to 

Number of Contractor Subject Chapter 


NDCrc-75 

NDCrc-141 

NDCrc-177 

NDCrc-88 

Project C-9 

NDCrc-191 

OEMsr-200 

OEMsr-227 

OEMsr-92 

OEMsr-378 

OEMsr-32 

OEMsr-89 

OEMsr-285 

OEMsr-49 

OEMsr-50 

OEMsr-120 

OEMsr-174 

OEMsr-225 

OEMsr-357 

OEMsr-316 

OEMsr-311 

OEMsr-420 

OEMsr-678 

OEMsr-442 

Project C-44 

OEMsr-632 

OEMsr-573 

OEMsr-590 

OEMsr-558 

Project C2-49 

OEMsr-594 


Radio Corporation of America 
Camden, New Jersey 
General Radio Company 
Boston, Massachusetts 
Western Electric Company, Incorporated 
New York, New York 
Radio Corporation of America 
Camden, New Jersey 
National Bureau of Standards 
Washington, D. C. 

Westinghouse Electric and Manufactur¬ 
ing Company 
Baltimore, Maryland 
Carnegie Institution of Washington 
Washington, D. C. 

Stanford University 

Stanford University, California 
Oregon State College 
Corvallis, Oregon 
Louisiana State University 
University, Louisiana 
Radio Corporation of America 
Camden, New Jersey 
Farnsworth Television Company 
Fort Wayne, Indiana 
International Telephone and Radio 
Laboratories 
New York, New York 
Western Electric Company, Incorporated 
New York, New York 
Western Electric Company, Incorporated 
New York, New York 
Brush Development Company 
Cleveland, Ohio 
Jansky and Bailey 
Washington, D. C. 

Rensselaer Polytechnic Institute 
Troy, New York 

Western Electric Company, Incorporated 
New York, New York 
Western Union Telegraph Company 
New York, New York 
Western Electric Company, Incorporated 
New York, New York 
Willard Storage Battery Company 
Cleveland, Ohio 
University of New Mexico 
Albuquerque, New Mexico 
Radio Corporation of America 
Camden, New Jersey 
National Bureau of Standards 
Washington, D. C. 

University of Puerto Rico 
Rio Piedras, Puerto Rico 
Louisiana State University 
University, Louisiana 
Stanford University 

Stanford University, California 
Carnegie Institution of Washington 
Washington, D. C. 

National Bureau of Standards 
Washington, D. C. 

Carnegie Institution of Washington 
Washington, D. C. 


2-3,000 me receivers, 2 IF amplifiers, 1 signal generator. 
UHF field intensity measuring equipment. 

RF generator. 

Aircraft facsimile. 

Radio wave handbook. 

2-3,000 me transmitters. 

Coordination of DF-ionospheric measurements. 
DF-ionospheric measurements. 

Precipitation static reduction. 

DF-ionospheric measurements. 

3,000 me communication field research. 

Radio interference generator. 

Radio interference generator. 

Panoramic receiver with moving screen indicator. 
Extra-high speed “flash” telegraphy. 

Substitute for quartz crystals. 

Mobile transmitter radio field survey. 

Shielding for diathermy. 

Improved panoramic receiver. 

Device for locating faults in wire lines. 

Panoramic receiver for pulse signals. 

Storage batteries for cold climates. 

Precipitation static reduction. 

Voice communication on CM waves. 

Supplement to radio wave handbook. 

Ionospheric measurement research. 

Ionospheric measurement research. 

Ionospheric measurement research. 

Ionospheric measurement research. 

Radio transmission measurements. 

Correlation of solar and geomagnetic observations with 
conditions of the ionosphere. 


2 

7 


2 


7 

6 


o 


b 

6 


3 


6 


2 


5 

5 

4 

7 

8 
1 
8 
4 
8 
4 
8 


3 


2 

6 

6 

6 

b 

b 

b 

b 


209 










CONTRACT NUMBERS, CONTRACTORS, 


AND SUBJECTS OF CONTRACTS 


( Continued ) 


Contract Name and Address Refer to 

Number of Contractor Subject Chapter 


OEMsr-626 

OEMsr-690 

OEMsr-706 

OEMsr-679 

OEMsr-748 

OEMsr-893 

OEMsr-S33 

OEMsr-848 

OEMsr-879 

OEMsr-1018 

OEMsr-1010 

OEMsr-1086 

OEMsr-1413 

OEMsr-1441 
Problem No. 2 

OEMsr-1441 
Problem No. 4 

OEMsr-1441 
Problem No. 6 

OEMsr-1441 
Problem No. 8 

OEMsr-1441 
Problem No. 9 

OEMsr-1475 

OEMsr-1478 


Western Electric Company, Incorporated 
New York, New York 
Radio Corporation of America 
Camden, New Jersey 
Radio Corporation of America 
Camden, New Jersey 
Purdue Research Foundation 
Lafayette, Indiana 
Radio Corporation of America 
Camden, New Jersey 
United Air Lines 
Chicago, Illinois 
Brush Development Company 
Cleveland, Ohio 
State College of Washington 
Pullman, Washington 
Western Electric Company, Incorporated 
New York, New York 
Western Electric Company, Incorporated 
New York, New York 
Jansky and Bailey 
Washington, D. C. 

Radio Corporation of America 
Camden, New Jersey 
Western Electric Company, Incorporated 
New York, New York 
Central Communications Research 
Cruft Laboratory, Harvard University 
Cambridge, Massachusetts 
Central Communications Research 
Cruft Laboratory, Harvard University 
Cambridge, Massachusetts 
Central Communications Research 

Cruft Laboratory, Harvard University 
Cambridge, Massachusetts 
Central Communications Research 

Cruft Laboratory, Harvard University 
Cambridge, Massachusetts 
Central Communications Research 

Cruft Laboratory, Harvard University 
Cambridge, Massachusetts 
General Electric Company 
Schenectady, New York 
University of Pennsylvania 
Philadelphia, Pennsylvania 


Study of interference generation. 

Frequency stabilized master oscillator. 

Reconnaissance television. 

Study of effect of aircraft surface treatment on electrical 
charges. 

Electrical cancellation and indicating system. 
Precipitation static research. 

Sound recording on magnetic materials. 

Precipitation static exploration. 

Wire laying by aircraft. 

Systems engineering for AAF communications. 

Measure of effect of hills and trees on radio propagation. 
A survey of magnetic recording. 

General research in the field of communications. 

Study of FM versus AM. 

Investigation of principles underlying maximizing com¬ 
munications intelligibility. 

Aircraft transmitter antenna tuning network survey. 

Measurement of magnetic properties of ferrite core 
materials. 

Microwave survey 


Investigation of radio noise generation in aircraft elec¬ 
trical machinery. 

Investigation of radio interference conditions on ship¬ 
board up to 1150 me. 


5 

7 

7 
3 

8 
3 

7 
3 

8 
1 
1 

7 

8 
1 

8 

8 

8 


8 

8 


I 


CONFIDENTIAL 


V 


210 









SERVICE PROJECT NUMBERS 

The projects listed below were transmitted to the Executive Secretary, 

NDRC, from the War or Navy Department through either the War 

Department Liaison Officer for NDRC or the Office of Research and 

Inventions (formerly the Coordinator of Research and Development), 

Navy Department. 

Service 


Refer to 

Project Number 

Subject 

Chapter 

AC-54 

Systems engineering for AAF communications. 

1 

AC-54 

Measure of effect of hills and trees on radio propagation. 

1 

AC-230.03 

Study of FM versus AM. 

1 

AC-238.03 

Investigation of radio noise generation in aircraft electrical machinery. 

8 

AC-238.07 

Aircraft transmitter antenna tuning network survey. 

8 

SC-2 

Precipitation static reduction. 

3 

SC-2 

Precipitation static reduction. 

3 

SC-2 

Study of effect of aircraft surface treatment on electrical charges. 

3 

SC-2 

Precipitation static research. 

3 

SC-2 

Precipitation static exploration. 

3 

SC-13 

2-3,000 me receivers, 2 IF amplifiers, 1 signal generator. 

2 

SC-13 

3,000 me communication field research. 

2 

SC-13 

Voice communication on CM waves. 

2 

SC-15 

Aircraft facsimile. 

7 

SC-16 

UHF field intensity measuring equipment. 

7 

SC-18 

Frequency stabilized master oscillator. 

7 

SC-19 

Radio interference generator. 

5 

SC-19 

Radio interference generator. 

5 

SC-19 

Study of interference generation. 

5 

SC-20 

Storage batteries for cold climates. 

8 

SC-23 

Device for locating faults in wire lines. 

8 

SC-24 

Substitute for quartz crystals. 

8 

SC-42 

Wire laying by aircraft. 

8 

NA-110 

Reconnaissance television. 

7 

NS-365 

Investigation of principles underlying maximizing of communications intelligibility. 

8 

NS-368 

Investigation of radio interference conditions on shipboard up to 1150 me. 

8 


CONFIDENTIAL 


211 















INDEX 


The subject indexes of all STR volumes are combined in a master index printed in a separate volume. For access 
to the index volume consult the Army or Navy Agency listed on the reverse of the half-title page. 


AAF communications systems, 3-6 
see also Communications systems 
AFC 

see Automatic frequency control 
Air Warning Service 

choice of frequency band, 4 
communications systems studies, 3 
Aircraft antenna impedance studies, 
200-201 

Aircraft electric machinery noise, 193 
Aircraft facsimile system, 141-143 
Aircraft lightning strike hazard, 45 
Aircraft static charge 

see Precipitation static studies 
Aircraft surface treatment for static 
reduction, 51-52 

Aluminum phosphate crystals for fre¬ 
quency control, 174 
Amplitude modulation systems 
band space requirements, 13, 14 
comparison with f-m systems, 9 
crosstalk in multiplexed transmis¬ 
sion, 13 

effect of mistuning, 8 
in directional microwave telephone 
transmitter, 36 
response to random noise, 8 
AN/ARQ-9 radio set, scanning receiver- 
research, 108-116 
Antennas 

directive antennas for interference 
protection, 15 

effect of height on microwave propa¬ 
gation, 20 

effect on frequency stability of micro- 
wave transmitters, 22 
for directional microwave telephone, 
34 

for mobile radiotelephone communi¬ 
cations, 7 

for noise meters, 194 
for omnidirectional microwave tele¬ 
phone receiver, 27 

for radar pulse scanning receiver, 110 
impedance studies, 200-201 
ionosphere recording antennas, 132 
lightweight antenna mast, 4 
loading range for specific transmit¬ 
ters, 200-201 
site studies, 4 

AN/TRC-5 multiplying method, 12 
Atmospheric disturbances 

in mobile communications systems, 7 
ionosphere studies, 131-133 
precipitation static studies, 45-60 
Attenuation, microwave 
in plywood, 26-27 
in rectangular guide at 9.8 cm, 25 
mutual inductance type attenua¬ 
tors, 139 

of radio waves by absorption, 132 
panoramic receiver input attenuator, 
89 


Audio limiters 
see Limiters 

Aurora displays, effect on radio trans¬ 
mission, 133 

Automatic frequency control 
cavity stabilization, 16 
effect on frequency stability, 158 
microwave transmitters, 23 
Automatic volume control 
panoramic receivers, 66-67 
radar pulse scanning receiver, 112 
Azimuth indicating system, radar pulse 
scanning receiver, 110 

Balancers for noise reduction 
in a-m systems, 8 
in f-m systems, 9 

reduction of diathermy interference, 
175 

Band space requirements 
a-m systems, 12 
f-m systems, 8, 9, 11 
jamming systems, 119 
multiplex microwave transmission, 14 
pulsed emission, 12 
simplex microwave transmission, 13- 
14 

Band-pass amplifiers, response analy¬ 
sis, 196-197 

Batteries, low temperature, 181-182 
BC-603 receivers, 9 
Beam deflection converter receiver, 25 
Beam deflection methods for panoramic 
indicators, 73, 78 
Bendix Aviation Corporation, 51 
Bendix charge dissipator, 48, 58 
Blanking for panoramic reception, 82, 
100 

Block-and-squirter system for control 
of precipitation static, 59-60 
Bolometer bridge, 31 
Boonton Type 100-A Q meter, 198, 201 
Bridge stabilized oscillator, 158 
Bridges for wire fault location, 176-177 
Butterfly circuit, 137-139 
Buzzer noise for jamming speech sig¬ 
nals, 119 

C-25 jammer, 127 
C-26 jammer, 124-127 
C-59 frequency-stabilized master oscil¬ 
lator, 154-161 
circuit studies, 155-158 
humidity effects, 159 
importance of high Q, 159 
stability tests, 154 
temperature effects, 159 
tube studies, 154, 156 
tuning methods, 159-161 
Camouflage paints, effect on aircraft 
static, 51-52 

Cancellation and indicating system for 
complex waves, 183-190 


alarm and indicating system, 187-188 
approximation by rectangular pulses, 
183-184 

complex wave generator, 184-185 
electrical cancellation, 188 
monitor oscilloscope, 187-188 
multiple station monitoring, 190 
panoramic reception, 188-189 
use as coding signal generator, 189- 
190 

Carnegie Institution of Washington, 
field intensity and ionosphere 
studies, 131 
Carrier frequency 

effect of doppler shift on bandwidth 
requirements, 13 

number of channels in a given band 
space, 13 
selection, 13 
stabilization, 13, 15, 16 
Cathode-ray tubes as panoramic indi¬ 
cators 

amplitude vs. frequency diagrams, 78 
azimuth vs. frequency diagrams, 79 
choice of cathode ray screen type, 76 
facsimile type diagrams, 77 
for long scale 3- to 10-mc panoramic 
receiver, 99-100 
long persistence screens, 76 
moving screen tube, 77, 92-93 
recording radio telegraph signals, 77- 
78, 83-85 
resolution, 76-77 

signal presence vs. frequency dia¬ 
grams, 73-76 

Cavity for 1290-CT r-f oscillator tube, 
37 

Cavity monitor, r-f generator, 38-39 
Cavity oscillator for directional tele¬ 
phone, 35 

Cavity stabilization scheme for afc, 16 
CFVD modulation (continuous fre¬ 
quency, variable duration) pulse 
system, 22, 23, 36 
r-f generator, 41 

Charge dissipators, aircraft static 
flame discharge, 48 
radioactive cups, 48 
trailed wire type, 47, 4S, 55-56 
Charge indicator, aircraft static, 56 
Charging rates of aircraft camouflage 
paints, 51-52 
Clippers 
see Limiters 

Coding signal generator, electrical can¬ 
cellation system, 189-190 
Colpitts oscillator circuit, frequency 
stability studies, 155-156 
Communication uses for panoramic re¬ 
ceivers, 86 

Communications systems 

see also Microwave communications 
systems, design factors 




214 


INDEX 


directional microwave telephone, 32- 
36 

facsimile system, 141-143 
flash telegraphy systems, 144-1.53 
mobile radio telephone communica¬ 
tions, 6-7 

omnidirectional microwave tele¬ 
phone, 27-32 
Complex impedance 
see Impedance 

Complex waveform approximation 
using rectangular pulses, 183 
Complex-wave generator, commutator 
type, 184 

Compression amplifier, moving-screen 
panoramic receiver, 91, 95 
Continuous waves for communications, 
advantages, 15 

Core materials, magnetic properties 
see Magnetic properties of ferrite core 
materials 

Corona discharge from planes 
current measurements, 58-59 
voltage induction effects, 47 
CR-301 microwave receiver, 25 
Cross-band signaling, 14-16 
Crosstalk in multiplexed transmission, 
13 

Crystal control for ultra high frequen¬ 
cies, 13 

Crystal converter microwave receivers 
disadvantages, 21-22 
frequency stability, 24 
input noise characteristics, 24 
sensitivity, 24 

Crystals, for frequency control, syn¬ 
thetics, 173-174 

CuZn ferrite, magnetic permeability, 
199 

C-W Morse telegraph operation at 
v-h-f, 4 

D 160127 coaxial line tube, 24, 139 
Dampers, for reducing precipitation 
static, 47 

Deaf persons aided by visible speech 
system, 177 

Diathermy equipment shielding, 175 
Dielectric constant of plywood, 26 
Direction finding, panoramic receivers, 
87 

Directional microwave telephone 
antenna system, 34 
field tests, 36 
power supply, 35 
requirements, 32-34 
r-f unit, 35 
transmitter, 36 

Discharge wires for static dissipation, 
48 

Discriminator, f-m systems, effect on 
pulse interference, 9, 10 
Dry cells for low temperature use, 
181-182 

Duty cycle, panoramic receiver, 109 
Dynamotor noise studies, 193 

Electrolytic recorder, facsimile, 141 


Electron gun, television, 162, 165 
Electrostatic charging rates of aircraft 
in flight, 58-59 
ENR (excess noise ratio) 
definition, 31-32 

directional microwave telephone re¬ 
ceiver, 35 

omnidirectional microwave telephone 
receiver, 31 

ENSI (equivalent noise sideband input) 
definition, 21 
of crystal receivers, 21 
microwave receivers, 24-25 
Erasing magnetic recordings, 144 
E-region ionization, effect on radio 
transmission, 133 

Facsimile system, aircraft, 141-143 
Facsimile-type diagrams, panoramic re¬ 
ceivers, 77 

Fault location in wire lines, 176 
Ferrite, magnetic cores, 198-199 
permeability, 198 
specific loss factor, 198 
Fiberglas, use in floating insulated wire, 
192 

Fibrite as storage battery insulator, 181 
Field intensity studies 

ionosphere studies, 131-132 
u-h-f measuring equipment, 137-140 
Field wire, laying by airplane over 
jungle, 191 

Flame discharger for precipitation static 
reduction, 48 

Flash telegraphy system, 144-153 
advantages, 144 
booster amplifier, 148 
distortion from tape recording, 150- 
152 

flasher design, 147-148 
magnetic tape recording system, 144- 
145, 152-153 

radio circuit requirements, 153 
starting pulse problems, 146-147 
tape drive mechanism, 148 
teletypewriter and perforated tape 
methods, 145 

Floating wire cable on water, methods, 
192 

F-M systems 

see Frequency modulation systems 
Fortisan, use for floating insulated wire, 
192 

Free space field, field intensity studies, 
7 

Frequency allocation 

Air Warning Service radio circuits, 4 
mobile radiotelephone communica¬ 
tion, 6 

panoramic receivers, 64-65 
Frequency control, automatic, 16, 23, 
158 

Frequency determination, panoramic 
receivers, 81-82 
beating oscillators, 82 
calibrated signal generator, 81 
electrical markings, 81 
fixed frequency markings, 81 


Frequency difference compensator, 
microwave telephone, 36 
Frequency modulation systems, 8-10, 
23 

band space requirements, 12-14 
comparison with a-m, 9-10 
crosstalk in multiplexed transmis¬ 
sion, 13 

jamming, 125, 126 
narrow band and wide band systems, 
8, 9 

overmodulation, 9 
response to random noise, 8-10 
signal-to-noise ratio improvement, 24 
sweep oscillators for panoramic re¬ 
ception, 64, 106 

Frequencies for communication systems 
sec Band space requirements 
Frequency spectrum of precipitation 
static, 48 

Frequency stability 
circuit studies, 155-157 
humidity effects, 159 
importance of high Q, 159 
microwave telephone oscillator, 32 
microwave transmitter, 22-25 
stability tests, 154 
temperature effects, 159 
tube studies, 154-156 
2-20 me oscillators, 154-159 
Frequency-division multiplexing, 12-13 
Frost crystals, effect on airplane static 
charge, 51-52 

Geomagnetic phenomena, ionosphere 
studies, 131-133 

Glass wool as storage battery insulator, 

181 

Glide-path indication, use of panoramic 
receivers, 87 

G-meter for vertical acceleration mea¬ 
surements, 50 

Gradient indicator, aircraft, electro¬ 
static, 56 

Grounding, effect on shielded room, 175 
Gum-bichromate process for television 
tubes, 164 

Hartley oscillator 

for frequency stabilized 2-20 me oscil¬ 
lator, 154 

frequency stability studies, 156 
in moving-screen panoramic receiver, 
90 

Heterodyne methods in panoramic re¬ 
ception, 65 

Hills, obstructions to radio propaga¬ 
tion, 5, 6 

Horn antennas for microwave use, 25 
Humidity effect on oscillator stability, 
159 

Ignition disturbances in mobile com¬ 
munications system, 7 
Impedance 

aircraft antennas, 200-201 
effect of weather on antenna match¬ 
ing, 22 

measuring equipment, 201 


CONFIDENTIAL 





INDEX 


215 


plywood filled waveguide, 26 

Impulse noise for jamming speech 
signals, 119 

Indicators for panoramic receivers 
amplitude vs. frequency diagram, 78 
azimuth vs. frequency diagrams, 79 
choice of screen type, 76 
facsimile type diagram, 77 
long scale indicating equipment, 100- 
102 

mechanical indicators, 79 
moving screen tube, 77 
recording radio telegraph signals, 
77-78 

repetition rate, 76 
resolution, 76-77 
signal alarms, 80 

signal presence vs. frequency dia¬ 
grams, 73-76 

350-750 me receiver, 109-110 

Information center communications 
systems, 3 

Infrared rays for wiping out CRO 
traces, 95 

Instrument landing, use of panoramic 
receivers, 87 

Insulation, low temperature storage 
batteries, 181 

Intelligibility tests, speech transmis¬ 
sion jamming, 119-120 

Interference 

see also Jamming studies 
effect on omnidirectional microwave 
telephone, 32 
pulse interference, 8, 9 
random noise in a-m systems, 8 
random noise in f-m systems, 8-10 
reduction in AAF communications 
systems, 4, 195 

shielding for diathermy equipment, 
175 

Interference Reduction Committee, 195 

Interservice Radio Propagation Lab¬ 
oratory (IRPL), 131 

Inverse vacuum-tube voltmeter, 49 

Ionosphere studies, 131-133 
apparatus, 132-133 
field strength studies, 131-132 
solar and geomagnetic observation, 
133 

Jamming, 117 
C-25 jammer, 127 
C-26 jammer, 124-127 
criteria for successful jamming, 123- 

124 

effect on a-m and f-m receivers, 9 
efficiency of noise limiters against 
high intensity pulses, 126 
field test results, 126-127 
form and degree of modulation, 124, 
126, 127 

fundamental masking studies, 122- 
123 

interference with f-m transmission, 

125 

interference with speech transmis¬ 
sion, 119-120 


interference with telegraph transmis¬ 
sion, 120-121, 123 
military considerations, 124 
protection against jamming, 15 
resistance noise generation, 121 
tuning to enemy frequency, 124-125, 
127 

use of panoramic receivers, 87 

Jungle terrain, laying field wire by air¬ 
plane, 191 

Klystron transmitter, 14, 19, 23-24 

Latex sponge as storage battery insu¬ 
lator, 181 

Laying field wire by airplane, 191 

Leland Stanford University, field in¬ 
tensity and ionosphere studies, 
131 

Lightning strikes, aircraft, 45 
gradient warning indicator, 56 

Lighthouse tubes 

in directional microwave telephone 
oscillator, 35 

limitations for butterfly tuning cir¬ 
cuits, 139 

Limiters 

effect on jamming efficiency, 126 
for microwave telephone transmitter, 
30 

for noise reduction, 8-10, 15 
for panoramic receivers, 66-67 
for precipitation static reduction, 47 
in a-m systems, 8, 9-10 
in band-pass amplifiers, 196-197 
in f-m systems, 9 

Line mosaic pick-up tube, 162-165 
electron gun, 162, 165 
mosaic types, 163-164 
optical system, 162 
test results, 165 
tube construction, 164-165 

Lineman’s portable open circuit locator, 
177-178 

Line-of-sight path for microwave propa¬ 
gation, 19, 21 

Loading range measurements, r-f trans¬ 
mitter, 200, 201 

Long-scale panoramic receiver, 98-104 
balanced modulators, 103 
beam modulation method of indica¬ 
tion, 99 

blanking out recognized stations, 100 
cathode-ray tube, 99-100 
length and shape of scale, 98-99 
method of obtaining spiral trace, 99 
modulating amplifier, 102 
operation of indicating equipment, 
100-102 

phase shift circuits, 103 
scanning filters, 99-100 
scanning rate, 100 
synchronization amplifier, 104 

Loss factor, magnetic core materials, 
198, 199 

Louisiana State University, field inten¬ 
sity and ionosphere studies, 131 


“Magic T” waveguide section, for 
cross-band operation, 16 
Magnetic disturbances 
effect on radio transmission, 132 
radio propagation ionosphere studies, 
131-133 

Magnetic properties of ferrite cores, 
198-199 

permeability, 198, 199 
Q, 198-199' 
specific loss factor, 198 
Magnetic recording 
a-c erase, 144 
a-c recording, 145 
advantages, 166 

blanking out recognized stations in 
panoramic reception, 82 
circuits, 152 
d-c erase, 144 
d-c recording, 144-145 
distortion, 150-152 
drive mechanism for tape, 148 
flash telegraphy system, 144-145, 
152-153 

pocket-sized wire recorder, 166-169 
recording without bias, 146-147 
reproducing system, 152-153 
Magnetron 

in omnidirectional microwave tele¬ 
phone transmitter, 27 
output power, 14 

Marking arrangements, panoramic re¬ 
ceivers 

broad band 0.1- to 30-mc scanning 
receiver, 107 

moving screen receiver, 94, 95 
Masking studies, 122-123 
by clicks and pulses, 122 
by noise, 122 
code signals, 123 
pure tones, 122 

speech intelligibility, in the presence 
of noise, 120 

Mechanical indicators for panoramic 
receivers, 79 

Micromax line-drawing type of re¬ 
corder, 132 

Micrometer-tuned wavemeter, 38 
Microphone, single-button carbon, 28 
Microwave communications systems, 
design factors, 11-43 
antennas, 25, 34 
cross-band signaling, 14, 16 
i-f power, 14 

multiplexing of signals, 12-13 
omnidirectional microwave tele¬ 
phone, 27-32 

propagation studies over land and 
sea water, 19 

protection from interference and jam¬ 
ming, 15-16 
receivers, 24, 31-32 
recommendations for universal set, 
15-16 

r-f generator, 37-43 
selection and stabilization of carrier 
frequency, 13, 16 
spectrum widths, 11-12 


CONFIDENTIAL 





216 


INDEX 


transmitters, 22-24, 28-31, 36 
types of operation, 11 
use of band space, 13-14 
waveguides, 25 

Microwave frequency discriminator for 
stabilization, 16 

Mistuning, effect on receiver perform¬ 
ance, 8, 9 

MnZn ferrite, magnetic permeability, 
199 

Mobile microwave communications sys¬ 
tem 

see Directional microwave telephone 
Mobile radiotelephone communication, 
6-7 

antenna characteristics, 7 
choice of frequency, 6, 7 
field intensity studies, 7 
Modulating amplifier 

long-scale 3- to 10-mc panoramic re¬ 
ceiver, 102 

moving screen panoramic receiver, 92 
Modulation 

see also Amplitude modulation sys¬ 
tems; Frequency modulation 
systems 

cathode modulation of directional 
microwave telephone transmit¬ 
ter, 36 

CFVD system, 41 

overmodulation effect on f-m sys¬ 
tems, 9 

types of pulse modulation, 11 
Monitor cavity, r-f generator, 41-42 
Monitor circuit use of electrical can¬ 
cellation system, 190 
Monitor oscilloscope for electrical can¬ 
cellation system, 187 
Morse operation, at v-h-f, c-w, 4 
Mosaic types, line mosaic television 
pick-up tube, 163-165 
Motor noise studies, radio, 193 
Mountainous terrain, propagation 
studies, 5-6, 7 

Moving-screen panoramic receiver, 77, 
88-97 

advantages, 88 
amplifiers, 92-95 
band-pass filters, 90, 91 
cathode-ray tube, 92-93 
design principles, 88-89 
input attenuator, 89 
marking arrangement, 94, 95 
modulating amplifier, 92-93 
power supplies, 94 
pulse reception studies, 96-97 
reproduction and reading of tele¬ 
graph code, 88 

r-f and modulator circuits, 90 
scanning oscillator, 90, 94-95 
three dimensional pattern, 88 
Multipath transmission, f-m systems, 9 
Multiple responses in panoramic re¬ 
ception, 65 

Multiple selection circuits, 86 
Multiplex operation, microwave com¬ 
munications system, 11-14 
band space requirements, 14 


crosstalk problem, 13 

c-w carrier frequency stabilization, 16 

frequency division multiplexing, 12 

13 

time division multiplexing, 12-13 
Multivibrator circuit for pulse gener¬ 
ator, 42 

Narrow-band scanning, panoramic re¬ 
ception, 82-83 

National Bureau of Standards, field in¬ 
tensity and ionosphere studies, 

131 

Navigation, using panoramic radio re¬ 
ceivers, 87 

Network analysis, band-pass, 196 
Network analysis, use of complex-wave 
generator, 190 

Noise for jamming speech transmission, 
119-120 

Noise spectrum from precipitation 
static, 47, 48 
Noise limiters 
see Limiters 
Noise studies 
see also Jamming 
aircraft electrical machinery, 193 
meter studies, 193-194 
precipitation static, 45-60 
random noise, 8-10 
Nomograph for optimum filter band¬ 
width, 70 

Nylon for floating insulated wire, 192 

Omnidirectional microwave telephone, 
27-32 

antennas, 27, 32 
effect of interference, 32 
probable range with one set in air¬ 
plane, 32 
receiver, 31-32 
transmitter, 27-31 
116 me communication systems 
effect of increasing antenna height, 6 
for transmission over broken coun¬ 
try, 5 

mobile radiotelephone communica¬ 
tion, 7 

Open circuits in lines, location 

capacitance bridge method, 176-177 
errors due to resistance and leakage, 
177 

portable locator, 177-180 
Oscillators 

300 to 1,000 me, 137-139 
410 to 690 me, 111 
1,000 to 3,000 me, 139 
2,000 to 3,000-mc r-f generator, 37-43 
C-59 frequency stabilized oscillator, 
154, 159-161 

frequency stability studies, 154-161 
in directional microwave telephone, 
35 

sweep oscillators in panoramic re¬ 
ceivers, 64-65, 111 
Oscilloscopes 

see also Cathode-ray tubes as panor¬ 
amic indicators 


CONFIDENTIAL 


monitor for electrical cancellation 
system, 187 

Overmodulation effect in f-m systems, 9 
Owen bridge for measuring inductance, 
198 

Paillard Bolex 16-mm camera, 50 
Painted surfaces, precipitation static 
accumulation, 56 

PAM (pulse amplitude modulated) 
emission, 11 

Panoramic facsimile recorder, 83-85 
Panoramic receivers, 61-116 

automatic control of signal intensity, 
66-67 

blanking out recognized stations, 82 
broad band 0.1- to 30-mc scanning 
receiver, 104-108 

comparison with ordinary radio re¬ 
ception, 63-64 

electrical cancellation and indicating 
system, 188-189 
frequency allocations, 64-65 
frequency determination, 80-82 
long-scale model for 3 to 10 me range, 
98-107 

moving-screen indicator, 88-97 
narrow band scanning, 82 
panoramic indicators, 72-80 
radar pulse reception, 85-86 
range expansion, 83 
receivers without frequency sweep, 86 
recording, 83-85 

research recommendations, 107, 114, 
115 

resolution, 76-77 
sensitivity, 80 

scanning filter design and perform¬ 
ance, 67-72 

350-750 me scanning receiver, 108- 
114 

tuning jammer to enemy frequency, 
124-125 
uses, 86-87 

Parallel traces for panoramic indicator, 
74-76 

Perforated tape methods of flash teleg¬ 
raphy, 145 

Permeability, magnetic 
formula, 198 
of ferrite materials, 199 
PFM (pulse frequency modulation) 
emission, definition, 11 
Pickup tube for reconnaissance tele¬ 
vision 

see Line mosaic pickup tube 
Tr-type attentuator for panoramic re¬ 
ceiver input, 89 

PLM (pulse length modulated) emis¬ 
sion, definition, 11 
Plywood antennas 
dielectric constant, 27 
microwave losses, 26-27 
PNM (pulse number modulated) emis¬ 
sion, definition, 11 
Polarization 

horizontal vs. vertical for broken 
country, 5 





INDEX 


217 


transmission over salt water, 19-20 
Polyethelene for floating insulated wire, 
192 

Portable wire recorder, 166-169 
Power dividing wave guide sections, 

16 

Power measurement 
adjustable load method measure¬ 
ments for r-f transmitter, 200 
microwave standing wave detector, 27 
Power supply 

directional microwave telephone, 35 
in moving screen panoramic indi¬ 
cator, 94 

requirements for microwave com¬ 
munications systems, 14 
r-f generator, 43 

Power tubes for microwave systems 
klystron, 14 
magnetron, 14 

PPM (pulse position modulated) emis¬ 
sion 

advantages, 15-16 
definition, 11 

Precipitation static studies, 45-60 
airplanes for test flights, 54 
apparatus, 47, 49-50, 54-55, 58 
block-and-squirter discharge system, 
59-60 

causes of static, 57 
effect of aircraft surface treatment, 
51-52 

flight tests, 56, 58-59 
flame discharger, 48 
noise frequency spectrum, 47, 48 
nonlinear elements for receiver cir¬ 
cuits, 47 

trailed wire charge dissipators, 47-48, 
54, 55-56, 59 

voltmeter for high voltage measure¬ 
ments, 49 

Propagation studies, radio waves 
choice of transmitter and receiver 
locations, 6 

effect of antenna heights, 6 
effect of polarization, 19-20 
effect of type of terrain, 5-6, 7 
ionosphere studies, 131-133 
over land at microwave frequencies, 
20-21 

radiotelephone communication be¬ 
tween mobile units, 6-7 
Psycho-Acoustic Laboratory, Harvard 
University, 8 

PTM (pulse time modulated) emission, 
definition, 11 

Pulse amplitude modulated (PAM) 
emission, definition, 11 
Pulse control, microwave telephone 
transmitter, 30 

Pulse frequency modulated (PFM) 
emission, definition, 11 
Pulse generator, r-f generator, 42-43 
Pulse jamming 

in a-m systems, 8, 9 
in f-m systems, 9 

Pulse length modulated (PLM) emis¬ 
sion, definition, 11 


Pulse number modulated (PNM) emis¬ 
sion, definition, 11 

Pulse position modulated (PPM) emis¬ 
sion, definition, 11 
Pulse reception 

by panoramic receivers, 85-86 
moving screen panoramic receiver, 
96-97 

pulse scanning receiver, 108-116 
Pulse scanning receiver, 350-750 me, 
108-116 

alarm and motor control, 112-115 

antenna, 110 

a-v-c circuit, 113 

converter, 110 

detector, 113 

duty cycle, 109 

energy distribution in radar pulses, 
108 

filter width, 108 
i-f amplifiers, 113 
indicating systems, 109, 110 
monitor and sweep circuits, 115 
oscillator, 111-113 
receiver characteristics, 115 
recommendations for future research, 
115-116 

signal-to-noise ratio, 109 
video amplifier, 113 
Pulse time modulated (PTM) emission, 
definition, 11 

Pulse transmitter, omnidirectional 
microwave telephone, 28-31 
audio limiter, 30 
circuit operation, 30-31 
magnetron, 28-29 
output circuit, 31 
pulse control, 30 
wavemeter, 31 
Pulsed wave emission, 11-15 

band space requirements, 11-12, 13- 
14 

bandwidth formula, 12 
for relay and trunkline service, 15 
time division multiplexing, 12-13 
types of modulation, 11 
Pulse-modulated telephone generator 
see R-f generator, 2,000 to 3,000 me 
Purdue Research Foundation, 51 


of frequency stabilized oscillator, 
159-160 

of magnetic core materials, 198 
Quartz crystal substitutes for frequency 
control, 173-174 

Radar pulse reception, panoramic re¬ 
ceivers, 85-86 

Radar pulse scanning receiver 

see Pulse scanning receiver, 350-750 
me 

Radio navigation, use of panoramic re¬ 
ceivers, 87 
Radio propagation 
see Propagation studies, radio waves 
Radio transmission, ionosphere studies, 
131-135 

CONFIDENTIAL 


Radioactive cups for precipitation static 
dissipation, 48 

Radiotelephone communication 
antenna characteristics, 7 
between mobile units, 6-7 
choice of frequency, 6, 7 
directional microwave telephone, 32- 
36 

field intensity studies, 7 
omnidirectional microwave telephone, 
27-32 

studies of effective speech jamming, 
119-120 

Rain, electrostatic charge from, 58 
Random noise, 8-10 
Range, maximum 
for f-m systems, 9 

for microwave communications, 14 
microwave propagation studies over 
salt water, 19 

of directional microwave telephone, 
34, 36 

of omnidirectional microwave tele¬ 
phone system, 32 

Range expansion, panoramic reception, 
82 

RCD-23 microwave receiver, 25 
Reactance 
see Impedance 
Receivers, radio 

see also Panoramic receivers 
0.1 to 30 me, 104-108 
3 to 10 me, 99-100 
300 to 1,000 me, 139 
350 to 750 me, 108-116 
1,000 to 3,000 me, 139 
antenna for microwave receivers, 32 
apparatus for ionosphere field and 
intensity studies, 132 
CR-301 beam deflection converter 
receiver, 25 
effect of mistuning, 8 
for omnidirectional telephone system, 
31-32 

frequency stability, 24, 25, 32 
input noise characteristics, 24, 25 
location for radio transmission over 
broken country, 6 
microwave crystal receivers, 21, 24 
RCD-23 receiver, 24-25 
Recommendations for future research 
camouflage paint binder for reduc¬ 
tion of precipitation static, 52 
charging rate of semiconducting rub¬ 
ber, 52 

charging rates measurements on air¬ 
craft wing surface, 52 
impedance measuring equipment, 201 
noise measurement studies, 195 
panoramic reception, 107, 115-116 
quartz crystal substitutes for fre¬ 
quency control, 173-174 
two dimensional mosaic television 
pickup tube, 165 

universal microwave communication 
equipment, 15-16 
Recorders 

for aircraft facsimile system, 141 









218 


INDEX 


for ionosphere field intensity studies, 
132 

Recording, magnetic tape 
see Magnetic recording 
Recording head, pocket sized wire re¬ 
corder, 169 

Recording radio telegraph code from 
panoramic indicator, 77-78, 83- 
85 

Refraction studies, microwave, 21 
Rejection circuits for panoramic recep¬ 
tion, 82 

Repetition rates 

for frequency diagrams, panoramic 
indicators, 76 

for panoramic facsimile recorder, 84 
Resistance 
see Impedance 
Resistance noise 

for jamming speech signals, 119, 120, 
121 

for jamming telegraph, 120-121 
interference criteria, 123 
methods of generation, 121 
Resolution in panoramic receivers, 76- 
77 

R-F generator, 2,000 to 3,000 me, 37-43 
automatic tuning, 38 
CFVD modulation, 41 
magnetic focusing, 37 
monitor cavity, 38-39 
oscillator tube, 37 
power supply, 43 
pulse generator, 42-43 
Rochelle salt crystals for frequency 
control, 173 

Runway localizing and marking, use of 
panoramic receivers, 87 

Sawtooth generator for panoramic in¬ 
dicator, 93-94 

Scanning filters, panoramic receivers, 
67-72 

analogy with quasi-stationary filter 
response, 68-70 

bandwidth and resolution for equal- 
level signals, 69-70 
bandwidth formula, 108 
desirable characteristics, 72 
effect of level difference, 70-71 
effect on resolution, 68-69 
general design considerations, 67-68 
in long scale 3- to 10-mc receiver, 
99-100 

location of scanning filter, 72 
radar pulse reception requirements, 
85-86 

use of one filter at two or more 
scanning speeds, 72 
Scanning oscillators in panoramic re¬ 
ceivers 

see Sweep oscillators, panoramic re¬ 
ceivers 

Scanning receiver, 0.1- to 30-mc, 104- 
107 

see also Panoramic receivers 
beating oscillators, 106 
f-m oscillator, 106 


f-m tripler and amplifier receiver, 106 
input filters, 105 
modulators, 106 

monitoring receiver and marker, 107 
r-f amplifier, 106 

SCR-508 f-m transmitter-receiver equip¬ 
ment, 9 
Secrecy, radio 

advantages of flash telegraphy, 144 
use of 116 me for mobile radiotele¬ 
phone communications, 7 
Selection circuits, panoramic reception 
for receivers without frequency sweep, 
86 

frequency allocations, 65-66 
Sensitivity, microwave receivers, 24 
Sensitivity, panoramic receivers, 80-81 
radar pulse reception, 86 
radar pulse scanning receiver, 109 
Shielding diathermy equipment, 175 
Shock excitation, effect on radio re¬ 
ceivers, 47 

Signal amplitude selection circuits, 15 
Signal generator, coding, 1S9-190 
Signal generators 
see Oscillators 

Signal intensity control, panoramic re¬ 
ceivers, 66-67 

Signal strength studies, microwave 
propagation, 21 
Signal-to-noise ratio, 119 

see also Sensitivity, microwave re¬ 
ceivers; Sensitivity, panoramic 
receivers 

advantages of f-m, 8, 15 
effect of i-f bandwidth in a-m sys¬ 
tems, 8 

for radar pulse reception, 86, 109 
improvement using f-m in klystron 
transmitter, 24 

improvement with long persistence 
screen, 76 

scanning filters for panoramic re¬ 
ceivers, 71 

6C4 tube, frequency stability studies, 
156 

Slaytor Electronics Corporation, static 
discharger, 56 

Slit-aperture electron gun, 162, 164 
Snow crystals, effect on airplane static 
charge, 51, 58 

Solar phenomena, ionosphere studies, 
131, 132, 133 
Solar-activity cycles 
effect on radio transmission, 133 
Specific loss factor, magnetic core ma¬ 
terials, 198, 199 

Spectrograms of radio noise, 119 
Spectrum widths 

see Band space requirements 
Speech transmission, effective jamming 
see also Jamming 

continuous vs. interrupted jamming, 
120 

interference criteria, 123 
noise spectrogram, 119 
required frequency spectrum, 119 
required noise-to-signal ratios, 119 


speech mixtures for jamming, 119 
Spiral trace, panoramic receiver indi- 
dication, 99 

Spiral trace, signal presence diagrams, 
panoramic indicator, 74 
“Spiral-four” audio-frequency multi¬ 
plexing, 12 

Square wave approximation of complex 
waveforms, 183-184 
Stabilization of carrier frequency, 16 
Stabilization of microwave carrier fre¬ 
quency, 13 

Stagger-tuned circuits, 196-197 
Standing wave measurements, 

26, 27 

Stanford University, field intensity and 
ionosphere studies, 131 

Static 

see Precipitation static studies 
Storage batteries for low temperature 
use, 181-182 

Storm forecasting from ionosphere stud¬ 
ies, 131-133 

Sun-spot cycles, effect on radio trans¬ 
mission, 133 

Supersonic-frequency multiplexing, 12 
Sweep oscillators, panoramic receivers, 
64-65 

electronically controlled oscillators, 
64-65 

for moving-screen receiver, 90 
mechanically controlled oscillators, 
64 

use of an RC oscillator, 94-95 

Tape recording 

see Magnetic recording 
Telegraph transmission 

flash telegraphy system, 144-153 
interference criteria, 123 
jamming studies, 120-121 
masking studies, 123 
reproduction and reading code from 
panoramic receiver indicator, 
77-78, 83-85, 88 

single-channel teletypewriter system, 
4 

Telephone systems, radar 

directional microwave telephone, 32- 
36 

omnidirectional microwave telephone, 
27-32 

speech jamming studies, 119-120 
Telephone wire laying from airplanes, 4 
Teletypewriter method of flash teleg¬ 
raphy, 145 

Teletypewriter system, single channel, 4 
Television, line mosaic pick-up tube, 
162-165 

Temperature effects on frequency sta¬ 
bility of oscillator circuits, 32, 
159 

350 to 750 me pulse scanning receiver 
see Pulse scanning receiver, 350-750 
me 

“Three-gap” oscillator tube, 37 
3,000 me communication systems, de¬ 
sign 


CONFIDENTIAL 





Ll! 'v ' NATION: BEFORE SERVic^.l 
OR’EEx’RODUCTNG ANY PART OF THiri 
DOCUMENT, all classification 
INDEX MANNINGS MUST BE CANCELLED219 


see Microwave communication sys¬ 
tems, design factors 
Threshold of understandability of 
speech in the presence of con¬ 
tinuous and interrupted noise, 
120 

Thunderclouds indicator, 56, 59 
Time-division multiplexing, 12-13 
Trailed wire charge dissipators, 47-48, 
54-56, 59 

Transmitters, microwave, 22-24 

automatic frequency control, 22-23 
CFVD modulator, 22, 23 
effect of ambient temperature, 22 
frequency stability, 22-23 
in directional microwave telephone, 36 
in omnidirectional telephone system, 
28-31 

loading and tuning range of aircraft 
antennas, 200-201 

location for transmission over broken 
country, 5, 6 
push-to-talk circuit, 24 
r-f power output measurements, 200 
Trap circuits for blanking in panoramic 
reception, 82 

Trees obstructing wave propagation, 
5,22 

Triboelectric series 
camouflage paints, 52 
snow and frost crystals, 52 
Triple heterodyne circuit, 104, 106 


Tuning systems 

adjustable load method, 200 
butterfly circuit, u-h-f oscillators, 
137-138, 139 
C-26 jammer, 124-125 
dummy antenna method of testing, 
200, 201 

frequency stabilized 2-20 me oscil¬ 
lator, 159-161 
r-f generator, 37-42 

U-H-F field intensity measuring equip¬ 
ment, 137-140 

300-1000 me oscillator, 137-139 
1000-3000 me oscillator, 139 
receivers, 140 

University of Puerto Rico, field inten¬ 
sity and ionosphere studies, 131 
Vario-losser circuits, 67 
Velocity-modulated r-f generator, 2000- 
3000 me, 37-43 
V-H-F frequency band 

antenna site selection studies, 4 
c-w Morse operation, 4 
effect of obstructions on radio propa¬ 
gation, 4-6 

for Air Warning Service radio cir¬ 
cuits, 4 

reduction of radio interference, 4 
Video amplifier, radar pulse scanning 
receiver, 112 

Visible speech system for aiding the 


deaf in learning to talk, 77 
Washington State College, 57 
Wave generator for complex wave¬ 
forms, 184 
Wave propagation 

see Propagation studies, radio waves 
Waveguide 

curvature of bends for negligible re¬ 
flection, 25 

for microwave transmitter antenna 
feed, 25 

loss in rectangular guide at 9.8 cm, 
25 

use of “magic T” sections in cross¬ 
band signaling, 16 
Wavemeter 

in microwave telephone transmitter, 
31 

in r-f generator tuning system, 38 
Weather, effect on microwave trans¬ 
mission, 21, 22 
Wire, floating insulated, 192 
Wire, trailing, as charge dissipator for 
precipitation static, 47-48, 54-59 
Wire laying by airplane over jungle 
terrain, 4, 191 

Wire lines, fault location, 176-180 
Wire recorder, pocket sized, 166-169 
d-c erase and bias, 167 
drive mechanism, 166-167, 169 
play back amplifier, 169 
wire details, 167-169 



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